Switched-mode power supply comprising a module for charging and discharging an energy store including an electrical transformer

Abstract

The disclosure concerns a switched-mode power supply comprising a module for charging and discharging an energy store including an electrical transformer. The device provides high configuration flexibility.

Claims

1. A switched-mode electrical power supply device for regulating a DC input voltage provided between two input terminals of the switched-mode electrical power supply device by an electrical network, the switched-mode electrical power supply device comprising: at least one conversion module to convert the DC input voltage into a regulated DC voltage, an energy store module to store electrical energy to yield to the at least one conversion module in case of outage of the DC input voltage, a disconnection module for disconnecting the switched-mode electrical power supply device from the electrical network in case of outage of the DC input voltage, a management module distinct from the disconnection module, wherein the management module is directly connected to first and second switching modules and configured to manage charging and discharging of the energy store module by controlling the first and second switching modules, and an electrical transformer, and wherein: a primary circuit connected between the two input terminals comprises a primary winding of the electrical transformer connected in series with the first switching module controlled by the management module, a closed secondary circuit comprises a secondary winding of the electrical transformer connected in series with the energy store module and the second switching module controlled by the management module, and the closed secondary circuit is closed because an input end of the closed secondary circuit and an output end of the closed secondary circuit are directly coupled to a common electrical node, and wherein the input end is electrically coupled to the secondary winding of the electrical transformer and the output end is electrically coupled to the energy store module.

2. The switched-mode electrical power supply device according to claim 1, wherein the closed secondary circuit is closed on one of the two input terminals.

3. The switched-mode electrical power supply device according to claim 1, wherein the disconnection module comprises a control circuit configured to control a disconnection switch connected in parallel with a diode.

4. The switched-mode electrical power supply device according to claim 3, wherein the disconnection module is connected to one of the two input terminals different to another one of the two input terminals where the secondary winding is closed.

5. The switched-mode electrical power supply device according to claim 3, wherein the disconnection module is connected to a cold point of the switched-mode electrical power supply device.

6. The switched-mode electrical power supply device according to claim 1, wherein the closed secondary circuit is closed on an output terminal of the conversion module supplying the regulated DC voltage.

7. The switched-mode electrical power supply device according to claim 1, wherein at least one of the first and second switching modules has a connection terminal connected to a cold point of the switched-mode electrical power supply device.

8. The switched-mode electrical power supply device according to claim 1, wherein at least one of the first and second switching modules has a connection terminal connected to a hot point of the switched-mode electrical power supply device.

9. The switched-mode electrical power supply device according to claim 1, wherein corresponding ends of the primary and secondary windings are connected to a voltage that is not subjected to switched mode.

10. The switched-mode electrical power supply device according to claim 1, wherein each of the first and second switching modules comprises a transistor connected in parallel with a diode.

11. The switched-mode electrical power supply device according to claim 1, wherein each of the first and second switching modules is referenced to a cold point of the switched-mode electrical power supply device.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) The features and advantages of the present subject matter will be more readily understood from the following detailed description which should be read in conjunction with the accompanying drawings that are given merely by way of explanatory and non-limiting example, and in which:

(2) FIG. 1 illustrates a switched-mode power supply circuit diagram according to the state of the art;

(3) FIG. 2 illustrates a switched-mode power supply circuit diagram according to some aspects of the present subject matter;

(4) FIGS. 3a-3f illustrate simplified circuit diagrams of devices according to some aspects of the present subject matter;

(5) FIG. 4a illustrates a switched-mode power supply circuit diagram that is capable of taking measurements of current by shunt resistors referenced to the ground according to some aspects of the present subject matter; and

(6) FIG. 4b illustrates a control circuit diagram for generating signals according to some aspects of the present subject matter.

DETAILED DESCRIPTION

(7) Input terminals 200, 201 receive a DC input voltage to convert from a DC electrical network (or from a rectified AC network). A conversion module 202 (which may comprise a plurality of converters) regulates that input DC voltage and provides regulated output voltages on the output terminals 203. In order to protect the switched-mode power supply, an energy store 204 is connected to the terminals of the conversion module.

(8) A control circuit 205 (for example a NOT gate) controls a switch 206 (for example a MOSFET transistor) connected in parallel with a diode 207. All the components 205, 206 and 207 form part of a module 208 for connecting and disconnecting the supply of the electrical network providing the DC voltage input.

(9) The storage module is charged and discharged by a charging and discharging module 209.

(10) This module 209 comprises two switching modules 210, 211. For example, each switching module comprises a switch (for example a MOSFET transistor) connected in parallel with a diode. The switches are controlled by a charging and discharging management module 212. The control of the switching modules is simplified by them being referenced to the input terminal of which the voltage is taken as the ground of the device (or “cold point”), for example here the terminal 201. For example, the source of a MOSFET transistor or the emitter of an IGBT transistor capable of being used as a switch, is connected to that cold point.

(11) The charging and discharging module further comprises a transformer 213 with a primary winding 214 and a secondary winding 215.

(12) The primary winding 214 is connected in series with the switching module 210 between the input terminals 200 and 201, so forming a primary circuit. For example, the switching module is connected to the input terminal of which the voltage is taken as the ground of the device (or “cold point”), for example here the terminal 201.

(13) The secondary winding is connected in series with the switching module 211 and the energy store module 204, thus forming a secondary circuit. The secondary circuit is for example closed. It is for example closed on the input terminal of which the voltage is taken as the ground of the device (or “cold point”), for example here the terminal 201. For example, the secondary winding is connected between the switching module 211 and the storage module 204.

(14) The circuit of FIG. 2 relies on an isolated inverter topology, of “flyback” type.

(15) The primary and secondary windings are integrated into circuits connected together (by the cold point). Thus, the parasitic capacitive currents, due to the parasitic capacitance between windings are fed back in a connection path which is as short as possible (they pass into the connection between the two windings, without extending into the remainder of the circuit or even beyond that circuit), and only generate a little electromagnetic disturbance. It is thus possible to have a relatively high parasitic capacitance, and thus to pair-wind two wires in order to limit leakage inductances.

(16) The charging voltage of the energy store may be chosen on the basis of the transformation ratio of the transformer. However, other parameters may be used, for example such as the technology of the capacitance, the storage density, the safety of persons, the switch technology, or other parameter. The “flyback” topology makes it possible to have different ratios of input voltage to output voltage, regardless of the transformation ratio.

(17) According to a first approach, the transformation ratio may be equal to one, that is to say that the two windings of the transformer have the same number of turns. Such a transformation ratio makes it possible to facilitate the production of the inductor and to improve the characteristics thereof, in particular to minimize leakage inductance. The choice of a transformation ratio equal to one does not very strongly constrain the charging voltage of the energy store module. However, as the transformation ratio is fixed, according to the voltage ratio between the network voltage and the charging voltage of the energy module, the circuit may be caused to operate with duty cycles close to 0 or close to 1.

(18) According to a second approach, the transformation ratio may be chosen so as to optimize the circuit duty cycle range to use given the charging voltage of the energy store module chosen. The yield of the charging and discharging module may thus be optimized.

(19) A switched-mode power supply according to the diagram of FIG. 2 may be implemented in accordance with several variants. In particular, the position of the connection and disconnection module on one or other of the input terminals, the arrangement of the primary circuit, the arrangement of the secondary circuit, and the referencing of the secondary circuit, provide degrees of freedom of implementation.

(20) Other degrees of freedom are linked to the galvanic insolation between the primary and secondary windings.

(21) The device according to the disclosure has a multitude of implementations which enable its employment in supply systems with various characteristics. Thus, the supply designer may adapt the device to the specific needs of his or her system. Examples of device variants are presented below with advantages. The advantages presented are not limited to a circuit. Certain advantages presented for one circuit may apply to others.

(22) FIGS. 3a to 3f are simplified circuits of devices according to embodiments. In particular, the charging and discharging module is not represented.

(23) In the circuit of FIG. 3a, the connection and disconnection module 208 is connected to terminal 201 of the device corresponding to the voltage taken as ground (“cold point” of the device).

(24) Module 208 comprises a controlled switch 301 (for example a MOSFET transistor). The switch is controlled by the control circuit 205 connected in parallel with a diode. Relative to module 208 of FIG. 2, the transistor 301 and the diode 302 are connected in reverse to the transistor 206 and the diode 207.

(25) When the switch comprises an N-channel MOSFET transistor and module 208 is connected to the cold point, the transistor may be controlled without having to generate a higher voltage than that of the hot point (voltage of terminal 200).

(26) In the circuit of FIG. 3b, the charging and discharging module 209 comprises, in the primary circuit, a switching module 303 and a primary winding 304 of which the connection order is not the same as in FIG. 2. In the circuit of FIG. 3b, it is the primary winding that is connected to the cold point of the circuit.

(27) In this configuration, the effect of the parasitic capacitances of the switching module switches is reduced. The drain (if is a MOSFET transistor) or the collector (if it is an IGBT transistor) of the switch is connected to a voltage not subjected to switched mode.

(28) Furthermore, the referencing of the switching modules to different voltages (module 211 is referenced to the cold point and not module 303) makes it possible to cancel the effect of the parasitic currents in the transformer, in particular in a transformer with “two pair-wound wires” and in a transformer with a transformation ratio equal to one.

(29) In the circuit of FIG. 3c, the charging and discharging module 209 comprises, in the secondary circuit, a switching module 305 and a secondary winding 306 of which the connection order is not the same as in FIG. 2. In the circuit of FIG. 3c, it is the switching module 305 that is connected between the secondary winding and the storage module 204.

(30) The configuration of the circuit of FIG. 3c provides similar advantages to those of the circuit of FIG. 3b.

(31) In the circuit of FIG. 3d, the secondary circuit is not closed on the voltage of the cold point of the circuit as is the case in the circuit of FIG. 2. The storage module 307 and the switching module 308 of the circuit of FIG. 3d are connected to the output of connection and disconnection module 208. The winding 309 of the secondary is connected in series between the two of them.

(32) In the circuit of FIG. 3e, the secondary circuit is closed on an output terminal of the converter module 202. Thus, the storage module 310 and the switching module 311 are connected to that output terminal. The secondary winding 312 of the transformer is connected in series between the two of them.

(33) In the circuit of FIG. 3f, the secondary circuit, comprising, connected in series, the storage module 313, the switching module 314 and the secondary winding 315 of the transformer, is closed on itself, without being referenced to a voltage of the circuit. The secondary circuit is not connected to any point on the circuit.

(34) The different variants described above offer various advantages. These advantages are not necessarily exclusive of each other, and may be combined in the same circuit.

(35) In particular, in the variants in which a switch controlled at its source (for a MOSFET) or its emitter (for an IGBT) connected to the voltage to which the circuit control is referenced, that switch is simpler to control.

(36) When the drain (for a MOSFET) or the collector (for an IGBT) of a controlled switch is connected to a voltage not subjected to switch mode (as is the case in the circuits of FIGS. 3b and 3c), it is possible to reduce the parasitic currents linked to the parasitic capacitances usually present between the drain or the collector and the surroundings. To be precise, the substrate of the chip constituting the switch is usually the drain or the collector thereof.

(37) When the corresponding ends of the two winding are both connected to a potential not subjected to switched mode (as is the case in the circuits of FIGS. 3b and 3c), the temporal variation dv/dt of the voltage between corresponding turns of the two winding may be cancelled. Thus, a winding with “two pair-wound wires” may be used while having very low capacitive currents. The winding with “two pair-wound wires” makes it possible to minimize the leakage inductance and thus to optimize operation, at the expense of an increase in the parasitic capacitances between the primary and secondary, which usually results in an increase in the parasitic capacitive currents.

(38) Apart from the variant of FIG. 3e, when the converters of the conversion module have galvanic isolation, the parasitic capacitive currents may be fed back by wired pathways internal to the device. Thus, low levels of electromagnetic disturbance are generated outside the assembly.

(39) After having presented architectures of the circuit according to the embodiments, the following portion of the description details the regulation of these circuits.

(40) Generally, to measure the current in the device, it is possible to use a shunt resistor and/or a current transformer.

(41) In particular, it is possible to only take measurements of current by shunt resistors referenced to ground, as illustrated by the circuit of FIG. 4a.

(42) Another advantage is that these shunt resistors deliver voltages that are always positive. This makes it possible to dispense with the need for a negative supply and makes it possible to envision implementations in which only one current comparator is used.

(43) FIG. 4a includes the circuit of FIG. 2 with the same reference signs. Two shunt resistors 401 and 402 have been added, connected between the cold point of the circuit and, respectively, the switching modules 210 and 208. The switching modules 210 and 208 are respectively controlled by signals) X.sub.ch, X.sub.dch for charging and for discharging the energy store module.

(44) The circuit control strategy is described below, with reference to FIG. 4b which represents a control circuit for generating the signals X.sub.ch and X.sub.dch for controlling the switching modules 210 and 208.

(45) The voltage u.sub.r* represents the control signal for the store voltage u.sub.r at the terminals of the energy store module 204. The voltage u.sub.b* represents the control signal for the bus voltage u.sub.b at the terminals of the conversion module and of the charging and discharging module 209.

(46) The voltage u.sub.r is supplied to a shaping module 451 of which the output is compared to the voltage u.sub.r* by a comparator 452. Similarly, the voltage u.sub.b is supplied to a shaping module 453 of which the output is compared to the voltage u.sub.b* by a comparator 454. The comparators apply respective gains to the comparison of their input signals.

(47) The store voltage control signal u.sub.r* is chosen so as to protect the energy store module (comprising for example a capacitor), and to store sufficient energy therein. The bus voltage control signal u.sub.b* is chosen at the limit value (of network voltage) above which the energy of the network is used, and below which the energy of the store is used.

(48) The operation of the charging and discharging module is governed by a single control value i*, which is the control signal for current in the winding 214 of the primary circuit, the latter being defined by:
i=i.sub.2−(n.sub.1/n.sub.2)×i.sub.1

(49) n.sub.1 and n.sub.2 being the number of respective turns of the two windings, primary 214 and secondary 215, of the transformer.

(50) The control signal for current is saturated positively at the value i.sub.max* and negatively at the value i.sub.min*.

(51) The current i.sub.b*, supplied at the output of the comparator 454 represents the control signal for current established in order to regulate the bus voltage. The current i.sub.r*, supplied at the output of the comparator 452 represents the control signal for current established in order to regulate the store voltage.

(52) The output of the comparator 454 is supplied to a Schottky diode 455. The control signal for current i*, at the output of the Schottky diode 455, is equal to that one of the two control signals i.sub.b* and i.sub.r* which is algebraically the greatest (that is to say which has the greatest capability for a transfer of energy from the energy store module to the conversion module).

(53) To be precise, if the store voltage is greater than the control signal (i.sub.r*>0) and the bus voltage also (i.sub.b*0), it is necessary to discharge the store (i*>0), and not the bus. Similarly, if the store voltage is less than the control signal (i.sub.r*<0) and the bus voltage also (i.sub.b*>0), it is necessary to discharge the store (i*>0), and not the bus.

(54) The output from the comparator 452 is connected to a current limitation module 456, via a resistor 457. The output from the comparator 455 is also connected, via the Schottky diode 455, to the current limitation module.

(55) The association between the resistor 457 and the diode 455 enables arbitration between the control signals for current.

(56) Depending on the amplitude and the sign of the current i*, the current limitation module enables the charging or discharging of the energy store module. The output from module 456 is provided as input to three comparators 458, 459, 460 and 461. The output from module 456 is supplied to comparator 461 via a gain module 462 of which the gain value depends on transformation ratio of the transformer: −(n.sub.1/n.sub.2).

(57) The comparators 459 and 458 make it possible to avoid inadvertently commanding the charging and discharging of the energy storage module. They furthermore make it possible to avoid simultaneously commanding charging and discharging.

(58) Thus, the comparator 458 compares the output from module 456 with a minimum discharge current value i.sub.dch to provide a signal en.sub.dch as output. The comparator 459 compares the output from module 456 with a minimum charge current value i.sub.ch to provide a signal en.sub.ch as output. Thus, the charging and discharging of the energy storage module are only commanded when the signal output from module 456 is significant enough.

(59) The signals en.sub.dch and en.sub.ch are provided to a synchronization logic to synchronize the orders for charging and discharging the energy storage module with a clock signal.

(60) The signal en.sub.dch is provided as input to the input D of a first D latch 463. The signal en.sub.ch is provided as input to the input D of another D latch 464. The respective inputs C of the latches 463 and 464 receive a clock signal from a clock 465 which is inverted by an inverting gate 466. The output Q from latch 463 is supplied as input to a NAND gate 467 and the output Q from the latch 464 is provided as input to a NAND gate 468. The D latches 463 and 464 are conventional. A detail of their wiring is represented in window 469.

(61) Control of the switching modules 208 and 210 depends in particular on the control signals u.sub.r and u.sub.b, and on the measurements of current in the shunt resistors 401 and 402.

(62) Thus, the comparator 460 compares the output from the current limitation module 456 with the voltage u.sub.i2 representing the current passing through the shunt resistor 402 while the comparator 461 compares that output, amplified by the gain (−(n.sub.1/n.sub.2)) of the gain module 462, with the voltage u.sub.i1 representing the current passing through the shunt resistor 401. The output from the comparator 461 is supplied as input to the NAND gate 468 and the output from the comparator 460 is provided as input to the NAND gate 467.

(63) The charging X.sub.ch and discharging X.sub.dch control signals are supplied by SR flip-flops 469 and 470.

(64) The SR flip-flop 469, which delivers the signal X.sub.ch at its output Q, receives, on its R input (reset) the inverted output from the NAND gate 468. It furthermore receives on its S input (set) the clock signal from clock 465. The SR flip-flop 470, which delivers the signal X.sub.dch at its output Q, receives, on its R input (reset) the inverted output from the NAND gate 467. It furthermore receives on its S input (set) the clock signal from clock 465. The SR flip-flops 469 and 470 are conventional. A detail of their wiring is represented in window 471.

(65) The control arrangement with the SR flip-flops makes it possible to control the switching modules 208 and 210 in Peak Current Mode.

(66) Thus, in a regulation mode, it is the bus voltage which is regulated and in another regulation mode, it is the store voltage which is regulated. The passage from one regulation mode to the other is directly determined by the control signal values i.sub.b* and i.sub.r*.

(67) There is an operating point common to both regulation modes when i.sub.b*=i.sub.r*.

(68) The passage from one regulation mode to the other is not strictly equivalent to the change in sign of the control signal for current i*.

(69) This strategy has several advantages.

(70) The construction of the control signal i* as choice of the largest of the two control signals i.sub.b* and i.sub.r* ensures that the passage from one mode to the other (from i*=i.sub.b* to i*=i.sub.r* and vice-versa) does not introduce any discontinuity into i*, and therefore no disturbance of the system.

(71) This furthermore enables an operation without oscillation at the intersection between the two modes. In particular, when the network is of high impedance and/or the charging current very great, if the current absorbed on the network to charge the store module with energy makes the network voltage drop to the neighborhood of u.sub.b*, the bus voltage regulation enters into play even though the energy store module is still in course of charging (with a reduced current), so avoiding a more significant collapse of the network or possible oscillations. This property makes it possible to adjust the store charging current (via i.sub.min*) so as to obtain a very fast charge when the network is of low impedance, without this posing any problem when the network is of very high impedance.

(72) The existence of two separate voltage loops enables different regulation according to needs. To be precise, the presence of capacitors with different values on the two voltages to regulate may require different gains.

(73) Of course, the present disclosure is not limited to the described embodiments, other variants and combinations of features are possible. The various components described may be substituted by one or more other components configured to fulfill the functions described for each component of the circuits described above.