Reference signal construction

Abstract

There is disclosed a radio node for a wireless communication network, the radio node being adapted for processing reference signaling based on a coding, the coding being based on a Golay sequence. The disclosure also pertains to related devices and methods.

Claims

1. A radio node for a wireless communication network, the radio node being configured for: pruning a predetermined number of elements from a Golay sequence, the Golay sequence having a total number of elements; zero-padding the total number of elements of the Golay sequence; and processing reference signaling based on a coding, the coding being based on the pruned Golay sequence and the zero-padded total number of elements.

2. The radio node according to claim 1, wherein the coding is based on a pair of complementary Golay sequences comprising a first sequence and a second sequence.

3. The radio node according to claim 1, wherein the coding maps first reference signaling to a first frequency range and a second reference signaling to a second frequency range.

4. The radio node according to claim 1, wherein elements of a Golay sequence are determined differentially.

5. The radio node according to claim 1, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

6. A method for operating a radio node in a wireless communication network, the method comprising: pruning a predetermined number of elements from a Golay sequence, the Golay sequence having a total number of elements; zero-padding the total number of elements of the Golay sequence; and processing reference signaling based on a coding, the coding being based on the pruned Golay sequence and the zero-padded total number of elements.

7. The method according to claim 6, wherein the coding is based on a pair of complementary Golay sequences comprising a first sequence and a second sequence.

8. The method according to claim 6, wherein the coding maps first reference signaling to a first frequency range and a second reference signaling to a second frequency range.

9. The method according to claim 6, wherein elements of a Golay sequence are determined differentially.

10. The method according to claim 6, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

11. A computer storage device storing executable instructions, when executed, the instructions causing processing circuitry to at least one of perform and control a method for operating a radio node in a wireless communication network, the method comprising: pruning a predetermined number of elements from a Golay sequence, the Golay sequence having a total number of elements; zero-padding the total number of elements of the Golay sequence; and processing reference signaling based on a coding, the coding being based on the pruned Golay sequence and the zero-padded total number of elements.

12. The radio node according to claim 2, wherein the coding maps first reference signaling to a first frequency range and a second reference signaling to a second frequency range.

13. The radio node according to claim 12, wherein elements of a Golay sequence are determined differentially.

14. The radio node according to claim 13, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

15. The radio node according to claim 2, wherein elements of a Golay sequence are determined differentially.

16. The radio node according to claim 15, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

17. The radio node according to claim 3, wherein elements of a Golay sequence are determined differentially.

18. The radio node according to claim 2, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

19. The radio node according to claim 3, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

20. The radio node according to claim 4, wherein reference signaling comprises at least one of primary synchronisation signaling and secondary synchronisation signaling.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) The drawings are provided to illustrate the approaches and concepts described herein, and are not intended to limit their scope. They comprise:

(2) FIG. 1, showing PSS and SSS as in LTE FDD release 8;

(3) FIG. 2, showing initial cell search procedure for LTE release 8;

(4) FIG. 3, showing detectors for PSS and SSS;

(5) FIG. 4, showing receiver correlation with timing and frequency offset candidates,

(6) FIG. 5, showing autocorrelations of the two complementary sequences and the sum of these autocorrelations,

(7) FIG. 6, showing correlation with two Golay complementary sequences;

(8) FIG. 7, showing an exemplary Golay sequence;

(9) FIG. 8, showing autocorrelation of a repeated Golay sequence;

(10) FIG. 9, showing an exemplary Golay sequence with timing and frequency offset candidates;

(11) FIG. 10, showing a block based synchronization signal transmitter using DFTS-OFDM;

(12) FIG. 11, showing a linear synchronization signal transmitter using frequency shift;

(13) FIG. 12, showing a receiver for differential encoded sequences with block-based transmission;

(14) FIG. 13, showing a receiver of two frequency shifted differential encoded sequences;

(15) FIG. 14, showing pruning a differential sequence in time (before DFT);

(16) FIG. 15, showing zero padding in time (before DFT);

(17) FIG. 16, showing autocorrelation of a differential decoded Golay sequence with frequency and timing errors;

(18) FIG. 17, showing an exemplary radio node;

(19) FIG. 18, showing an exemplary method for operating a radio node; and

(20) FIG. 19, showing a further exemplary radio node.

DETAILED DESCRIPTION

(21) Synchronization sequences respective associated signaling are described as example for reference signaling in the following. Herein, a radio node like a UE is referred to as a receiver receiving and decoding/detecting signaling, and a radio node like a base station or eNB (LTE base station) or gNB (NR base station) is referred to as transmitter transmitting and/or encoding.

(22) When a UE is powered on, or when it moves between cells in LTE release 8, it receives and synchronizes to downlink signals in a cell search procedure. The purpose of this cell search is to identify the best cell and to achieve time and frequency synchronization to the network in downlink (i.e. from base station to UE). The Primary and Secondary Synchronization Signals (PSS and SSS) are used at cell search in the UE for LTE. Here, in the case of FDD (Frequency Division Duplexing), the PSS is transmitted in the last OFDM symbol of slots 0 and 10 within a frame and the SSS is transmitted in the OFDM symbol preceding PSS, see the illustration in FIG. 1. In the case of TDD (Time Division Duplex), the PSS is transmitted in the third OFDM symbol of slots 3 and 13 within a frame, and the SSS is transmitted in the last OFDM symbol of slots 2 and 12, i.e., three symbols ahead of the PSS.

(23) Detectors are discussed in the following. A detector may be implemented in a receiver, e.g. a user equipment or terminal. A detector may generally be adapted for detecting reference signaling. Detecting reference signaling may be considered to comprise identifying the signals or symbols belonging to a reference signaling sequence of signals or symbols.

(24) A simplified initial cell search procedure is illustrated in FIG. 2. Here the UE tries to detect PSS from which it can derive the cell identity within a cell-identity group, which consists of three different cell identities corresponding to three different PSS. In this detection, as shown in the illustration in FIG. 3, the UE thus has to blindly search for all of these three possible cell identities. This is done by calculating a cross-correlation between received signals and the candidate PSS sequence. Here a cross-correlation is calculated in a time domain matched filter, followed by an absolute square operation.

(25) The UE also achieves OFDM symbol synchronization and a coarse frequency offset estimation with an accuracy of about 1 kHz. The latter is estimated by the UE by evaluating several hypotheses of the frequency error. Many operations have to be done for each of these frequency offset hypothesis. Typically, the lowpass filter, three time domain matched filters, and three absolute square operations have to be done for each hypothesis. Then the search for peak is done over all frequency offset hypothesis and cell id groups.

(26) The UE can then continue to detect SSS (coherent detection thanks for the PSS decoding) from which it acquires the physical cell id and achieves radio frame synchronization, see FIG. 3. Here, the UE also detects if normal or extended cyclic prefix is used. If the UE is not preconfigured for either TDD or FDD, the UE can detect the duplex mode by the position in the frame of detected SSS in relation to detected PSS. Fine frequency offset estimation can be estimated by correlating PSS and SSS. Alternatively, this fine frequency offset estimation is estimated by using the Cell specific Reference Signals (CRS) which are derived from the Physical Cell Identity (PCI) encoded in the PSS/SSS. Once the UE is capable of decoding the CRSs the UE can receive and decode cell system information which contains cell configuration parameters starting with the Physical Broadcast Channel (PBCH).

(27) Frequency errors can occur due to imperfect oscillators and movement of the UE relative to the base station. Typically, frequency errors up to 20 ppm (part per million) of the carrier frequency must be handled during initial cell search when receiving synchronization signals such as PSS. For a carrier frequency of 2.5 GHz, the frequency error can thus be +−50 kHz. As mentioned earlier, this frequency error can be estimated by detecting PSS for several pre-rotations with different candidates of the frequency errors. A few illustrations of receiver cross correlations in the detector are given in FIG. 4

(28) The drawings are provided to illustrate the approaches and concepts described herein, and are not intended to limit their scope. They comprise:

(29) FIG. 1, showing PSS and SSS as in LTE FDD release 8;

(30) FIG. 2, showing initial cell search procedure for LTE release 8;

(31) FIG. 3, showing detectors for PSS and SSS;

(32) Figure, with frequency error candidates between +−50 kHz, for a Zadoff-Chu sequences as used in LTE release 8. The correct frequency and timing errors are 0 Hz and 0 microseconds, respectively, in these illustrations.

(33) Golay complementary sequences are discussed in the following.

(34) Golay complementary sequences were introduced by Marcel Golay. These sequences are interesting for reference signaling, in particular in the context of synchronization, for at least two reasons:

(35) 1. The sum of the autocorrelations from a complementary pair is ideal

(36) 2. Low computational complexity when calculating correlations A pair of Golay complementary sequences may be denoted as
{a(0),a(1), . . . ,a(N−1)}  (1)
and
{b(0),b(1), . . . ,b(N−1)}.  (2)
Also, the aperiodic autocorrelations of these sequences are denoted as

(37) $\begin{array}{cc}{C}_a\left(k\right)=\underset{n=0}{\overset{N-k-1}{.Math.}}a*\left(n\right)a\left(n+k\right)\mathrm{and}& \left(3\right)\\ C_b\left(k\right)=\underset{n=0}{\overset{N-k-1}{.Math.}}b*\left(n\right)b\left(n+k\right).& \left(4\right)\end{array}$
and
C.sub.b(k)=Σ.sub.n=0.sup.N-k-1b*(n)b(n+k).  (4)

(38) Then the sum of these autocorrelations is a perfect “Dirac delta function” i.e.
C.sub.a(k)+C.sub.b(k)=δ(k)  (5)
where δ(k)=1 for k=0 only and zero otherwise, which is illustrated in FIG. 5. Equation 3 to 5 may be considered to define (complementary) Golay sequences.

(39) A second property of Golay complementary sequences is that the computational complexity for calculating a correlation is low. A sequence diagram is given in FIG. 6 from which correlations with two complementary sequences are calculated simultaneously.

(40) A Golay sequence may be used for timing synchronization as illustrated in FIG. 7. Here
G.sub.a.sub.128={a(0),a(1), . . . ,a(N−1)}  (6)
is a 128 symbol long Golay sequence. This sequence is repeatedly transmitted 16 times within the short training field, as illustrated in FIG. 7, followed by one transmission with changed sign. A very simple receiver, as in FIG. 6, can be constructed in which the received signal is correlated with the known 128 samples long Golay sequence.

(41) The received signal in the detector will include a few of the last samples of a transmitted Golay sequence followed by the first samples of an identical Golay sequence in most correlations. This will result in a non-ideal correlation, as illustrated in FIG. 8.

(42) A few illustrations of receiver correlations in the detector are given in FIG. 9, with frequency error candidates between +−50 kHz, for a three different Golay sequences.

(43) The use of Zadoff-Chu as in current LTE sequences as synchronization signals is very sensitive to frequency errors. The autocorrelations, both with and without frequency errors have many false peaks which cause detections for other than the true timing and frequency error. This leads to timing and frequency offsets after synchronization. This is visible in FIG. 4 as additional peaks for nonzero delays and frequency offsets. False peaks and estimation errors also occur when using a Golay sequence as the synchronization signal, as shown in FIG. 9.

(44) For a variant, it is proposed constructing a time synchronization and/or beam finding reference signal by two differentially encoded complementary sequences which are multiplexed in frequency into one time interval.

(45) Reference signaling, in particular a synchronization signaling sequence, may be constructed which has a detector with very low correlation peaks for non-zero delay errors has a detector which is robust against frequency errors has a detector with very low computational complexity Transmitters are discussed in the following. A transmitter may in particular be implemented as, and/or as part of, radio node like a network node.

(46) An illustration is given in FIG. 10 of a transmitter for the proposed method of processing (constructing) a reference signal from differential encoded Golay complementary sequences. Here two DFTs (Discrete Fourier Transform), one for each of the complementary sequences, are followed by a concatenation and an inverse FFT (Fast Fourier Transform). This construction of a DFT followed by a larger IFFT is also known as DFTS-OFDM and is block based since a block of the transmitted signal is constructed within one IFFT.

(47) An alternative transmitter structure is illustrated in FIG. 11, where an oversampling and interpolation is followed by a frequency shift, which modulates (at least) one of the differential encoded sequences such that they span different, essentially non-overlapping, frequency intervals. This transmitter is linear since all processing is done with linear processing blocks in a sample-by-sample manner.

(48) The construction by DFTS-OFDM in FIG. 10 may be beneficial for base-station implementations built with an IFFT close to the antenna ports. Here, a construction with an IFFT close to the antenna ports can be a software implementation build upon all downlink signals processed with IFFTs. Also, the base station might be designed with IFFTs in hardware which is physically placed close or inside the radio equipment such that having a signal which is not processed with an IFFT would add a significant amount of extra software or hardware support.

(49) The block of signal values constructed by the DFTS-OFDM in FIG. 10 is cyclic within the OFDM symbol, in contrast to the linear processing as illustrated in FIG. 11. A receiver structure is presented next which is based on linear processing. The receiver is thus constructed for a linear signal and will have some performance loss when combined with the block based cyclic signals as compared to using a linear transmitter.

(50) A receiver is discussed in the following. A receiver may in particular be implemented as, and/or as part of, a radio node like a user equipment or terminal.

(51) The signals generated by the two transmitter versions are preferable decoded using corresponding matching receiver formulations; a mismatch will lead to slightly degraded performance.

(52) Block-based generated signaling is discussed in the following, as an example of processing reference signaling.

(53) The block-based transmitter in FIG. 10 is matched by a receiver depicted in FIG. 12. The frequency-domain signal is separated into the two constituent DFT components and converted back to the original sequences. Differential decoding operations are followed by correlation stages with the two complementary Golay sequences. Finally, the two correlations are added before detection.

(54) If several possible synchronization sequences exist, the correlation stage pairs are repeated for each such sequence, while the FFT processing complexity does not change.

(55) To extract the synchronization signal contents in the frequency domain, the FFT input samples must cover the entire OFDM symbol. Since the symbol timing is not known a priori, a simple solution is to perform an FFT and repeat the processing above at several sample timing candidates, which however may be computationally expensive.

(56) To remove the FFT window placement uncertainty, the approach of oversampled FFT may be used. The receiver places double-length, 2N-FFT windows in time domain (2N is an example, other FFT length longer than N work as well), with adjacent FFTs overlapping by N samples (the symbol length). That way, every second FFT window captures an entire OFDM symbol. The transmitted OFDM symbol may contain reference symbols (not shown in FIG. 10) to estimate and remove the resulting frequency rotation, effectively restoring the proper symbol timing. This approach works particularly well if the neighboring OFDM symbols are known to not to contain the sequence.

(57) Another possible variant of the receiver for the block-based signal applies the sliding window FFT approach: a full FFT is performed at an initial time, but at each subsequent sample time, only FFT (DFT) output contribution (negative contribution) due to one new input sample (due to one input sample falling outside the new DFT window) is computed, while a linearly increasing exponential factor is applied to the corrected FFT output values to account for the time shift of the previous input samples. The number of multiplications for each FFT update is O(N), more specifically proportional to the allocated subcarriers in the FFT.

(58) To formulate yet another variant for the receiver, it should be noted that the transmitter operations Golay encoding-Diff encoding.fwdarw.DFT.fwdarw.IFFT create an N-sample long time-domain sequence. A matched filter may be implemented by correlating the received signal with this time-domain sequence, which provides an alternative receiver. However, the correlation complexity is higher since the Golay sequence properties are not utilized.

(59) Linearly generated signaling is discussed in the following as example of determining a coding.

(60) A receiver structure for the linear transmitter in FIG. 11 is illustrated in FIG. 13, where two different bandpass filters are used to extract the different frequency allocations for the two differentially encoded Golay sequences. Again, differential decoding operations are followed by correlation stages with the two complementary Golay sequences. Finally, the two correlations are added before detection.

(61) The linear receiver processing amounts to time-domain matched filtering and correlation over a sliding window in time. If several possible synchronization sequences exist, the correlation stage pairs are repeated for each such sequence and complexity is proportional to the number of sequences.

(62) Detailed descriptions of some of the above processing steps for the linear transmitter and receiver realizations of FIG. 10 and FIG. 13 are given below.

(63) Differential encoding is discussed in the following. Differential encoding may be seen as implementation of differentially determining elements.

(64) Example Golay sequences may be denoted as:
{a(0),a(1), . . . ,a(N−1)}  (7)
or
{b(0),b(1), . . . ,b(N−1)}.  (8)

(65) Equation (7) may represent a first Golay sequence, Equation (8) a second Golay sequence. Reference signaling may be based on one of such sequences, or on more than one. In particular, the first and second sequence may be complementary Golay sequences.

(66) Differential coded Golay sequences may be defined as:

(67) $\begin{array}{cc}\tilde{a}\left(n\right)=\frac{a\left(n-1\right)}{\tilde{a}*\left(n-1\right)}\mathrm{and}& \left(9\right)\\ \tilde{b}\left(n\right)=\frac{b\left(n-1\right)}{\tilde{b}*\left(n-1\right)},& \left(10\right)\end{array}$
with ã(0)=1; {tilde over (b)}(0)=1 for n=1 . . . N. Other differential encoding (and corresponding decoding schemes) are also possible, e.g. utilising a constant factor, and/or referring to different preceding elements.

(68) Modulation (encoding) is discussed in the following.

(69) Sub-carrier mapping (mapping to frequency resources) may be formulated as rotation (with starting sub-carrier N.sub.c), as an example of encoding or determining coding:
ā(n)=ã(n)  (11)
and
b(n)={tilde over (b)}(n)e.sup.j2πnN.sup.c.sup./N.sup.fft  (12)

(70) The sub-carrier shift is typically selected as N.sub.c=N+1. Selection of larger values (than N.sub.c=N+1) may be beneficial in terms of better separation (and less interference between) the two differentially coded Golay sequences.

(71) An exemplary channel is discussed in the following.

(72) The radio channel is a model of the propagation from transmission (in the base station) to the receiver (in the UE). Here a simple channel model is used, equal to a constant complex scaling, delay n.sub.0 and additive noise and interference. Within the current section, this radio channel also includes the interpolation, bandpass filters and decimations. The same model is also used both for the DFTS-OFDM in FIG. 10 and the frequency shift transmitters in FIG. 11.

(73) The two complementary sequences are modelled separately as
{tilde over (r)}.sub.a(n)=hā(n−n.sub.0)+w.sub.a(n)  (13)
and
{tilde over (r)}.sub.b(n)=hb(n−n.sub.0)+w.sub.b(n)  (14)

(74) De-rotation is discussed in the following. De-rotation may be part of decoding and/or receiving. The radio node may comprise a derotation module for such derotation, and/or may be adapted for using the processing circuitry therefor.

(75) Compensate for the sub-carrier shift (or sub-carrier mapping) by
{tilde over (r)}.sub.a(n)=r.sub.a(n)  (15)
and
{tilde over (r)}.sub.b(n)=r.sub.b(n)e.sup.j2πnN.sup.c.sup./N.sup.fft  (16)

(76) Inserting (11) and (13) into (15) results in
{tilde over (r)}.sub.a(n)=hã(n−n.sub.0)+w.sub.a(n)  (17)
Also, inserting (12) and (14) into (16) results in
{tilde over (r)}.sub.b(n)=e.sup.−j2π.sup.0.sup.N.sup.c.sup./N.sup.ffth{tilde over (b)}(n−n.sub.0)+w.sub.b(n)  (18)

(77) Note that the received, de-rotated signal in (16) still has a complex exponential factor which depends on the unknown delay n.sub.0. This complex exponential will be eliminated by the following differential decoding.

(78) Differential decoding is discussed in an example in the following. Differential decoding may be part of, and/or an implementation of, differentially determining a coding.

(79) The two complementary Golay sequences are differentially decoded according to
r.sub.a(n)={tilde over (r)}.sub.a(n+1)={tilde over (r)}.sub.a*(n)=|h|.sup.2a(n−n.sub.0)+w.sub.a(n)w.sub.a*(n−1)  (19)
and
r.sub.b(n)={tilde over (r)}.sub.b(n+1)={tilde over (r)}.sub.b*(n)=|h|.sup.2b(n−n.sub.0)+w.sub.b(n)w.sub.b*(n−1).  (20)

(80) Correlations are discussed in the following.

(81) Correlation with the two complementary Golay sequences can be done with the receiver structure as illustrated in FIG. 6, which corresponds to

(82) $\begin{array}{cc}{\hat{C}}_a\left(k\right)=\underset{n=0}{\overset{N-k-1}{.Math.}}a*\left(n\right)r_a\left(n+k\right)\mathrm{and}& \left(21\right)\\ {\hat{C}}_b\left(k\right)=\underset{n=0}{\overset{N-k-1}{.Math.}}b*\left(n\right)r_b\left(n+k\right)& \left(22\right)\end{array}$

(83) Detection is discussed in the following.

(84) Detection of timing equals the delay k, corresponding to the largest absolute square of the sum of these correlation, i.e.

(85) $\begin{array}{cc}\hat{k}=\arg \underset{k}{\max }{.Math.{\hat{C}}_a\left(k\right)+{\hat{C}}_b\left(k\right).Math.}^2& \left(23\right)\end{array}$

(86) Aspects of DFTS-OFDM are discussed in the following.

(87) The detection of the synchronization signals in the receiver, as illustrated in FIG. 13, is very much simplified if the decimation is done by an integer factor. For the transmitter based on DFTS-OFDM, as illustrated in FIG. 10, this requirement of integer decimations leads to restrictions in the relation between the length of the DFT and the length of the IFFT. Denote the size of the DFT as N.sub.DFT such that

(88) $\begin{array}{cc}\tilde{A}\left(k\right)=\underset{n=0}{\overset{N_{\mathrm{DFT}}-1}{.Math.}}\tilde{a}\left(n\right)e^{-\frac{j2\pi kn}{N_{\mathrm{DFT}}}}\mathrm{and}& \left(24\right)\\ \tilde{B}\left(k\right)=\underset{n=0}{\overset{N_{\mathrm{DFT}}-1}{.Math.}}\tilde{b}\left(n\right)e^{-\frac{j2\pi kn}{N_{\mathrm{DFT}}}},& \left(25\right)\end{array}$

(89) Also, denote the size of the IFFT as N.sub.IFFT such that

(90) $\begin{array}{cc}\stackrel{\_}{s}\left(n\right)=\underset{k=0}{\overset{N_{\mathrm{DFT}}-1}{.Math.}}\tilde{A}\left(k\right)e^{\frac{j2\pi kn}{N_{\mathrm{IFFT}}}}+\underset{k=0}{\overset{N_{\mathrm{DFT}}-1}{.Math.}}\tilde{B}\left(k\right)e^{\frac{j2\pi N_{c}n}{N_{\mathrm{IFFT}}}}{e}^{\frac{j2\pi kn}{N_{\mathrm{IFFT}}}},& \left(26\right)\end{array}$

(91) Note that the summations above are only needed over the N.sub.DFT values of Ã(k) and {tilde over (B)}(k). The decimation in the receiver can now be expressed as

(92) $\begin{array}{cc}D=\frac{N_{\mathrm{IFFT}}}{N_{\mathrm{DFT}}}& \left(27\right)\end{array}$

(93) Typical examples are N.sub.DFT=64 and N.sub.IFFT=2048 such that D=32.

(94) From the design of Golay sequences, the length of a Golay sequence is typically
N=2.sup.m10.sup.k13.sup.i  (28)

(95) With m=6, k=0, l=0 follows that N=64.

(96) However, within the differential encoding, the length of sequence “over air” increases with one such that a sequence of 64 values increase to 65 values. Two different solutions are listed below in order to achieve integer decimation:

(97) 1. Prune differential sequence in time (before DFT), i.e. select N.sub.DFT=N.

(98) 2. Zero padding in time (before DFT), i.e. N.sub.DFT>N, e.g. N.sub.DFT=2N

(99) An illustration of the autocorrelation of pruned differential sequences is given in FIG. 14, and for zero padding in FIG. 15.

(100) Frequency offset is discussed in the following.

(101) The detection of synchronization signals is independent of frequency offset with the use of differential encoding and decoding, a related illustration is provided in FIG. 16. The actual frequency offset is estimated in a subsequent step using a symbol sequence ({tilde over (r)}.sub.a(n) or {tilde over (r)}.sub.b(n)) in a non-differentially-decoded form, using the previously detected timing alignment value.

(102) Alternative multiplexing of Golay-encoded sequences is described in the following. In FIG. 10, the two constituent (or component) sequences are multiplexed side-by-side in frequency by virtue of directly stacking the DFT outputs for the two sequences. This effectively yields a signal that is a sum of two single-carrier signals.

(103) In some alternative variants, the DFT outputs may be interleaved in frequency according to a regular or pseudorandom pattern, e.g. to increase robustness to fast fading in dispersive channels. The receiver in FIG. 12 is then adapted by changing the mapping of FFT outputs to DFT inputs accordingly.

(104) Separate time and frequency allocations are discussed in the following.

(105) In yet another variant, the sequences are separated both in time and frequency. Then the single-carrier properties could be preserved. A drawback is that sequences would need to be shorter (given same total amount of resources), so that fewer orthogonal sequences are possible. However, if only a few different sequences are needed in some scenario, this solution could be a reasonable trade-off to reduce PAPR and thereby increase coverage.

(106) Generally, there may be considered a synchronization signal constructed by differential encoding of two complementary sequences, which are mapped to different frequency intervals.

(107) FIG. 17 shows an exemplary radio node 100, which may be implemented as a network node or user equipment. Radio node 100 comprises processing circuitry (which may also be referred to as control circuitry) 120, which may comprise a controller connected to a memory. Any module, e.g. receiving module and/or transmitting module and/or configuring module of the radio node may be implemented in and/or executable by the processing circuitry 120. The processing circuitry is connected to control radio circuitry 122 of the radio node 100, which provides receiver and transmitter and/or transceiver functionality (e.g., comprising one or more transmitter and/or receiver and/or transceiver). An antenna circuitry 124 may be connected or connectable to radio circuitry 122 for signal reception or transmittance and/or amplification. The radio node 100 may be adapted to carry out any of the methods for operating a radio node disclosed herein; in particular, it may comprise corresponding circuitry, e.g. processing circuitry. The antenna circuitry may be connected to and/or comprise an antenna array.

(108) FIG. 18 shows an exemplary method for operating a radio node. The method comprises an action RS10 of processing reference signaling based on a coding, the coding being based on a Golay sequence.

(109) FIG. 19 shows an exemplary radio node. The radio node comprises a processing module RM10 for performing action RS10.

(110) Receiving or transmitting on a cell or carrier may refer to receiving or transmitting utilizing a frequency (band) or spectrum associated to the cell or carrier. A cell may generally comprise and/or be defined by or for one or more carriers, in particular at least one carrier for UL communication/transmission (called UL carrier) and at least one carrier for DL communication/transmission (called DL carrier). It may be considered that a cell comprises different numbers of UL carriers and DL carriers. Alternatively or additionally, a cell may comprise at least one carrier for UL communication/transmission and DL communication/transmission, e.g., in TDD-based approaches.

(111) A cell may be generally a communication cell, e.g., of a cellular or mobile or wireless communication network, provided by a node like a network node. A cell may be provided in a RAN. A serving cell may be a cell on or via which a network node (the node providing or associated to the cell, e.g., base station or eNodeB) transmits and/or may transmit data (which may be data other than broadcast data) to a user equipment, in particular control and/or user or payload data, and/or via or on which a user equipment transmits and/or may transmit data to the node; a serving cell may be a cell for or on which the user equipment is configured and/or to which it is synchronized and/or has performed an access procedure, e.g., a random access procedure, and/or in relation to which it is in a RRC_connected or RRC_idle state, e.g., in case the node and/or user equipment and/or network follow the LTE-standard. One or more carriers (e.g., uplink and/or downlink carrier/s and/or a carrier for both uplink and downlink) may be associated to a cell.

(112) There is disclosed a carrier (or storage) medium arrangement carrying and/or storing at least any one of the program products described herein and/or code executable by processing and/or control circuitry, the code causing the processing and/or control circuitry to perform and/or control at least any one of the methods described herein. A carrier medium arrangement may comprise one or more carrier media. Generally, a carrier medium may be accessible and/or readable and/or receivable by processing circuitry. Storing data and/or a program product and/or code may be seen as part of carrying data and/or a program product and/or code. A carrier medium generally may comprise a guiding/transporting medium and/or a storage medium. A guiding/transporting medium may be adapted to carry and/or carry and/or store signals, in particular electromagnetic signals and/or electrical signals and/or magnetic signals and/or optical signals. A carrier medium, in particular a guiding/transporting medium, may be adapted to guide such signals to carry them. A carrier medium, in particular a guiding/transporting medium, may comprise the electromagnetic field, e.g. radio waves or microwaves, and/or optically transmissive material, e.g. glass fiber, and/or cable. A storage medium may comprise at least one of a memory, which may be volatile or non-volatile, a buffer, a cache, an optical disc, magnetic memory, flash memory, etc. Code may generally comprise instructions.

(113) An uplink direction may refer to a data transfer direction from a terminal to a network node, e.g., base station and/or relay station. A downlink direction may refer to a data transfer direction from a network node, e.g., base station and/or relay node, to a terminal. UL and DL may be associated to different frequency resources, e.g., carriers and/or spectral bands. A cell may comprise at least one uplink carrier and at least one downlink carrier, which may have different frequency bands. A network node, e.g., a base station or eNodeB, may be adapted to provide and/or define and/or control one or more cells, e.g., a PCell and/or a LA cell. Cellular DL operation and/or communication of a wireless device or UE may refer to receiving transmissions in DL, in particular in cellular operation and/or from a radio node/network node/gNB/base station. Cellular UL operation of a wireless device or UE may refer to UL transmissions, in particular in cellular operation, e.g. transmitting to a network or radio node/network node/gNB/base station.

(114) Configuring (e.g., with or for a configuration) a device like a UE or terminal or radio node or network node may comprise bringing the device into a state in accordance with the configuration. A device may generally configure itself, e.g. by adapting a configuration. Configuring a terminal, e.g. by a network node, may comprise transmitting a configuration or configuration data indicating a configuration to the terminal, and/or instructing the terminal, e.g. via transmission of configuration data, to adapt the configuration configured. Configuration data may for example be represented by broadcast and/or multicast and/or unicast data, and/or comprise downlink control information, e.g. DCI according to 3GPP standardization. Scheduling may comprise allocating resource/s for uplink and/or downlink transmissions, and/or transmitting configuration or scheduling data indicative thereof.

(115) In this disclosure, for purposes of explanation and not limitation, specific details are set forth (such as particular network functions, processes and signaling steps) in order to provide a thorough understanding of the technique presented herein. It will be apparent to one skilled in the art that the present concepts and aspects may be practiced in other variants and variants that depart from these specific details. For example, the concepts and variants are partially described in the context of Long Term Evolution (LTE) or LTE-Advanced (LTE-A) or LTE Evolution or NR mobile or wireless or cellular communications technologies; however, this does not rule out the use of the present concepts and aspects in connection with additional or alternative mobile communication technologies such as the Global System for Mobile Communications (GSM). While the following variants will partially be described with respect to certain Technical Specifications (TSs) of the Third Generation Partnership Project (3GPP), it will be appreciated that the present concepts and aspects could also be realized in connection with different Performance Management (PM) specifications.

(116) Moreover, those skilled in the art will appreciate that the services, functions and steps explained herein may be implemented using software functioning in conjunction with a programmed microprocessor, or using an Application Specific Integrated Circuit (ASIC), a Digital Signal Processor (DSP), a Field Programmable Gate Array (FPGA) or general purpose computer. It will also be appreciated that while the variants described herein are elucidated in the context of methods and devices, the concepts and aspects presented herein may also be embodied in a program product as well as in a system comprising control circuitry, e.g. a computer processor and a memory coupled to the processor, wherein the memory is encoded with one or more programs or program products that execute the services, functions and steps disclosed herein.

(117) It is believed that the advantages of the approaches, aspects and variants presented herein will be fully understood from the foregoing description, and it will be apparent that various changes may be made in the form, constructions and arrangement of the exemplary aspects thereof without departing from the scope of the concepts and aspects described herein or without sacrificing all of its advantageous effects. Because the aspects presented herein can be varied in many ways, it will be recognized that any scope of protection should be defined by the scope of the claims that follow without being limited by the description.