Radar system and method for operating a radar system

Abstract

Transmitting-receiving devices, such as within a radar system, can use a clock generator from which various higher-frequency signals are derived. For example, respective transmitting-receiving devices can include high-frequency (HF) generators. The present subject matter concerns a system and a method for providing measurement signals having increased coherence as compared with signals originally transmitted by the transmitting-receiving devices. Such measurement signals can be exchanged for synchronization. Increased coherence can enhance overall system performance, such as to assist in separating returns associated with weaker targets from those associated with stronger targets, or to provide enhanced angular resolution, as illustrative examples.

Claims

1. A radar system comprising: a first (SE1) transmitting-receiving device and a second (SE2) transmitting-receiving device, each having a transmitting antenna, a receiving antenna, and a high frequency (HF) generator; and a common clock generator to feed the HF generators of the transmitting-receiving devices, wherein at least one of the first (SE1) transmitting receiving device, the second (SE2) transmitting-receiving device, or a separate evaluation device comprising a processor is configured to process transmitting and receiving signals of the transmitting-receiving devices (SE1, SE2) to provide measurement signals with increased coherence in comparison with the transmitting and receiving signals of the transmitting-receiving devices (SE1, SE2).

2. The system according to claim 1, wherein the first (SE1) transmitting-receiving device and its HF generator or the second (SE2) transmitting-receiving device and its HF generator is arranged on a printed circuit board, together with the common clock generator.

3. The system according to claim 1, wherein the HF generators are arranged in the vicinity of the transmitting and receiving antennae.

4. The system according to claim 1, comprising a phase locked loop to enhance coherence of the HF signals of respective HF generators.

5. The system according to claim 1, wherein the first (SE1) transmitting-receiving device is configured to generate a first signal (sigTX1) for transmission over a path (SP), the second (SE2) transmitting-receiving device is configured to generate a further first signal (sigTX2) for transmission over the path (SP), in the first (SE1) transmitting-receiving device is configured to form a first comparative signal (sigC12) from the first signal (sigTX1) of the first (SE1) transmitting-receiving device and from the further first signal (sigTX2) received from the second (SE2) transmitting-receiving device via the path (SP); and the second (SE2) transmitting-receiving device is configured to form a further comparative signal (sigC21) from the further first signal of the second (SE2) transmitting-receiving device (sigTX2) and from the first signal (sigTX1) received from the first (SE1) transmitting-receiving device via the path (SP), wherein the second (SE2) transmitting-receive device is configured to transmit the further comparative signal (sigC21) to the first (SE1) transmitting-receiving device.

6. The system according to claim 5, wherein at least one of the first (SE1) transmitting-receiving device, the second (SE2) transmitting-receiving device, or the evaluation device is configured to form another comparative signal from the first comparative signal (sigC12) and the further comparative signal (sigC21).

7. The system according to claim 5, wherein at least one of the first (SE1) transmitting-receiving device, the second (SE2) transmitting-receiving device, or the evaluation device is configured to compensate for deviations of the first and second comparative signals which are caused by systematic deviations in the transmitting-receiving devices (SE2, SE1), and to use at least one complex value from the first of the two comparative signals or from a signal which was derived from this first of the two comparative signals, to adapt at least one complex value of the second of the two comparative signals or a value of the second signal which was derived from this second comparative signal, and thus form an adapted signal (sigCC), wherein the adaptation comprises forming a vectorial sum or a difference of the complex values or forming a sum or a difference of phases of the complex values.

8. The system according to claim 1, wherein the first (SE1) transmitting-receiving device or the second (SE2) transmitting-receiving device comprises two or more transmitting antennae two or more receiving antennae.

9. The system according to claim 1, wherein the transmitting-receiving devices (SE1, SE2) are arranged spatially as a sparse array.

10. The system according to claim 1, wherein the first (SE1) transmitting device and the second (SE2) transmitting-receiving device are configured to transmit simultaneously or in a time-overlapping manner so that transmission signals of the first (SE1) transmitting device and the second (SE2) transmitting-receiving device differ from one another.

11. The system according to claim 1, wherein at least one of the first (SE1) transmitting-receiving device, the second (SE2) transmitting-receiving device, or the evaluation device is configured to: mix or correlate the received signals; and separate the received signals on the basis of a distinguishing feature.

12. The system according to claim 1, wherein at least one of the first (SE1) transmitting-receiving device, the second (SE2) transmitting-receiving device, or the evaluation device is configured to: form complex-valued spectra of indirect reflected signals received from the transmitting-receiving devices (SE1, SE2), wherein a sum or a difference of the complex-valued spectra is formed or a sum or a difference of phase values of the indirect reflected signals received from the transmitting-receiving devices is formed.

13. A method for operating a radar system, the method comprising: transmitting a signal from a first (SE1) transmitting-receiving device and a second (SE2) transmitting-receiving device, each having a transmitting antenna, a receiving antenna, and a high frequency (HF) generator; and using at least one of the first (SE1) transmitting receiving device, the second (SE2) transmitting-receiving device, or a separate evaluation device comprising a processor, processing the transmitted and received signals of the transmitting-receiving devices (SE1, SE2) to provide measurement signals with increased coherence in comparison with the transmitted and received signals of the transmitting-receiving devices (SE1, SE2); wherein a common clock generator is used for feeding the HF generators of the transmitting-receiving devices (SE1, SE2).

14. The method according to claim 13, wherein in the first (SE1) transmitting-receiving device, a first signal (sigTX1) is generated and transmitted over a path (SP), in a second (SE2) transmitting-receiving device, a further first signal (sigTX2) is generated and transmitted over the path (SP), a first comparative signal (sigC12) is formed from the first signal (sigTX1) of the first (SE1) transmitting-receiving device and from the first signal (sigTX2) received from the second (SE2) transmitting-receiving device via the path (SP), and a further comparative signal (sigC21) is formed from the first signal of the second (SE2) transmitting-receiving device (sigTX2) and from such a first signal (sigTX1) received from the first (SE1) transmitting-receiving device via the path (SP), and wherein the further comparative signal (sigC21) is transmitted, from the second transmitting-receiving device (SE2) to the first transmitting-receiving device.

15. The method according to claim 14, wherein at least one of the first (SE1) transmitting-receiving device or the second (SE2) transmitting-receiving device compensates for deviations of the first and second comparative signals which are caused by systematic deviations in the transmitting-receiving devices (SE2, SE1), and using at least one complex value from the first of the two comparative signals or from a signal which was derived from this first comparative signal, to adapt at least one complex value of the second of the two comparative signals or a value of the first signal which was derived from this second comparative signal, and thus form an adapted signal (sigCC), wherein the adaptation comprises forming a vectorial sum or a difference of the complex values or forming a sum or a difference of phases of the complex values.

Description

(1) Exemplary embodiments are explained in detail hereinafter with reference to the figures. In the figures:

(2) FIG. 1 shows two inter-communicating transmitting-receiving units and individual components thereof;

(3) FIG. 2 shows the components from FIG. 1 with illustration of a process sequence;

(4) FIG. 3 shows at the top beat signals of the two transmitting-receiving units with non-correlated noise components before synchronization and at the bottom, a synthetic mixed product with correlated phase noise after synchronization;

(5) FIG. 4 shows spectrograms of all the ramps of the two transmitting-receiving units before synchronization;

(6) FIG. 5 shows a schematic diagram of a conventional radar array;

(7) FIG. 6 shows a schematic diagram of a radar array according to the invention;

(8) FIG. 7 shows a phase-noise diagram; and

(9) FIG. 8 shows a schematic diagram of direct and indirect signal paths.

(10) As can be seen from FIG. 1 two transmitting-receiving units SE1, SE2 communicate with one another via a radio interface. In this case, a first or a second signal sigTX1, sigTX2 is transmitted. The transmitting-receiving units SE1, SE2 each have a signal source 1, a unit for clock matching or comparative signal modification 2 and a transmission comparison unit (SigComp1, SigComp2). The (non-coherent) transmitting-receiving units preferably form transmitting-receiving devices. Thus hereinafter, SE1 can be seen as a first transmitting-receiving device and SE2 as a second transmitting-receiving device.

(11) FIG. 2 additionally shows respectively one unit for phase modification 4. Data exchange takes place between the two units for phase modification 4.

(12) Hereinafter the exact mathematical derivation of the operation mode of the method II is performed. In a first (non-coherent) transmitting-receiving unit (SE1), a first signal (sigTX1) is generated and sent over a path (SP), in particular emitted. In a further, in particular second (non-coherent) transmitting-receiving unit (SE2), a second signal (sigTX2) is generated and sent via the path (SP), in particular emitted. In this case, the emission of the signals takes place as far as possible simultaneously but at least temporally matched to one another so that the two signal forms preferably overlap at least half the transmission time. The signal sources can be completely or partially independent.

(13) As is usual in communications engineering, the transmission signals used (siTX1, sigTX2) can be represented as broken down into an equivalent base band signal (bbTX1) and a carrier signal.

(14) Since the system according to the invention should preferably be used for distance measurement or for imaging, preferably signals with so-called good correlation properties can be used as base band signals. Signals with good correlation properties are, for example, wide-band pulses, noise signals, pseudo-random pulse sequences (PN codes) such as M sequences, Gold codes or Barker codes, Kasami sequences, Huffmann sequences, chirps, linear frequency-modulated signals (FMCW), chirp or FMCW sequences etc. Such signal forms have been known for a long time and in many forms in radar technology and communications technology (in particular in the area of CDMA).

(15) The transmission signal (sigTX1) of the transmitting-receiving unit (SE1) can be represented as follows:
sigTX1(t)=Re{bbTX1(t−T.sub.01).Math.e.sup.j(ω.sup.c1.sup.(t−T.sup.01.sup.)+ϕTX1(t−T.sup.01.sup.))}.

(16) The time offset T01 defines the transmission time of the signal sigTX1;

(17) the phase term ϕTX1(t)+φTX1+ξTX1(t) comprises a constant phase offset and the phase noise of the carrier signal.

(18) The circular frequency ω.sub.c1 characterizes the frequency of the carrier signal of sigTX1.

(19) The transmission signal (sigTX2) of the transmitting-receiving unit (SE2) can be formed in the same way. It holds that:
sigTX2(t)=Re{bbTX2(t−T.sub.02).Math.e.sup.j(ω.sup.c2.sup.(t−T.sup.02.sup.)+ϕTX2(t−T.sup.02.sup.))}

(20) The transmitted signals (sigTX1 and sigTX2) arrive—on the direct path or reflected at objects—at the respectively other transmitting-receiving station and are received then and further processed as receiving signals sigRX12 and sigRX21.

(21) The receiving signal, which is received at the second (non-coherent) transmitting-receiving unit (SE2) corresponds to the transmission signal (sigTX1), wherein however this is changed in amplitude and delayed by the transit time τ.sub.21. To simplify the mathematical representation and without restricting the general disclosure, all the signals should subsequently be represented as complex-value signals. It therefore holds that:
sigRS21(t)=ARX21.Math.bbTX1(t−T.sub.01−τ.sub.21).Math.e.sup.j(ω.sup.c1.sup.(t−T.sup.01.sup.−τ.sup.21.sup.)+ϕTX1(t−T.sup.01.sup.−τ.sup.21.sup.))

(22) If the transmission signal (sigTX1) is transmitted on several (a number of I) different-length transmission paths to the second transmitting-receiving unit (SE2), the receiving signal can be represented as follows as a linear superposition of amplitude-weighted and time-delayed signals as follows:

(23) $\mathrm{sigRX}21\left(t\right)=\underset{i=1}{\overset{I}{.Math.}}\mathrm{sigRX}21i\left(t\right)$

(24) where
sigRX21i(i)=ARX21i.Math.bbTX1(t−T.sub.01−τ.sub.21).Math.e.sup.j(ω.sup.c1.sup.(t−T.sup.01.sup.−τ.sup.221.sup.)+ϕTX1(t−T.sup.01.sup.−τ.sup.222.sup.))

(25) For the signal transmitted from the second transmitting-receiving unit (SE2) to the first transmitting-receiving unit (SE1), it holds accordingly that

(26) $\mathrm{sigRX}12\left(t\right)=\mathrm{ARX}12.Math.\mathrm{bbTX}2\left(t-T_{02}-{\tau }_{12}\right).Math.e^{j\left({\omega }_{c2}\left(t-T_{02}-{\tau }_{12}\right)+\varphi \mathrm{TX}2\left(t-T_{02}-{\tau }_{12}\right)\right)}$$\mathrm{or}$$\mathrm{sigRX}12\left(t\right)=\underset{i=1}{\overset{I}{.Math.}}\mathrm{sigRX}12i\left(t\right)$

(27) where
sigRX12i(i)=ARX12i.Math.bbTX2(t−T.sub.02−τ.sub.121).Math.e.sup.j(ω.sup.c2.sup.(t−T.sup.02.sup.−τ.sup.02.sup.−τ.sup.11.sup.)+ϕTX2(t−T.sup.01.sup.−τ.sup.12.sup.))

(28) The transmitting-receiving units (SE1, SE2) are designed so that they comprise signal comparison units SigComp1, SigComp2 in which the respective receiving signals of a transmitting-receiving unit is calculated with its transmitting signal—i.e. in SE1 the signal sigRX12 with the signal sigTX1 and in SE2 the signal sigRX21 with the signal sigTX2. The signal comparison units SigComp1, SigComp2 are executed as a mixer Mix in the exemplary embodiment. That is, here in SE1 the signal sigRX12 is mixed with the signal sigTX1 and in SE2 the signal sigRX21 is mixed with the signal sigTX2. It is generally known as such that a mixing process can be expressed system-theoretically as multiplication or a down-mixing in the case of two complex sine signals as multiplication of the signals with the complex conjugate (*=sign for conjugation) of the other signal. It therefore holds that:

(29) $\begin{array}{c}\mathrm{sigC}12=\mathrm{sigRX}{12}^{*}.Math.\mathrm{sigTX}1\\ =\mathrm{ARX}12.Math.\mathrm{bbTX}{2}^{*}\left(t-T_{02}-{\tau }_{12}\right).Math.e^{-j\left({\omega }_{c2}\left(t-T_{02}-{\tau }_{12}\right)+\varphi \mathrm{TX}2\left(t-T_{02}-{\tau }_{12}\right)\right)}.Math.\\ \mathrm{bbTX}1\left(t-T_{01}\right).Math.e^{j\left({\omega }_{c1}\left(t-T_{01}\right)+\varphi \mathrm{TX}1\left(t-T_{01}\right)\right)}\\ =\mathrm{ARX}12.Math.\mathrm{bbTX}{2}^{*}\left(t-T_{02}-{\tau }_{12}\right).Math.\mathrm{bbTX}1\left(t-T_{01}\right).Math.\\ e^{j\left({\omega }_{c1}\left(t-T_{01}\right)+\varphi \mathrm{TX}1\left(t-T_{01}\right)-{\omega }_{c2}\left(t-T_{02}-{\tau }_{12}\right)-\varphi \mathrm{TX}2\left(t-T_{02}-{\tau }_{12}\right)\right)}\end{array}$

(30) Another advantageous case consists in forming a comparative signal in that SE1 does not mix the signal sigRX12 with the signal sigTX1 but only with its carrier. That is:

(31) $\mathrm{sigC}12=\mathrm{sigRX}{12}^{*}.Math.e^{j\left({\omega }_{c1}\left(t-T_{01}\right)+\varphi \mathrm{TX}1\left(t-T_{01}\right)\right)}=\mathrm{ARX}12.Math.\mathrm{bbTX}{2}^{*}\left(t-T_{02}-{\tau }_{12}\right).Math.e^{j\left({\omega }_{c1}\left(t-T_{01}\right)+\varphi \mathrm{TX}1\left(t-T_{01}\right)-{\omega }_{c2}\left(t-T_{02}-{\tau }_{12}\right)-\varphi \mathrm{TX}2\left(t-T_{02}-{\tau }_{12}\right)\right)}$

(32) For the signals in the SE2 it holds accordingly:

(33) $\begin{array}{c}\mathrm{sigC}21=\mathrm{sigRX}{21}^{*}.Math.\mathrm{sigTX}2\\ =\mathrm{ARX}21.Math.\mathrm{bbTX}{1}^{*}\left(t-T_{01}-{\tau }_{21}\right).Math.e^{-j\left({\omega }_{c1}\left(t-T_{01}-{\tau }_{21}\right)+\varphi \mathrm{TX}1\left(t-T_{01}-{\tau }_{21}\right)\right)}.Math.\\ \mathrm{bbTX}2\left(t-T_{02}\right).Math.e^{j\left({\omega }_{c2}\left(t-T_{02}\right)+\varphi \mathrm{TX}2\left(t-T_{02}\right)\right)}\\ =\mathrm{ARX}21.Math.\mathrm{bbTX}{1}^{*}\left(t-T_{01}-{\tau }_{21}\right).Math.\mathrm{bbTX}2\left(t-T_{02}\right).Math.\\ e^{j\left({\omega }_{c2}\left(t-T_{02}\right)+\varphi \mathrm{TX}2\left(t-T_{02}\right)-{\omega }_{c1}\left(t-T_{01}-{\tau }_{21}\right)-\varphi \mathrm{TX}1\left(t-T_{01}-{\tau }_{21}\right)\right)}\end{array}$

(34) or in the alternative embodiment:

(35) $\mathrm{sigC}21=\mathrm{sigRX}{21}^{*}.Math.e^{j\left({\omega }_{c2}\left(t-T_{02}\right)+\varphi \mathrm{TX}2\left(t-T_{02}\right)\right)}=\mathrm{ARX}21.Math.\mathrm{bbTX}{1}^{*}\left(t-T_{01}-{\tau }_{21}\right).Math.e^{j\left({\omega }_{c2}\left(t-T_{02}\right)+\varphi \mathrm{TX}2\left(t-T_{01}-{\tau }_{21}\right)-{\omega }_{c1}\left(t-T_{01}-{\tau }_{21}\right)-\varphi \mathrm{TX}1\left(t-T_{01}-{\tau }_{21}\right)\right)}$

(36) It is now assumed that means are provided in SE which ensure that the following conditions are satisfied:
T.sub.01=T.sub.02=T.sub.0 and ω.sub.c2=ω.sub.c1=ω.sub.c

(37) How these means can preferably be configured has already been explained above or will be explained further below in an exemplary embodiment. Under these boundary conditions, it follows:

(38) $\mathrm{sigC}12=\mathrm{ARX}12.Math.\mathrm{bbTX}{2}^{*}\left(t-T_0-{\tau }_{12}\right).Math.\mathrm{bbTX}1\left(t-T_0\right).Math.e^{j\left({\omega }_c\left(t-T_0\right)+\varphi \mathrm{TX}1\left(t-T_0\right)-{\omega }_c\left(t-T_0-{\tau }_{12}\right)-\varphi \mathrm{TX}2\left(t-T_0-{\tau }_{12}\right)\right)}=\mathrm{ARX}12.Math.\mathrm{bbTX}{2}^{*}\left(t-T_0-{\tau }_{12}\right).Math.\mathrm{bbTX}1\left(t-T_0\right).Math.e^{j\left({\omega }_c{\tau }_{12}+\varphi \mathrm{TX}1\left(t-T_0\right)-\varphi \mathrm{TX}2\left(t-T_0-{\tau }_{12}\right)\right)}$$\mathrm{sigC}21=\mathrm{ARX}21.Math.\mathrm{bbTX}{1}^{*}\left(t-T_0-{\tau }_{21}\right).Math.\mathrm{bbTX}2\left(t-T_0\right).Math.e^{j\left({\omega }_c\left(t-T_0\right)+\varphi \mathrm{TX}1\left(t-T_0\right)-{\omega }_c\left(t-T_0-{\tau }_{21}\right)-\varphi \mathrm{TX}1\left(t-T_0-{\tau }_{21}\right)\right)}=\mathrm{ARX}21.Math.\mathrm{bbTX}{1}^{*}\left(t-T_0-{\tau }_{21}\right).Math.\mathrm{bbTX}2\left(t-T_0\right).Math.e^{j\left({\omega }_c{\tau }_{21}+\varphi \mathrm{TX}1\left(t-T_0\right)-\varphi \mathrm{TX}2\left(t-T_0-{\tau }_{21}\right)\right)}$

(39) If a reciprocal transmission channel is assumed, it further holds that:
τ.sub.21=τ.sub.12=τ

(40) In the next step with a data communication it is ensured that both comparative signals are transmitted to a common evaluation unit and both are present there for the evaluation. The common evaluation unit can be SE1, SE2 or another evaluation unit.

(41) Now in a further processing step, the phases of the two comparative signals are added. If only the carrier phases with the phase noise component are considered here, since unknown phase contributions are present in this component and the two carrier phase terms are added, this gives:

(42) $\Delta \varphi =\left({\omega }_c\tau +\varphi \mathrm{TX}1\left(t-T_0\right)-\varphi \mathrm{TX}2\left(t-T_0-\tau \right)\right)+\left({\omega }_c{\tau }_{21}+\varphi \mathrm{TX}2\left(t-T_0\right)-\varphi \mathrm{TX}1\left(t-T_0-{\tau }_{21}\right)\right)=2{\omega }_c\tau +\varphi \mathrm{TX}1\left(t-T_0\right)-\varphi \mathrm{TX}1\left(t-T_0-\tau \right)+\varphi \mathrm{TX}2\left(t-T_0\right)-\varphi \mathrm{TX}2\left(t-T_0-\tau \right)$

(43) If it is now borne in mind that as a result of the high propagation velocity of electromagnetic waves, the transit time τ is usually very short and that the definitive phase noise components in an oscillator according to the known relationships of oscillator phase noise typically decrease substantially with increasing distance from the carrier, and ϕTX1 or ϕTX2 consequently exhibit a defined low-pass behaviour and specifically a low pass behaviour with a limiting frequency, which is usually significantly less than 1/τ, it follows that:
δϕ1(1)=ϕTX1(t−T.sub.0)−ϕTX1(t−T.sub.0−τ) where δϕ1(t)<<ϕTX1(t)
δϕ2=ϕTX2(t−T.sub.0)−ϕTX2(t−T.sub.0−τ) where δϕ2(t)<<ϕTX2(t)

(44) The proposed processing whereby in one of the comparative signals the phase of the respectively other comparative signal is added, therefore has the result that the perturbations due to phase noise are reduced quite considerably. This phase noise reduction results in a better detectability of targets, a larger measurement range and an improved measurement accuracy.

(45) Depending on the selected mixer topology, whether for example a same-position or an inverted position mixer, is used, it is possible that the phase terms presented above have different signs. Depending on the sign, the preferred linking of the phase terms, is not necessarily an addition but possibly also a subtraction. It is crucial that the link results in a reduction of the phase noise terms and the transit-time-dependent phase term, i.e. an expression comprising the term ω.sub.cT is maintained. It is further generally known that for the case where the phase values are represented by complex numbers, the complex numbers are multiplied by one another, divided or multiplied with the complex conjugate of the respectively other number in order to form the sum or the difference of the phases.

(46) A possible preferred variant for reducing the phase noise components shall be described in the following. In many cases, it is favourable that in the first and second (non-coherent) transmitting-receiving unit (SE1, SE1) base band signals of the same type are generated, i.e. it holds that:
bbTX1=bbTX2=bbTX.

(47) In an at least approximately reciprocal radio channel, it should further be assumed that:
ARX12=ARX21=ARX.

(48) Under these boundary conditions, it follows that:
sigC12=ARX.Math.bbTX*(t−T.sub.0−τ.sub.12).Math.bbTX(t−T.sub.0).Math.e.sup.j(ω.sup.c.sup.τ+ϕTX1(t−T.sup.0.sup.)−ϕTX2(t−T.sup.0.sup.−τ))
sigC21=ARX.Math.bbTX*(t−T.sub.0−τ.sub.21).Math.bbTX(t−T.sub.0).Math.e.sup.j(ω.sup.c.sup.τ+ϕTX2(t−T.sup.0.sup.)−ϕTX1(t−T.sup.0.sup.−τ))

(49) As can easily be identified, the two signals are identical apart from their phase terms.

(50) Easily distinguishable amplitudes of the signals sigC12 and sigC21 can occur, however, despite a reciprocal radio channel as a result of different properties of the electronic components such as mixers or amplifiers etc. If the amplitudes of the signals sigC12 and sigC21 are different, in the preferred variant described here the signals must initially be normalized to the same amplitude.

(51) During the process for forming the signals sigC12 and sigC21, additional systematic phase offsets can also occur. If these phase offsets of the signals sigC12 and sigC21 are different, in the preferred variant described here, these phase offsets must be initially compensated.

(52) For a certain time t, the signals sigC12 and sigC21 can be interpreted as complex indicators. As a result of a complex addition of the indicators, the vector components of the phase terms with different signs cancel out in the same manner as was described above in the addition of the phase terms. Consequently, as a possible preferred variant for reducing the phase noise components it is proposed to add the complex signals sigC12 and sigC21, i.e., to form a signal as follows:
sigCC=sigC12+sigC21

(53) The signal sigCC then has a significantly lower phase noise than the signal sigC12 or sigC21 and the signal sigCC is then used further for the purpose of distance measurement, angular measurement or for imaging. However, it is important that before the addition of the signals, the previously described systematic deviations of amplitude and phases which cause different carrier frequencies and transmission times, were compensated.

(54) Naturally not all values of sigC12 and sigC21 and also certainly not the signals sigC12 and sigC21 themselves must be added. However, at least one complex value from a first of the two comparative signals and from a signal that was derived from this first comparative signal should be used to adapt at least one complex value of the second of the two comparative signals or a value of a signal that was derived from this second comparative signal and thus form at least one value of a signal (sigCC), wherein the adaptation is made in such a manner that by means of a mathematical operation the vectorial sum or the difference of at least two complex values derived from sigC12 and sigC21 is formed or the sum or the difference of the phases of these complex values is formed.

(55) It should be pointed out here that the proposed mixing processes only form a possible embodiment and that the compensations of the phase noise components could also be achieved by alternative methods. Thus, possibly all the high-frequency signals could already be digitized before mixing, i.e. scanned with an analog-to-digital converter and all further operations could be accomplished computationally or digitally, for example, in a processor or FPGA (field-programmable gate array).

(56) In principle, the transmitted signals sigTX1 and sigTX2 can be modulated. Preferably in this case (before the mathematical operation), the spectra of the comparative signals are normalized to the highest value.

(57) A special embodiment of method II with FMCW signals and a plurality of successive N ramps is described hereinafter. In this case, the SE transmit several N signals with linearly increasing or decreasing frequency, hereinafter designated as frequency ramps. The comparative signals are then generated from the received signals in the SE and buffered for further processing. For example, ascending and descending ramps are used since a correct-sign determination of the relative speed is herewith possible.

(58) Firstly individual spectrograms of the beat signals sigC12 and sigC21 are created for each receiving channel for each ramp. These spectrograms are placed side by side in the amplitude diagram without phase information for all N consecutive ramps. This is shown in FIG. 4 for the ascending ramps, in which two maxima appear since no IQ mixing was carried out but a real-value scanning signal is present. When used in primary radars, for this step the at least one reflector in the detection range must be identified in advance and represented as described previously.

(59) Now the frequency band in which the beat signal is to be expected (ensured by a coarse pre-synchronization) is roughly cut out. Then in each case, the spectrogram of the first N/2 ramps is correlated with that of the second N/2 ramps along the frequency axis (step 1). The maximum thereby found reproduces the relative time drift of the two SEs (here a linear function can be assumed). When receiving the signals via one or more reflections, for example, the identification of the targets can be made by means of the opposite drift on both sides.

(60) Alternatively, a determination of the frequency offset can also be made in particular in primary radars via a common bus system whereby the systems exchange their measurement signals or more extensive synchronization signals via the cable of a bus system. The bus system is in this case in particular a CAN, FlexRay, Most, Gigabyte Ethernet system, USB, Firewire or TTP system.

(61) Then, all the ramps in the spectrogram are corrected by this drift, by multiplying, for example, with a complex correction signal having an opposite frequency offset in the unit for clock matching or comparative signal modification 2. The spectrograms of the various ramps thus obtained are (incoherently) added and as a result of the superposition, the maximum is sought which corresponds to the time offset (offset error). In primary radar, the identification of the related peaks made in the previous step can be used for the selection of the peaks.

(62) Alternatively, a determination of the time offset can also be made via a common bus system, in particular either by transferring measurement data or suitable correlation sequences.

(63) The parameters of the relative time offset determined in this way and relative time drift (=current frequency offset) are determined by means of the complete sequence of N ramps. This result contains a large part of the clock deviation. In addition, it is now known for each ramp and each station at which point in the spectrogram the energy of the incident signal is to be expected in each case.

(64) The originally recorded local mixing signals sigC12 and sigC21 are now initially shifted by integer values Tint (representation of the time offset between the two stations as ΔT=|T01-T02|=Tint+Tfrac) in order to obtain a uniform time basis. Due to the common time basis the phase noise is more strongly correlated. The remaining small time error Tfrac can now be compensated possibly by using a fractional delay filter. The signals thus shifted are now corrected by the different ramp steepness which occurs as a result of the frequency offset Δω=ω1−ω2 of the two local oscillators, by convolving or spectrally multiplying by a normalized correction signal, which corresponds to the frequency behaviour in the opposite direction.

(65) In these resharpened mixing signals, a peak is sought in each case after an FFT of the beat signal for the channel pulse response. In the case of secondary radar, preferably the strongest peak or alternatively the first peak is taken, in primary radar a peak obtained equally on both sides must be selected. For each ramp at both stations a maximum is thus obtained with the estimated spacing with the appurtenant phase position. These values fundamentally agree for the measurement on the forward and return path in the case of a reciprocal channel. The remaining deviations can be attributed to remaining frequency and phase differences between both signal sources 1 of the SE, for example, of the oscillators whose phase noise forms the basic cause. The precise frequency difference can now be determined absolutely and therefore corrected (the phase difference can be determined up to 180° ambiguity (in the case of IQ mixers) 360°. This ambiguity is eliminated by a restriction of the phase profile to +/−90° from ramp to ramp, which is also designated as unwrapping. After this precise correction of the remaining phase error, the synthetic signals of both stations now scarcely differ.

(66) After this pre-processing, the characteristic systematic errors of the radar system are completely corrected, which is why the phase shift of the two beat signals only differs by a small amount. At this point, on the one hand, a precise synchronization of the time and frequency base is achieved and on the other hand, the phase noise can be considered as an additive contribution and can be eliminated by linear combination. This is accomplished, for example, by means of 2D Fourier transformation of all N ramps at both SEs, whereupon finally the beat signals normalized in the amplitude are added. By incorporating the system parameters (scanning rate, ramp steepness, carrier frequency, . . . ), the maximum of the result of this linear combination forms the estimated value for this distance and speed.

(67) FIG. 5 shows as an example a conventional arrangement for a radar system with transmitting-receiving devices SE1, SE2, which each have at least two transmitting and receiving antennae as well as having an HF generator for the HF signal and a distributing device for distributing the HF signal to the transmitting-receiving devices SE1, SE2 and a clock generator for a system clock.

(68) FIG. 6 shows an arrangement of a radar system according to the invention with transmitting-receiving devices SE1, SE2, which each have at least two transmitting-receiving antennae as well as an HF generator and having a (common) clock generator for a system clock.

(69) FIG. 7 shows a phase noise diagram for IF signals which originate from the down-mixing of signals from different signal generators such as can be obtained with a radar system according to FIG. 6 and the use of a suitable phase-locked loop but without a method for the subsequent production of coherence.

(70) FIG. 8 shows a signal propagation for the radar system according to the invention. The signals received from each receiving path contain the dedicated signals reflected by the surroundings and the signals of the second transmission path reflected by the surroundings.

(71) The received signals are down-mixed or correlated with the local high-frequency signal. This results in a low-frequency signal S1.sub.beat which has both components from the direct reflection path (dedicated transmission signal) and from the indirect reflection path (external transmission signal). This is followed by a separation of the signal by means of the frequency difference or another type of modulation. The signals of the dedicated reflection path are then processed as normal radar signals, the signals of the indirect path are further processed as follows:

(72) Optionally, the indirect signals from both receiving paths are corrected by some frequency offset. In addition to the frequency offset, which was possibly introduced to distinguish/code the two signals, frequency offsets can also be perturbing system-induced frequency offsets. However, correction of the latter is usually not necessary as a result of the common clock source. Timing offsets, for example, due to (slightly) different times of the HF generators (frequency generators) are also corrected, for example, in the post processing by (for example) applying DE 101 57 931. Two spectra (spectrum I and spectrum II) can then be formed by both signals and preferably normalized in relation to their amplitudes. By means of a mathematical operation, the sum or the difference of the complex spectra or of signals which were derived from the spectra, are formed or the sum or the difference of phase values of the aforesaid signals is formed.

(73) A preferred variant of the evaluation can proceed as follows: one of the two calculated spectra (spectrum I) is preferably converted into a complex conjugate spectrum (spectrum 1C). This spectrum (spectrum 1C) and the non-complex-conjugate converted spectrum (spectrum 1) are added or subtracted or multiplied or divided in a mathematical operation. The resulting spectrum can then be processed as a normal radar spectrum.

(74) It is advantageous that due to the mathematical combination of the two signals (spectrum 1 and spectrum 2), the additional noise which was produced by the use of the separate HF signal sources can be very effectively suppressed.

(75) It is particularly preferred if the transmitting-receiving paths are arranged as sparse arrays. By this means, auxiliary maximum and complete aperture can be optimized so that accuracy and target separation can be significantly improved. Likewise preferred is the use of chips which comprise already-integrated antennae, either in the chip directly or in a corresponding package.

(76) Aspects and embodiments of method I or configuration I are described hereinafter. The reference numbers relate to the figures from DE 10 2014 104 273 A1. The transmitting-receiving units can be part of (optionally at least partially coherent) transmitting-receiving devices or can form these:

(77) 1st aspect: Method in a radar system, in which in a first (non-coherent) transmitting-receiving unit (SE1), a first signal (sigTX1) is generated and transmitted over a path (SP), in particular emitted in a further, in particular second (non-coherent) transmitting-receiving device (SE2), a first signal (sigTX2) is generated and transmitted over the path (SP), in particular emitted, in first transmitting-receiving unit (SE1), a comparative signal (sigC12) is formed from the first signal (sigTX1) and from such a first signal (sigTX2) received from the second transmitting-receiving unit (SE2) via the path (SP) and in the further transmitting-receiving device (SE2), a further comparative signal (sigC21) is formed from the first signal and from such a first signal (sigTX1) received from the first transmitting-receiving unit (SE1) via the path (SP), wherein the further comparative signal (sigC21) is preferably transmitted, in particular communicated, from the further transmitting-receiving unit (SE2) to the first transmitting-receiving unit (SE1).

(78) 2nd aspect: method according to the first aspect, in which a comparative-comparative signal (sigC21; sigC12) is formed from this comparative signal (sigC21) and the further comparative signal (sigC21).

(79) 3rd aspect: method according to the second aspect, in which the comparative-comparative signal (sigC21; sigC12) by processing together, in particular by complex-conjugate multiplying the two comparative signals (sigC21; sigC12), corresponds to a comparative signal generated with a coherent radar system.

(80) 4th aspect: method according to a preceding aspect, in which at least one of the comparative signal (sigC12), the further comparative signal (sigC21) or the comparative-comparative signal (sigC21; sigC12) is formed by at least one of mixing or correlation.

(81) 5th aspect: method according to a preceding aspect, in which at least one such further comparative signal (sigC21; sigC12) is transmitted between the transmitting-receiving units (SE2; SE1) as at least one of data, a data-containing signal or a signal containing data reconstructably.

(82) 6th aspect: method according to a preceding aspect, in which at least one of the first signals (sigTX1, sigTX2) is sent as a transmission signal via the path (SP) formed as an air interface.

(83) 7th aspect: method according to a preceding aspect, in which times for sending the first signals (sigTX1, sigTX2) are coordinated in such a manner that the first signals (sigTX1, sigTX2) overlap at least partially in time.

(84) 8th aspect: method according to a preceding aspect, in which from at least one comparative-comparative signal (sigC21; sigC12), a signal transit time (T12) required by such a first signal (sigTX1, sigTX2) for the path between the transmitting-receiving units (SE1, SE2) is determined by analysing at least one phase or a phase value (φ12, φ13, . . . , φ1N, φ22, φ23, φ24, . . . , φ2N, . . . , φN−N) of a frequency, an amplitude profile or a phase profile of the comparative-comparative signal (sigCC12).

(85) 9th aspect: method according to a preceding aspect, in which at least one of the first signals (sigTX1, sigTX2) is generated and sent as an FMCW or OFDM-modulated signal.

(86) 10th aspect: method according to a preceding aspect, in which at least one of the first signals (sigTX1, sigTX2) is generated and sent as a multi-ramp signal.

(87) 11th aspect: method according to a preceding aspect, in which a plurality of comparative-comparative signals (sigC21; sigC12) which are measured temporally successively with at least two transmitting-receiving units (SE1, SE2) of which at least one of the transmitting-receiving units (SE1, SE2) moves and by means of a synthetic aperture method at least one of a distance, a position, a speed or the presence of one of the transmitting-receiving units (SE1, SE2) or the presence of one such transmitting-receiving units (SE1, SE2) or at least one of a distance, a position, a speed relative to an object (O) or the presence of an object (O) is determined.

(88) 12th aspect: radar system in which at least one first (non-coherent) transmitting-receiving unit (SE1) is configured to generate a first signal (sigTX1) and send it, in particular emit it via a path (SP), at least one further, in particular second (non-coherent) transmitting-receiving unit (SE2) is configured to generate a first signal (sigTX2) and send it, in particular emit it via the path (SP), the first transmitting-receiving unit (SE1) is configured to form a comparative signal (sigC12) from the first signal (sigTX1) thereof and from such a first signal (sigTX2) received from the further transmitting-receiving unit (SE2) via the path (SP), the further transmitting-receiving unit (SE2) is configured to form a further comparative signal (sigC21) from the first signal (sigTX2) thereof and from such a first signal (sigTX1) received from the transmitting-receiving unit (SE1) via the path (SP) and the further comparative signal (sigC21) is transmitted, in particular communicated from the further transmitting-receiving unit (SE2) to the first transmitting-receiving unit (SE1).

(89) 13th aspect: radar system according to aspect 12, in which a comparative-comparative signal (sigC21; sigC12) is formed from this comparative signal (sigC21) and the further comparative signal (sigC21).

(90) 14th aspect: radar system according to aspect 12 or 13, with three or more spatially spaced-apart transmitting-receiving units (SE1, SE2, SE3, . . . , SE-N) in which from two or more comparative-comparative signals (sigCC12, sigCC12, sigCC13, sigCC22, . . . , sigCC32) which are measured with more that two pairs of respectively two of the spatially spaced-apart transmitting-receiving units (SE1, SE2; SE-N, SE2), a distance, a position, a speed or the presence of one of the transmitting-receiving units (SE1, SE2) or the presence of one such transmitting-receiving unit (SE2, SE1) or at least one of a distance, a position, a speed relative to an object (O) or the presence of an object (O) is determined.

(91) 15th aspect: radar system according to one of aspects 12 to 14, in which the first transmitting-receiving unit (SE1) and at least one such further transmitting-receiving unit (SE2) and/or an evaluation device (P) are configured to carry out a method according to one of the preceding claims.

(92) 16th aspect: apparatus of a radar system, in particular for carrying out a method according to one of aspects 1 to 11 and/or in a radar system according to one of aspects 12 to 15, wherein the apparatus is configured as a first (non-coherent) transmitting-receiving unit (SE1), in particular a first (non-coherent) transmitting-receiving unit (SE1) and has a signal generator and at least one antenna (TA1; RA1) configured to generate a first signal (sigTX1) to send, in particular emit, over a path (SP), has an arrangement configured to form a comparative signal (sigC12) from the first signal (sigTX1) and from such a first signal (sigTX2) received from the further transmitting-receiving unit (SE2) via the path (SP) and has at least one of an interface (CommTX) which is configured to transmit, in particular communicate the comparative signal (sigC12) to the further transmitting-receiving unit (SE2) or has an interface (CommRX) which is configured to obtain such a further comparative signal (sigC21) generated by the further transmitting-receiving unit (SE2) by means of transmitting, in particular communicating, in the first transmitting-receiving unit (SE1).

(93) 17th aspect: apparatus according to aspect 16 with a further comparison unit (sigComp12, which forms a comparative-comparative signal (sigCC12) from the comparative signal (sigC21) formed in the same transmitting-receiving unit (SE1) and the comparative signal (sigC21) transmitted to this transmitting-receiving unit (SE1).

(94) 18th aspect: apparatus according to aspect 16 or 17, in which the at least one interface (CommTX, CommRX) is a data interface.

(95) 19th aspect: apparatus according to one of aspects 16 to 18 in which between the arrangement which outputs the comparative signal (sigC12) and the further comparison unit (sigComp12) which forms the comparative-comparative signal (sigCC12), a filter (FLT) is arranged, wherein the filter (FLT) applies the comparative signal (sigC12) to the comparison unit (sigComp12), wherein the filter (FLT) does not apply a further comparative signal (sigC11) formed in the arrangement upstream of the filter (FLT) and suppresses or provides to a connection the comparative signal (sigC11) formed in the upstream arrangement.

(96) 20th aspect: apparatus according to one of aspects 16 to 19 which comprises a plurality of spatially spaced-apart receiving antennae (RA1,1, . . . , RA1,N: RA2,1, . . . , RA2,N) to which respectively one arrangement is assigned, which is configured to form respectively one comparative signal (sigC21,1, sigC21,2, sigC21,3) from the first signal (sigTX2) and from such a first signal (sigTX1) received from such a further transmitting-receiving unit (SE2) via the path (SP).

(97) Aspects and embodiments of method II or configuration II are described hereinafter. The reference numbers relate to FIGS. 1 to 4 of the present application. The transmitting-receiving units can be part of (optionally at least partially coherent) transmitting-receiving devices or can form these:

(98) 1st aspect: Method for reducing perturbations by phase noise in a radar system in which in a first (non-coherent) transmitting-receiving unit (SE1), a first signal (sigTX1) is generated and transmitted over a path (SP), in particular emitted in a further, in particular second (non-coherent) transmitting-receiving device (SE2), a first signal (sigTX2) is generated and transmitted over the path (SP), in particular emitted, the first signals (sigTX1 and sigTX2) are received in the respectively other transmitting-receiving unit by a direct or indirect path and are further processed there as receiving signals (sigRX12 and sigRX21), in first transmitting-receiving unit (SE1), a comparative signal (sigC12) is formed from the first signal (sigTX1) and from such a first signal (sigRTX2) received from the further transmitting-receiving unit (SE2) via the path (SP) and in the further transmitting-receiving device (SE2), a further comparative signal (sigC21) is formed from the first signal (sigTX2) and from such a first signal (sigTX1) received from the first transmitting-receiving unit (SE1) via the path (SP), wherein the further comparative signal (sigC21) is preferably transmitted, in particular communicated, from the further transmitting-receiving unit (SE2) to the first transmitting-receiving unit (SE1), wherein in a first step deviations of the comparative signals (sigC21 and sigC12) caused by systematic deviations in the transmitting-receiving units (SE2, SE1) are compensated, wherein in a second step at least one complex value from one of the two comparative signals or from one signal which was derived from this first comparative signal is used to adapt at least one complex value of the second of the comparative signals or a value of a signal which was derived from this second comparative signal, and thus form an adapted signal (sigCC), wherein the adaptation is accomplished in such a manner that by means of a mathematical operation, the vectorial sum or the difference of the complex values is formed or the sum or the difference of the phases of the complex values is formed.

(99) 2nd aspect: method according to aspect 1, wherein the transmitted signals (sigTX1 and sigTX2) FMCW are modulated.

(100) 3rd aspect: method according to aspect 1 or 2, wherein a clock rate matching, in particular of signal sources of the first signals (sigTX1 and sigTX2) is carried out via a bus system, preferably a communication bus and/or wherein a clock rate matching, in particular of clock rates of signal sources of the first signals (sigTX1 and sigTX2) is carried out via radio waves and/or via a cable connection, in particular during operation as primary radar.

(101) 4th aspect: method according one of the preceding aspects, wherein a synchronization of the (non-coherent) transmitting-receiving units SE1, SE2) in particular a pre-synchronization is carried out by determining a frequency drift via several ramps successively, in particular when using a secondary radar.

(102) 5th aspect: method according one of the preceding aspects, wherein an offset, in particular a time offset and/or a frequency offset is determined via a bus system, preferably during operation as primary radar.

(103) 6th aspect: method according one of the preceding aspects, wherein an/the offset, in particular a/the time offset and/or a/the frequency offset is determined via an evaluation of a position of, in particular, corrected maxima of the spectra of the comparative signals (sigC12 and sigC21).

(104) 7th aspect: method according one of the preceding aspects, wherein the first and/or the further (non-coherent) transmitting-receiving unit comprises at least one evaluation device for carrying out the individual process steps, in particular calculations and evaluations, wherein the respective evaluation device is

(105) possibly a physically independent evaluation device which is connected to the respective transmitting-receiving unit or the other components of the respective transmitting-receiving unit or

(106) possibly is integrated in the first and/or the further (non-coherent) transmitting-receiving unit, for example, in a common housing and/or as a structural component.

(107) 8th aspect: method according one of the preceding aspects, wherein the comparative signals (sigC12 and sigC21) are transmitted to an, in particular, common evaluation unit and are both present there for evaluation, wherein the common evaluation unit is optionally the first (non-coherent) transmitting-receiving unit (SE1) or optionally the second (non-coherent) transmitting-receiving unit (SE2) or optionally another, in particular separate evaluation unit.

(108) 9th aspect: method according one of the preceding aspects, wherein the first signals (TX1 and TX2) are sent at least in a temporally overlapping manner, wherein the further first signal (TX2) of the further (non-coherent) transmitting-receiving unit (SE2) is preferably sent at least during half the signal duration of the first signal (TX1) of the first (non-coherent) transmitting-receiving unit (SE2), further preferably at least approximately simultaneously.

(109) 10th aspect: method according one of the preceding aspects, wherein before the mathematical operation, the spectra of the comparative signals are normalized to the highest value.

(110) 11th aspect: system for reducing perturbations by phase noise in a radar system with units for carrying out the method according to one of the preceding claims, in particular comprising: a first (non-coherent) transmitting-receiving unit (SE1) to generate a first signal (sigTX1) and send, in particular emit, the first signal (sigTX1) via a path (SP), a further, in particular second, (non-coherent) transmitting-receiving unit (SE2) to generate a first signal (sigTX2) and send (in particular emit) the first signal (sigTX2) via the path (SP), wherein the (non-coherent) transmitting-receiving units (SE1 and SE2) are configured to receive the first signals (sigTX1 and sigTX2) in a direct or indirect path and further process them there as receiving signals (sigRX12 and sigRX21), wherein the first transmitting-receiving unit (SE1) is configured to form a comparative signal (sigC12) from the first signal (sigTX1) thereof and from a first signal (sigTX2) received from the further transmitting-receiving unit (SE2) via the path (SP), wherein the further transmitting-receiving unit (SE2) is configured to form a further comparative signal (sigC21) from the first signal (sigTX2) thereof and from such a first signal (sigTX1) received from the first transmitting-receiving unit (SE1) via the path (SP), wherein a transmission unit is provided to transmit, in particular to communicate the further comparative signal (sigC21) from the further transmitting-receiving unit (SE2) to the first transmitting-receiving unit (SE1), wherein at least one evaluation unit is provided which is configured to compensate, in a first step, for deviations of the comparative signals (sigC21 and sigC12) caused by systematic deviations in the transmitting-receiving units (SE2, SE1) and in a second step, to use at least one complex value from a first of the two comparative signals or from a signal which was derived from this first comparative signal to adapt at least one complex value of the second of the two comparative signals or a value of a signal which was derived from this second comparative signal, and thus form an adapted signal (sigCC) wherein the adaptation is performed in such a manner that by means of a mathematical operation the vectorial sum or the difference of the complex values is formed or the difference of the phases of the complex values is formed.

(111) 12th aspect: system according to aspect 11, wherein a bus system, in particular a communication bus, is provided for a clock rate matching, in particular of signal sources of the first signals (sigTX1 and sigTX2), and/or wherein a bus system is provided for determining an offset, in particular time offset and/or frequency offset.

(112) 13th aspect: system according to one of aspects 11 or 12, wherein a common transmitting and receiving antenna is provided in the first and/or further (non-coherent) transmitting-receiving unit (SE1 and/or SE2) and/or wherein a transmission mixer is provided in the path (SP).

(113) 14th aspect: use of the method according to one of claims 1 to 10, for a system having at least one common transmitting and receiving antenna in the first and/or second (non-coherent) transmitting-receiving unit (SE1 and/or SE2).

(114) 15th aspect: use of the system according to one of aspects 11 to 13 to reduce perturbations due to phase noise in a radar system.

(115) At this point, it should be pointed out that all the parts or functions described above when viewed by themselves alone or in any combination, in particular the details shown in the drawings, are claimed as essential to the invention. Modifications thereof are familiar to the person skilled in the art.

(116) FIG. 3 Uncorrelated receiver noise Correlated phase noise Correlated phase noise now interpreted as additive signal

(117) FIG. 4 Beat signals of all ramps: NKSE1 Beat signals of all ramps: NKSE2

(118) FIG. 5 HF signal generator System clock

(119) FIG. 6 HF signal generator HF signal generator System clock

(120) FIG. 7 Noise component of unequal reference clock Phase-locked loop High-frequency local oscillator

(121) FIG. 8 Object Indirect reflection path Direct reflection path HF signal generator HF signal generator System clock