CIRCULAR PILOT SEQUENCES FOR JOINT CHANNEL AND PHASE NOISE ESTIMATION

Abstract

The invention relates to a method for transmitting at least K reference signals in a radio signal to be transmitted over a wireless communication system, said radio signal being intended to be emitted by an emitter comprising at least a transmit antenna configured for emitting on a number M of subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal being provided by: inserting the at least K reference signals P.sub.1, . . . , P.sub.K so that the at least K reference signals P.sub.1, . . . , P.sub.K are respectively transmitted on the K contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K; emitting the radio signal including the at least K reference signals.

Claims

1. A method for transmitting at least K reference signals in a radio signal to be transmitted over a wireless communication system, said radio signal being intended to be emitted by an emitter comprising at least a transmit antenna configured for emitting on a number M of subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal being provided by: inserting the at least K reference signals P.sub.1, . . . , P.sub.K so that the at least K reference signals P.sub.1, . . . , P.sub.K are respectively transmitted on the K contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K; emitting the radio signal including the at least K reference signals; wherein, if K is odd, the values in the frequency domain of the reference signals P.sub.1, . . . , P.sub.(K−1)/2 are respectively equal to the values of the reference signals P.sub.(K+3)/2 , . . . , P.sub.K, if K is even, the values in the frequency domain of the reference signals P.sub.1, . . . , P.sub.K/2 are respectively equal to the values of the reference signals P.sub.K/2+1, . . . , P.sub.K.

2. The method according to claim 1, wherein, if K is odd the values of P.sub.1, . . . , P.sub.(K+1)/2 are respectively a sequence Q.sub.1, . . . , Q.sub.(K+1)/2 such as $\frac{2}{K + 1} {.Math.}_{k = 1}^{\frac{K + 1}{2}} Q_{k} \overline{Q_{{.Math. k - J + 1 .Math.}_{\frac{K + 1}{2}}}}$ is equal to a non-null predetermined value if j is equal to 1 and equal to zero otherwise, with custom-character n.sub.L being 1+mod(n−1, L), with mod(n−1, L) being [n−1] mod L, and if K is even the values of P.sub.1, . . . , P.sub.K/2 are respectively a sequence Q.sub.1, . . . , Q.sub.K/2 such as $\frac{2}{K} {.Math.}_{k = 1}^{\frac{K}{2}} Q_{k} \overline{Q_{{.Math. k - J + 1 .Math.}_{\frac{K}{2}}}}$ is equal to a non-null predetermined value if j is equal to 1 and equal to zero otherwise.

3. The method according to claim 1, wherein at least K+K′ reference signals are transmitted on the M subcarriersS.sub.1, . . . , S.sub.M amongst which at least a number K′ of different subcarriers S.sub.q′+1, S.sub.q′+2, . . . , S.sub.q′+K′ are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q′+1, S.sub.q′+2, . . . , S.sub.q′+K′ being ordered and q′ strictly superior than q+K, said radio signal being further provided by: inserting the at least K′ reference signals P′.sub.1, . . . P′.sub.K′ so that the at least K′ reference signals P′.sub.1, . . . , P′.sub.K′ are respectively transmitted on the K′ contiguous subcarriers S.sub.q′+1, S.sub.q′+2, . . . , S.sub.q′+K′; emitting the radio signal including the at least K+K′ reference signals; wherein, if K′ is odd, the values in the frequency domain of the reference signals P′.sub.1, . . . , P′.sub.(K′−1)/2 are respectively equal to the values of the reference signals P′.sub.(K′+3)/2, . . . , P′_.sub.K′, if K′ is even, the values in the frequency domain of the reference signals P′.sub.1, . . . , P′.sub.K′/2 are respectively equal to the values of the reference signals P′.sub.K′/2+1, . . . , P′.sub.K′.

4. The method according to clais 1, wherein K is set such as K.Δf is greater or equal than 2.ΔPN, with Δf being the subcarrier spacing of at least the subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K and ΔPN is the spectral occupancy of a phase noise of which suffer the radio signal.

5. A method for processing at a receiver a radio signal transmitted over a wireless communication system and received from an emitter comprising at least a transmit antenna configured for emitting on a number M of different subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal including K reference signals, said radio signal being provided according to any one of claim 1, said method comprising: determining a channel estimation, said channel estimation being dependent on a phase noise estimation; processing the radio signal using the channel estimation determined.

6. The method according to claim 5, wherein the determination of the channel estimation comprises: determining symbols Y.sub.n.sub.min, . . . , Y.sub.n.sub.max, said symbols Y.sub.n.sub.min, . . . , Y.sub.n.sub.max in the frequency domain being respectively received on subcarriers S.sub.n.sub.min, . . . , S.sub.n.sub.max, with: if K/2 is an even integer then $n_{\min} = q + \frac{K - 4}{4} and n_{\max} = q + \frac{1}{4} (3 K - 8),$ if K/2 is an odd integer then $n_{\min} = q + \frac{K - 6}{4} and n_{\max} = q + \frac{1}{4} (3 K - 10),$ if (K+1)/2 is an even integer then $n_{\min} = q + \frac{K - 3}{4} and n_{\max} = q + \frac{1}{4} (3 K - 5),$ if (K+1)/2 is an odd integer then $n_{\min} = q + \frac{K - 5}{4} and n_{\max} = q + \frac{1}{4} (3 K - 7);$ computing the channel estimation, said channel estimation being obtained through a linear estimation based on F.sub.K.sub.0.sup.−1{(P.sub.1, . . . , P.sub.K.sub.0)} and F.sub.K.sub.0.sup.−1{(Y.sub.n.sub.min, . . . , Y.sub.n.sub.max)}, with F.sub.K.sub.0.sup.−1{U} being the inverse DFT of order K.sub.0 of the vector U of size K.sub.0, with $K_{0} = \frac{K + 1}{2}$ if K is an oaa integer and $K_{0} = \frac{K}{2}$ if K is an even integer.

7. The method according to claim 6, wherein the determination of the channel estimation further comprises: computing a frequency domain representation H of the channel estimation, such as Ĥ is computed based on
F.sub.K.sub.0.sub.,M.sub.0{(λ.sub.1, . . . , λ.sub.K.sub.0)} where (λ.sub.1, . . . , λ.sub.K.sub.0) is the result of the linear estimation based on F.sub.K.sub.0.sup.−1{(P.sub.1, . . . , P.sub.K.sub.0)} and F.sub.K.sub.0.sup.−1{(Y.sub.n.sub.min, . . . , Y.sub.n.sub.max)} and F.sub.K.sub.0.sub.,j{u} is the j-th terme of DFT of order K.sub.0 of the vector u of size K.sub.0 and with $M_{0} = \frac{M}{2} + 1$ if M is an even integer and $M_{0} = \frac{M + 1}{2}$ if M is an odd integer.

8. The method according to claims 6, further comprises: computing a frequency domain representation {circumflex over (ψ)} of the phase noise estimation, {circumflex over (ψ)}=({circumflex over (ψ)}.sub.1, . . . , {circumflex over (ψ)}.sub.M), such that {circumflex over (ψ)}.sub.(j−M.).sub.M, $M_{*} = \frac{M}{2}$ if M is an even integer and $M_{*} = \frac{M - 1}{2}$ if M is an odd integer, is computed based on
F.sub.K.sub.0.sub.,j{(λ.sub.1, . . . , λ.sub.K.sub.0)} where (λ.sub.1, . . . , λ.sub.K.sub.0) is the result of the linear estimation based on F.sub.K.sub.0.sup.−1{(P.sub.1, . . . , P.sub.K.sub.0)} and F.sub.K.sub.0.sup.−1{(Y.sub.n.sub.min, . . . , Y.sub.n.sub.max)} and F.sub.K.sub.0,.sup.j{u} is the j-th terme of DFT of order K.sub.0 of the vector u of size K.sub.0.

9. The method according to claim 5, wherein, if K is odd the values of P.sub.1, . . . , P.sub.(K+1)/2 are respectively a sequence Q.sub.1, . . . , Q.sub.(K+1)/2 such as $\frac{2}{K + 1} {.Math.}_{k = 1}^{\frac{K + 1}{2}} Q_{k} \overline{Q_{{.Math. k - J + 1 .Math.}_{\frac{K + 1}{2}}}}$ is equal to a non-null predetermined value if j is equal to 1 and equal to zero otherwise, with custom-character n.sub.L being 1+mod(n−1, L), with mod(n−1, L) being [n−1] mod L, and if K is even the values of P.sub.1, . . . , P.sub.k/2 are respectively a sequence Q.sub.1, . . . , Q.sub.K/2 such as $\frac{2}{K} {.Math.}_{k = 1}^{\frac{K}{2}} Q_{k} \overline{Q_{{.Math. k - J + 1 .Math.}_{\frac{K}{2}}}}$ is equal to a non-null predetermined value if j is equal to 1 and equal to zero otherwise, and wherein the determination of the channel estimation comprises: determining symbols Y.sub.n.sub.min, . . . , Y.sub.n.sub.max, said symbols Y.sub.n.sub.min, . . . , Y.sub.n.sub.max in the frequency domain being respectively received on subcarriers S.sub.n.sub.min, . . . , S.sub.n.sub.max, with: if K/2 is an even integer then $n_{\min} = q + \frac{K - 4}{4} and n_{\max} = q + \frac{1}{4} (3 K - 8),$ if K/2 is an odd integer then $n_{\min} = q + \frac{K - 6}{4} and n_{\max} = q + \frac{1}{4} (3 K - 10),$ if (K+1)/2 is an even integer then $n_{\min} = q + \frac{K - 3}{4} and n_{\max} = q + \frac{1}{4} (3 K - 5),$ if (K+1)/2 is an odd integer then $n_{\min} = q + \frac{K - 5}{4} and n_{\max} = q + \frac{1}{4} (3 K - 7);$ computing a frequency domain representation Ĥ of the channel estimation, such as Ĥ is computed based on $\frac{1}{K_{0}} {.Math.}_{n = n_{\min}}^{n_{\max}} Y_{n} .Math. \overline{Q_{{.Math. n - q + 2 - M_{0} .Math.}_{K_{0}}}}$ with $K_{0} = \frac{K + 1}{2}$ if K is an odd integer and $K_{0} = \frac{K}{2}$ if K is an even integer and with $M_{0} = \frac{M}{2} + 1$ if M is an even integer and $M_{0} = \frac{M + 1}{2}$ if M is an odd integer; processing the radio signal using the channel estimation computed.

10. The method according to claim 9, further comprises: computing a frequency domain representation {circumflex over (ψ)} of the phase noise estimation, {circumflex over (ψ)}=({circumflex over (ψ)}.sub.1, . . . , {circumflex over (ψ)}.sub.M) such that ${\hat{ψ}}_{{.Math. j - M_{*} .Math.}_{M}}, M_{*} = \frac{M}{2}$ if M is an even integer and $M_{*} = \frac{M - 1}{2}$ if M is an odd integer, is computed, for k.sub.min≤j≤k.sub.max, based on: $\frac{1}{K_{0}} {.Math.}_{n = n_{\min}}^{n_{\max}} Y_{n} .Math. \overline{Q_{{.Math. n - q + 2 - j .Math.}_{K_{0}}}}$ with $k_{\min} = M_{0} - \frac{K - 4}{4} and k_{\max} = M_{0} + \frac{K}{4}$ if K is an even integer and K/2 is an even integer, $k_{\min} = M_{0} - \frac{K - 2}{4} and k_{\max} = M_{0} + \frac{K - 2}{4}$ if K is an even integer and K/2 is an odd integer, $k_{\min} = M_{0} - \frac{K - 3}{4} and k_{\max} = M_{0} + \frac{K + 1}{4}$ if K is an odd integer and K/2 is an even integer, $k_{\min} = M_{0} - \frac{K - 1}{4} and k_{\max} = M_{0} + \frac{K - 1}{4}$ if K is an odd integer and K/2 is an odd integer, and processing the radio signal using the phase noise estimation {circumflex over (ψ)} computed.

11. The method according to claim 8, wherein the processing of the radio signal comprises: computing estimated symbols {circumflex over (X)}.sub.1, . . . , {circumflex over (X)}.sub.M of symbols X.sub.1, . . . , X.sub.M respectively transmitted on the subcarriers S.sub.1, . . . , S.sub.M, said estimated symbols ({circumflex over (X)}.sub.1, . . . , {circumflex over (X)}.sub.M) being obtained by linear equalization based on Ĥ of R, R being a DFT of order M of e.sup.−i{circumflex over (φ)}⊙y with e.sup.−i{circumflex over (φ)} equal to $(\frac{F_{M, 1}^{- 1} {\hat{ψ}}}{| F_{M, 1}^{- 1} {\hat{ψ}} |}, . . ., \frac{F_{M, M}^{- 1} {\hat{ψ}}}{| F_{M, M}^{- 1} {\hat{ψ}} |}),$ with F.sub.M,m.sup.−1{U} being the m-th terme of the inverse DFT of order M of U and with y being the time domain signal received by the receiver.

12. A computer program product comprising code instructions to perform the method according to claim 1, when said instructions are run by a processor.

13. An emitter for transmitting at least K reference signals in a radio signal to be transmitted over a wireless communication system, said radio signal being intended to be emitted by the emitter, said emitter comprising: at least a transmit antenna configured for emitting on a number M of subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, a processor; and a non-transitory computer-readable medium comprising instructions stored thereon, which when executed by the processor configure the emitter to: insert the at least K reference signals P.sub.1, . . . , P.sub.K so that the at least K reference signals P.sub.1, . . . , P.sub.K are respectively transmitted on the K contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K; emit the radio signal including the at least K′- reference signals; wherein, if K is odd, the values in the frequency domain of the reference signals P.sub.1, . . . , P.sub.(K−1)/2 are respectively equal to the values of the reference signals P.sub.(K+3)/2, . . . , P.sub.K if K is even, the values in the frequency domain of the reference signals P.sub.1, . . . , P.sub.K/2 are respectively equal to the values of the reference signals P.sub.K/2+1, . . . , P.sub.K.

14. A receiver for processing a radio signal transmitted over a wireless communication system and received from an emitter comprising at least a transmit antenna configured for emitting on a number M of different subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal including K reference signals, said radio signal being provided according to claim 1, said receiver comprising: at least one receiving antenna; a processor; and a non-transitory computer-readable medium comprising instructions stored thereon, which when executed by the processor configure the receiver to: determine a channel estimation, said channel estimation being dependent on a phase noise estimation; processing the radio signal using the channel estimation determined.

15. A receiver for processing a radio signal transmitted over a wireless communication system and received from an emitter comprising at least a transmit antenna configured for emitting on a number M of different subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal including K reference signals, said radio signal being provided according to claim 2, said receiver comprising: at least one receiving antenna; a processor; and a non-transitory computer-readable medium comprising instructions stored thereon, which when executed by the processor configure the receiver to: determine a channel estimation, said channel estimation being dependent on a phase noise estimation; processing the radio signal using the channel estimation determined.

16. A receiver for processing a radio signal transmitted over a wireless communication system and received from an emitter comprising at least a transmit antenna configured for emitting on a number M of different subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal including K reference signals, said radio signal being provided according to claim 3, said receiver comprising: at least one receiving antenna; a processor; and a non-transitory computer-readable medium comprising instructions stored thereon, which when executed by the processor configure the receiver to: determine a channel estimation, said channel estimation being dependent on a phase noise estimation; processing the radio signal using the channel estimation determined.

17. A receiver for processing a radio signal transmitted over a wireless communication system and received from an emitter comprising at least a transmit antenna configured for emitting on a number M of different subcarriers S.sub.1, . . . , S.sub.M amongst which at least a number K of different subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K are contiguous, the respective frequencies of the contiguous subcarriers S.sub.q+1, S.sub.q+2, . . . , S.sub.q+K being ordered, said radio signal including K reference signals, said radio signal being provided according to claim 4, said receiver comprising: at least one receiving antenna; a processor; and a non-transitory computer-readable medium comprising instructions stored thereon, which when executed by the processor configure the receiver to: determine a channel estimation, said channel estimation being dependent on a phase noise estimation; processing the radio signal using the channel estimation determined.

Description

BRIEF DESCRIPTION OF DRAWINGS

[0100] The present invention is illustrated by way of example, and not by way of limitations, in the figures of the accompanying drawings, in which like reference numerals refer to similar elements and in which:

[0101] FIG. 1 illustrates a transmitter and receiver according to the invention.

[0102] FIG. 2 schematizes a block diagram of a transmitter according to the invention.

[0103] FIG. 3 details an example of a reference signal pattern according to the invention.

[0104] FIG. 4 schematizes a block diagram of a receiver according to the invention.

[0105] FIG. 5 illustrates a flowchart representing the steps of radio signal processing according to the invention.

[0106] FIG. 6 illustrates a flowchart representing the steps of radio signal decoding according to the invention.

DESCRIPTION OF EMBODIMENTS

[0107] Referring to FIG. 1, there is shown a transmitter 1.1 transmitting a radio signal to a receiver 1.2. The receiver 1.1 is in the cell of the transmitter 1.2. This transmission may be an OFDM based transmission. In this example the transmitter 1.1 is a fixed station and the receiver 1.2 is a mobile terminal. In the context of LTE the fixed station and the mobile terminal are respectively referred to as a base station and a user equipment. The transmitter 1.1 can as well be the mobile terminal and the receiver 1.2 the fixed station.

[0108] The transmitter 1.1 comprises one communication module (COM_trans) 1.3, one processing module (PROC_trans) 1.4 and a memory unit (MEMO_trans) 1.5. The MEMO_trans 1.5 comprises a non-volatile unit which retrieves the computer program and a volatile unit which retrieves the reference signal pattern parameters, for example the tuple (q.sub.i, μ.sub.i, K.sup.i, Q.sup.i.sub.1, . . . , Q.sup.i.sub.K.sub.0.sub.i). The PROC_trans 1.4 is configured to insert the reference signals according to the invention. The COM_trans is configured to transmit to the receiver 1.2 the radio signal. The processing module 1.4 and the memory unit 1.5 may constitute the device for inserting the reference signals, as previously described. The processing module 1.4 and the memory unit 1.5 can be dedicated to this task or also used for other functions of the transmitter like for processing the radio signal.

[0109] The receiver 1.2 comprises one communication module (COM_recei) 1.6, one processing module (PROC_recei) 1.7 and a memory unit (MEMO_recei) 1.8. The MEMO_recei 1.8 comprises a non-volatile unit which retrieves the computer program and a volatile unit which retrieves the reference signal pattern parameters, for example the tuple (q.sub.i, μ.sub.i, K.sup.i, Q.sup.i.sub.1, . . . , Q.sup.i.sub.K.sub.0.sub.i). The PROC_recei 1.7 is configured to deteiiiiine a channel estimation and a noise phase estimation and to process the radio signal according to the estimations to retrieve the other symbols emitted by the transmitter 1.1. The COM_recei 1.6 is configured to receive from the transmitter the radio signal. The processing module 1.7 and the memory unit 1.8 may be dedicated to these tasks, as previously described. The processing module 1.7 and the memory unit 1.8 may also be used for other functions of the receiver.

[0110] Referring to FIG. 2, there is shown a block diagram of a transmitter 1.1 according to the invention. Such OFDM transmitter 1.1 applies an OFDM scheme on a block of N′ symbols to obtain the radio signal. In the example of FIG. 2 the OFDM transmitter emits a radio signal by emitting on one transmit antenna Tx 2.0, this is none limiting and the OFDM transmitter can as well transmit by using several transmit antennas, for example in a MIMO context. However, in the context where several antennas are used the reference signal pattern may be identical for each antenna or only one antenna transmits the RS according to the invention, the RS of the RS pattern being replaced by zeros for the other antennas.

[0111] To provide the radio signal a serial to parallel (S/P) module 2.1 is applied to the block of N′ symbols X′=(X′.sub.1, . . . X′.sub.N). The symbols of the block of symbols may be N′ complex symbols obtained by a QPSK digital modulation scheme or any other digital modulation scheme as QAM, or may be symbols of a sequence with controlled PAPR (e.g. a CAZAC sequence).

[0112] At the output of the S/P module 2.1, the parallel symbols are mapped, with a subcarrier mapping module 2.2 in the frequency domain to N (>N′) out of M subcarriers (S.sub.1, . . . S.sub.M). Regarding the subcarrier mapping, the complex symbols are mapped to the N allocated subcarriers out of M existing subcarriers via subcarrier mapping module 2.2. The subcarrier mapping can be for example localized, that is, the N′ complex symbols are mapped throughout N consecutive subcarriers among the M existing. This subcarrier mapping is done in accordance with the reference signal pattern used by the transmitter 1.1. Therefore, the N-N′ allocated subcarriers to which none of the N′ complex symbols have been mapped correspond to the subcarriers which transmit the RS according to the RS pattern. Therefore, the RS insertion module 2.3 adds to these unused N-N′ subcarriers the RS according to the RS pattern as described in FIG. 3. The subcarrier mapping module 2.2 is therefore, parametrized to map the N′ symbols on other subcarriers than the N-N′ subcarriers intended to be used to transmit the reference signal according to the RS pattern of the invention. That is, for example to let unused the subcarriers ∪.sub.i=1.sup.L{S.sub.qi+1, . . . , S.sub.qi+K.sup.i}.

[0113] M-size inverse DFT module 2.4 is then applied to the resulting vector of M symbols X.sub.1, . . . , X.sub.M, the M symbols being the N non-null symbols (comprising the RS of the RS pattern) and M-N null symbols (according to the subcarrier mapping scheme), therefore generating an OFDM symbol which is transmitted via the transmit antenna 2.0. More precisely, at the output of the IDFT module 2.4 a signal {tilde over (x)} is obtained. This signal occupies during a time interval corresponding to an OFDM symbol, N allocated subcarriers out of the M existing subcarriers. This time-domains signal {tilde over (x)} corresponds to an OFDM symbol.

[0114] A cyclic prefix can be optionally appended after the IDFT by the CP module 2.5. In addition, the digital-to-analog converter (DAC) module 2.6 converts the digital signal resulting from the IDFT module 2.4 to an analog signal that can be transmitted through the antenna 2.0.

[0115] Referring to FIG. 3, there is shown an example of a reference signal pattern according to the invention.

[0116] The invention specifies specific positions (that is the subcarriers used to transmit the reference signals) and values for the reference signals. This specific reference signal pattern according to the invention (or simply the reference signal pattern) enables to have specific properties of the radio signal enabling to reduce errors during its decoding. However, this does not limit the use of the other subcarriers, that is, the N′ subcarriers can be used to transmit any types of symbols, for example other reference signals like DM-RS or PTRS, symbols transmitting control data or user data.

[0117] An example of a RS pattern specified by the invention is described at FIG. 3, the reference signals are positioned by groups of RS. On FIG. 3, L groups of RS are configured. The i-th group of RS is transmitted on the subcarriers S.sub.qi+1, . . . , S.sub.qi+K.sup.i, for i from 1 to L. The position of the first symbol in the group has to be greater than the last position of the previous group, that is, q.sub.i+K.sup.i<q.sub.i+1+1 for i from 1 to L−1. These positions are among the M positions of the subcarriers used in the bandwidth by the transmitter. Only one group of RS may be set in the RS pattern, then, the group of RS is transmitted on the subcarriers S.sub.q+1, . . . , S.sub.q+K.

[0118] For the i-th group, the values in the frequency domain of the reference signals P.sup.i.sub.1, . . . , P.sup.i.sub.K.sub.0.sub.i.sub.−1 (transmitted respectively on the subcarriers S.sub.qi+1, . . . , S.sub.qi+K.sub.0.sub.i.sub.−1) are respectively equal to the values of the reference signals P.sup.i.sub.K.sub.0.sub.i.sub.+1, . . . , P.sup.i.sub.K.sub.i (transmitted respectively on the subcarriers S.sub.qi+K.sub.0.sub.i.sub.+1, . . . , S.sub.K.sup.i), with

[00045] $K_{0}^{i} = \frac{K^{i} + 1}{2}$

if K.sup.i is an odd integer and

[00046] $K_{0}^{i} = \frac{K^{i}}{2}$

if K.sup.i is an even integer.

[0119] In addition, the group of reference signals P.sup.i.sub.1 , . . . , P.sup.i.sub.K.sup.i may be issued from a sequence of Q.sup.i.sub.1, . . . , Q.sup.i.sub.K.sub.0.sub.i which satisfies an auto-correlation condition, that is, such as

[00047] $\frac{1}{K_{0}^{i}} {.Math.}_{k = 1}^{K_{0}^{i}} Q_{k}^{i} \overline{Q_{{.Math. k -_{J} + 1 .Math.}_{K_{0}^{.Math.}}}^{.Math.}}$

equal to a non-null predetermined value if j is equal to 1 and equal to zero otherwise. Only some of the groups of RS may be issued from such sequences.

[0120] These sequences may be CAZAC sequences, for example Zadoff-Chu sequences.

[0121] The size K.sup.i of each group of reference signals may be chosen such as described after according to the spectral occupancy of the phase noise, or at least to the spectral occupancy of the modelized phase noise. The size K.sup.i of each group may be set such as the channel is constant or can be assimilated as such on a scale of K.sup.i.Δf. Therefore, the results are better if the channel is constant at least on a scale of 2.ΔPN.

[0122] The number L of groups of reference signals may be chosen according to the variation of channel in the spectrum. Indeed, if the channel is sensitive to frequency then it may be relevant to have an important density of groups of reference signals through the bandwidth used for the communication. Adventurously these groups of reference signals may be uniformly distributed through the bandwidth (all the μ.sub.i are equal or similar). If the channel is not sensitive to frequency then only one or two groups of reference signals may be needed to have good channel and/or phase noise estimation through all the bandwidth.

[0123] Referring to FIG. 4, there is shown a block diagram of a receiver 1.2 according to the invention. Such a receiver is configured to decode a radio signal emitted by a transmitter 1.1 as previously described. This example shows a receiver with a unique receive antenna but such receiver can have several receive antennas. When using several antennas, the radio signal received by each antenna differs which introduces receive diversity. In this example said radio signal is received on one antenna Rx 4.0. After applying a analog-to-digital converter ADC module 4.1 to the received radio signal and after an optional guard removal (by the CP module 4.2) the resulting signal y is inputted into a M-size DFT 4.3. The results, at the outputs of the DFT 4.3 are M symbols Y.sub.1, . . . , Y.sub.M respectively received on the subcarriers S.sub.1, . . . , S.sub.M.

[0124] The RS extraction module 4.4 extracts a block of symbols from the M symbols Y.sub.1, . . . , Y.sub.M. More specifically, the RS extraction module 4.4 extracts K.sub.0.sup.i received contiguous symbols Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i on the subcarriers S.sub.n.sub.min.sub.i, . . . , S.sub.n.sub.max.sub.i. n.sub.min.sup.i and n.sub.max.sup.i are defined so that each of these extracted symbols are only composed of samples of the reference signals P.sup.i.sub.1, . . . , P.sup.i.sub.K.sub.i of the block of K.sup.i reference signals. These received symbols Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i may also contain samples from other symbols but these samples are of low energy compared to the samples of the reference signals set in block. Indeed, by considering a typical model of phase noise such as a Wiener processed phase noise (also known as a Brownian motion)) it is possible to determine the number of contiguous symbols Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i that do not contain samples of other symbols. In addition, advantageously the sizes K.sup.i of the blocks of reference signals are chosen according to the spectral occupancy of the phase noise ΔPN, that is, for each i, K.sup.i.Δf is equal or greater than 2ΔPN. Advantageously, the K.sup.i are all equal since the spectral occupancy of the phase noise ΔPN is the same for each blocks of reference signals. This enables to ensure that a sufficiently important block of symbols Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i only composed of samples of the reference signals P.sup.i.sub.1, . . . , P.sup.i.sub.K.sub.i is received. For instance, if K.sup.i is greater than or equal to 2ΔPN/Δf, then n.sub.min and n.sub.max can be defined as: [0125] if K.sup.i/2 is an even integer then

[00048] $n_{\min}^{i} = q + \frac{K^{i} - 4}{4} and n_{\max}^{i} = q + \frac{1}{4} (3 K^{i} - 8),$ [0126] if K.sup.i/2 is an odd integer then

[00049] $n_{\min}^{i} = q + \frac{K^{i} - 6}{4} and n_{\max}^{i} = q + \frac{1}{4} (3 K^{i} - 1 0),$ [0127] if (K.sup.i+1)/2 is an even integer then

[00050] $n_{\min}^{i} = q + \frac{K^{i} - 3}{4} and n_{\max}^{i} = q + \frac{1}{4} (3 K^{i} - 5),$ [0128] if (K.sup.i+1)/2 an odd integer then

[00051] $n_{\min}^{i} = q + \frac{K^{i} - 5}{4} n_{\max}^{i} = q + \frac{1}{4} (3 K^{i} - 7) .$

[0129] In this case, for each i, Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i are:

[00052] ${\begin{matrix} Y_{n_{\min}^{i}} & = & ψ_{k_{\min}} X_{q^{i} + K_{0}^{i} - 1} H_{q^{i} + K_{0}^{i} - 1} + .Math. + ψ_{j} X_{q^{i} + k_{\max} - j} H_{q^{i} + k_{\max} - j} + .Math. + ψ_{k_{\max}} X_{q^{i}} H_{q^{i}} Z_{n_{\min}^{i}} \\ .Math. & .Math. & .Math. \\ Y_{n} & = & ψ_{k_{\min}} X_{q^{i} + K_{0}^{i} - 1 + n - n_{\min}^{i}} H_{q^{i} + K_{0}^{i} - 1 + n - n_{\min}^{i}} + .Math. + ψ_{j} X_{q^{i} + k_{\max} - j + n - n_{\min}^{i}} H_{q^{i} + k_{\max} - j + n - n_{\min}^{i}} + .Math. \\ .Math. & + & ψ_{k_{\max}} X_{q^{i} + n - n_{\min}^{i}} H_{q^{i} + n - n_{\min}^{i}} + Z_{n} \\ .Math. & .Math. & .Math. \\ Y_{n_{\max}^{i}} & = & ψ_{k_{\min}} X_{q^{i} + 2 K_{0}^{i} - 2} H_{q^{i} + 2 K_{0}^{i} - 2} + .Math. + ψ_{j} X_{q^{i} + k_{\max} - j + K_{0}^{i}} H_{q^{i} + k_{\max} - j + K_{0}^{i}} + \\ .Math. & + & ψ_{k_{\max}} X_{q^{i} + K_{0}^{i} - 1} H_{q^{i} + K_{0}^{i} - 1} Z_{n_{\max}^{i}} \end{matrix}$

[0130] Where X.sub.j is the symbol emitted on the subcarrier S.sub.j and (Z.sub.n.sub.min.sub.i, . . . , Z.sub.n.sub.max.sub.i) represents in the frequency domain additive white Gaussian noise (AWGN). Therefore, by replacing the values of the reference signals we obtain:

[00053] ${\begin{matrix} Y_{n_{\min}^{i}} & = & ψ_{k_{\min}} Q_{K_{0}^{i}}^{i} H_{q^{i} + K_{0}^{i} - 1} + .Math. + ψ_{j} Q_{k_{\max} - j + 1}^{i} H_{q^{i} + k_{\max} - j} + .Math. + ψ_{k_{\max}} Q_{1}^{i} H_{q^{i}} + Z_{n_{\min}^{i}} \\ .Math. & .Math. & .Math. \\ Y_{n} & = & ψ_{k_{\min}} Q_{{.Math. K_{0}^{i} + n - n_{\min}^{i} .Math.}_{K_{0}^{i}}}^{i} H_{q^{i} + K_{0}^{i} - 1 + n - n_{\min}^{i}} + .Math. + ψ_{j} Q_{{.Math. k_{\max} - j + 1 + n - n_{\min}^{i} .Math.}_{K_{0}^{i}}}^{i} H_{q^{i} + k_{\max} - j + n - n_{\min}^{i}} + .Math. \\ .Math. & + & ψ_{k_{\max}} {Q_{.Math. 1 + n - n_{\min}^{i} .Math.}^{i}}_{K_{0}^{i}} H_{q^{i} + n - n_{\min}^{i}} + Z_{n} \\ .Math. & .Math. & .Math. \\ Y_{n_{\max}^{i}} & = & ψ_{k_{\min}} Q_{K_{0}^{i} - 1}^{i} H_{q^{i} + 2 K_{0}^{i} - 2} + .Math. + ψ_{j} Q_{k_{\max} - j}^{i} H_{q^{i} + k_{\max} - j + K_{0}^{i}} + .Math. + ψ_{k_{\max}} Q_{K_{0}^{i}}^{i} H_{q^{i} + K_{0}^{i} - 1} + Z_{n_{\max}^{i}} \end{matrix}$

[0131] To simplify we assume that H is constant on the bandwidth corresponding to subcarriers S.sub.q.sub.i to S.sub.q.sub.i.sub.+2K.sub.0.sub.i.sub.−2, which are the subcarriers transmitting reference signals P.sup.i.sub.1, . . . , P.sup.i.sub.K.sub.i. In case of millimeter-Wave systems this assumption is generally not restrictive.

[0132] Therefore,

[00054] ${\begin{matrix} Y_{n_{\min}^{i}} & = & H^{i} (ψ_{k_{\min}} Q_{K_{0}^{i}}^{i} + .Math. + ψ_{j} Q_{k_{\max} - j + 1}^{i} + .Math. + ψ_{k_{\max}} Q_{1}^{i}) + Z_{n_{\min}^{i}} \\ .Math. & .Math. & .Math. \\ Y_{n} & = & H^{i} (ψ_{k_{\min}} Q_{{.Math. K_{0}^{i} + n - n_{\min}^{i} .Math.}_{K_{0}^{i}}}^{i} + .Math. + ψ_{j} Q_{{.Math. k_{\max} - j + 1 + n - n_{\min}^{i} .Math.}_{K_{0}^{i}}}^{i} + .Math. + ψ_{k_{\max}} Q_{{.Math. 1 + n - n_{\min}^{i} .Math.}_{K_{0}^{i}}}^{i}) + Z_{n} \\ .Math. & .Math. & .Math. \\ Y_{n_{\max}^{i}} & = & H^{i} (ψ_{k_{\min}} Q_{K_{0}^{i} - 1}^{i} + .Math. + ψ_{j} Q_{k_{\max} - j}^{i} + .Math. + ψ_{k_{\max}} Q_{K_{0}^{i}}^{i}) + Z_{n_{\max}^{i}} \end{matrix}$

[0133] As mentioned above, K.sup.i may be set greater than or equal to 2ΔPN/Δf and H may be assumed as constant on the bandwidth corresponding to subcarriers S.sub.q.sub.i to S.sub.q.sub.i.sub.+2K.sub.0.sub.1.sub.−2. However, in the case H is not constant and/or 2ΔPN is greater than K.sup.iΔf then the invention may still be applied with good results but less accurate than when those conditions are met.

[0134] Once these Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i are extracted, the channel and phase noise estimation module 4.5 computes channel estimation and a phase noise estimation.

[0135] Two different algorithms using the specificities of the specific RS pattern may be implemented by the channel and phase noise estimation module 4.5.

[0136] In the first algorithm, the channel and phase noise estimation module 4.5 computes the linear estimation of F.sub.K.sub.0.sub.i.sup.−1{(P.sub.1.sup.i, . . . , P.sub.K.sub.0.sub.i.sup.i)} by F.sub.K.sub.0.sub.i.sup.−1{(Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i)} with F.sub.K.sub.0.sub.i.sup.−1{U} the inverse DFT of order K.sub.0.sup.i of the vector U of size K.sub.0.sup.i, with

[00055] $K_{0}^{i} = \frac{K^{i} + 1}{2}$

if K.sup.i is an odd integer and

[00056] $K_{0}^{i} = \frac{K^{i}}{2}$

if K.sup.i is an even integer, to obtain the vector (λ.sub.1.sup.i, . . . , λ.sub.K.sub.0.sub.i.sup.i). The linear estimation may be: [0137] a zero-forcing-like estimation:

[00057] ${\hat{H}}^{i} .Math. F_{K_{0}^{i}}^{- 1} {(, .Math.,)} = F_{K_{0}^{i}}^{- 1} {(Y_{n_{\min}^{i}}, .Math., Y_{n_{\max}^{i}})} ⊙ \frac{1}{F_{K_{0}^{i}}^{- 1} {(P_{1}, .Math., P_{K_{0}^{i}})}};$ [0138] MMSE-like estimation:
Ĥ.sup.i.F.sub.K.sub.0.sub.i.sup.−1{( custom-character , . . . , )}=(F.sub.K.sub.0.sub.i.sup.−1{P.sub.1, . . . , P.sub.K.sub.0.sub.i)}.sup.†F.sub.K.sub.0.sub.i.sup.−1{(P.sub.1, . . . , P.sub.K.sub.0.sub.i)}+V).sup.−1F.sub.K.sub.0.sub.i.sup.−1{(P.sub.1, . . . , P.sub.K.sub.0.sub.i)}.sup.†F.sub.K.sub.0.sub.i.sup.−1{(Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i)} with V.sup.i the covariance matrix of size K.sub.0.sup.i of vector F.sub.K.sub.0.sub.i.sup.−1{(Z.sub.n.sub.min.sub.i, . . . , Z.sub.n.sub.max.sub.i )} and † is the conjugate transpose operator.

[0139] The channel and phase noise estimation module 4.5 computes the frequency domain representation Ĥ of the channel estimation such as Ĥ is equal to Ĥ.sup.i=F.sub.K.sub.0.sub.i.sub.,M.sub.0{(λ.sub.1.sup.i, . . . , λ.sub.K.sub.0.sub.i.sup.i)} (with

[00058] $M_{0} = \frac{M}{2} + 1$

if M is an even integer and

[00059] $M_{0} = \frac{M + 1}{2}$

if M is an odd integer) on the bandwidth corresponding to subcarriers S.sub.q.sub.i to S.sub.q.sub.i.sub.+2K.sub.0.sub.i.sub.−2. Indeed, the central component of the phase noise is the most powerful component, therefore, this central component, component M.sub.0, is less impacted by the additive noise, which therefore can be assumed as negligible.

[0140] The channel and phase noise estimation module 4.5 computes a group of parameters (Λ.sub.1.sup.i, . . . , Λ.sub.K.sub.min.sub.i.sup.i, . . . , Λ.sub.M.sup.i), with Λ.sub.j.sup.i=F.sub.K.sub.0.sub.i.sub.,j{(λ.sub.1.sup.i, . . . , λ.sub.K.sub.0.sub.i.sup.i)} for j from k.sub.min to k.sub.max and zero otherwise, with F.sub.K.sub.0.sub.i.sub.,j{u} is the j-th terme of DFT of order K.sub.0.sup.i of the vector u of size K.sub.0.sup.i. Then the frequency domain representation {circumflex over (ψ)} of the phase noise estimation, {circumflex over (ψ)}=({circumflex over (ψ)}.sub.1, . . . , {circumflex over (ψ)}.sub.M), can be computed based on the group of parameters (Λ.sub.1.sup.i, . . . , Λ.sub.K.sub.min.sub.i.sup.i, . . . , Λ.sub.k.sub.max.sub.i.sup.i, . . . , Λ.sub.M.sup.i). For example, {circumflex over (ψ)}.sub.j=e.sup.−i.arg Ĥ.sup.iΛ.sub.j.sup.i. In another example, each {circumflex over (ψ)}.sub.j can be computed based on an coherent average of the parameters through the RS pattern, that is, for example,

[00060] ${\hat{ψ}}_{j} = \frac{1}{L} {.Math.}_{i = 1}^{L} e^{- i .Math. \arg {\hat{H}}^{i}} Λ_{j}^{i} .$

This enables to enhance the accuracy of the phase noise estimation. Multiplying Λ.sub.j.sup.i by e.sup.−i.arg Ĥ.sup.i compute phase noise component {circumflex over (ψ)}.sub.j for which the phase does not depend on the channel estimation.

[0141] In the second algorithm, when the P.sup.i.sub.1, . . . , P.sup.i.sub.K.sub.i are issued from a sequence of Q.sup.i.sub.1, . . . , Q.sup.i.sub.K.sub.0.sub.i which satisfies the auto-correlation condition (for example a Zadoff-Chu sequences), channel and phase noise estimation module 4.5 computes the frequency domain representation Ĥ of the channel estimation such as Ĥ is computed based on

[00061] $\frac{1}{K_{0}^{i}} {.Math.}_{n = n_{\min}^{1}}^{n_{\max}^{i}} Y_{n} .Math. \overline{Q_{{.Math. n - q^{i} + 2 - M_{0} .Math.}_{K_{0}^{i}}}^{i}}$

with

[00062] $K_{0}^{i} = \frac{K^{i} + 1}{2}$

if K.sup.i is an odd integer and

[00063] $K_{0}^{i} = \frac{K^{i}}{2}$

if K.sup.i is an even integer.

[0142] For example, Ĥ is equal to

[00064] ${\hat{H}}^{i} = \frac{1}{A^{i} K_{0}^{i}} {.Math.}_{n = n_{\min}^{i}}^{n_{\max}^{i}} Y_{n} .Math. \overline{Q_{{.Math. n - q^{i} + 2 - M_{0} .Math.}_{K_{0}^{i}}}^{i}}$

on the bandwidth corresponding to subcarriers S.sub.q.sub.i to S.sub.q.sub.i.sub.+2K.sub.0.sub.i.sub.−2, with

[00065] $A^{i} = \frac{1}{K_{0}^{i}} {.Math.}_{n = 1}^{K_{0}^{i}} Q_{n}^{i} \overline{Q_{{.Math. n .Math.}_{K_{0}^{.Math.}}}^{.Math.}} .$

[0143] The channel and phase noise estimation module 4.5 computes a group of parameters (Λ.sub.1.sup.i, . . . , Λ.sub.k.sub.min.sub.i.sup.i, . . . , Λ.sub.k.sub.max.sub.i.sup.i, . . . Λ.sub.M.sup.i) with

[00066] $Λ_{j}^{i} = \frac{1}{A^{i} K_{0}^{i}} {.Math.}_{n = n_{\min}^{i}}^{n_{\max}^{i}} Y_{n} .Math. \overline{Q_{{.Math. n - q^{i} + 2 - j .Math.}_{K_{0}^{i}}}^{i}} .$

Then the frequency domain representation {circumflex over (ψ)} of the phase noise estimation, {circumflex over (ψ)}=({circumflex over (ψ)}.sub.1, . . . , {circumflex over (ψ)}.sub.M), can be computed based on the group of parameters (Λ.sub.1.sup.i, . . . , Λ.sub.k.sub.min.sub.i.sup.i, . . . , Λ.sub.k.sub.max.sub.i.sup.i, . . . Λ.sub.M.sup.i). For example, {circumflex over (ψ)}.sub.(j−M.).sub.M=e.sup.−i.arg H.sup.i.sup.M.sub.0Λ.sub.j.sup.i for K.sub.min.sup.i≤j≤k.sub.max.sup.i, with

[00067] $M_{*} = \frac{M}{2}$

if M is an even integer and

[00068] $M_{*} = \frac{M - 1}{2}$

if M is an odd integer, with

[00069] $k_{\min}^{i} = M_{0} - \frac{K^{i} - 4}{4} and k_{\max}^{i} = M_{0} + \frac{K^{i}}{4}$

if K.sup.i is an even integer and K.sup.i/2 is an even integer,

[00070] $k_{\min}^{i} = M_{0} - \frac{K^{i} - 2}{4} and k_{\max}^{i} = M_{0} + \frac{K^{i} - 2}{4}$

if K.sup.i is an even integer and K.sup.i/2 is an odd integer,

[00071] $k_{\min}^{i} = M_{0} - \frac{K^{i} - 3}{4} and k_{\max}^{i} = M_{0} + \frac{K^{i} + 1}{4}$

if K.sup.i is an odd integer and K.sup.i/2 is an even integer,

[00072] $k_{\min}^{i} = M_{0} - \frac{K^{i} - 1}{4} and k_{\max}^{i} = M_{0} + \frac{K^{i} - 1}{4}$

if K.sup.i is an odd integer and K.sup.i/2 is an odd integer. In another example, each {circumflex over (ψ)}.sub.(j−M.).sub.M can be computed based on an average of the parameters through the RS pattern, that is, for example,

[00073] ${\hat{ψ}}_{{.Math. j - M_{*} .Math.}_{M}} = \frac{1}{L} {.Math.}_{i = 1}^{L} e^{- i . \arg {\hat{H}}^{i}} Λ_{j}^{i},$

for k.sub.min.sup.i≤j≤k.sub.max.sup.i. This enables to enhance the accuracy of the phase noise estimation.

[0144] Once the channel and phase noise estimation are computed by the channel and phase noise estimation module 4.5, either by the first or the second algorithm, the equalization module 4.6 performs a linear equalization based on Ĥ of R, R being a DFT of order M of e.sup.−i{circumflex over (φ)}⊙y with e.sup.−i{circumflex over (φ)}equal to

[00074] $(\frac{F_{M, 1}^{- 1} {\hat{ψ}}}{.Math. F_{M, 1}^{- 1} {\hat{ψ}} .Math.}, .Math., \frac{F_{M, M}^{- 1} {\hat{ψ}}}{.Math. F_{M, M}^{- 1} {\hat{ψ}} .Math.}),$

with F.sub.M,m.sup.−1{U} being the m-th terme of the inverse DFT of order M of U and with y being the time domain signal received by the receiver. A⊙B is the Hadamard product. For example, R=F.sub.M{e.sup.−i{circumflex over (φ)}} custom-character (Y.sub.1, . . . , Y.sub.M), with F.sub.M{u} the DFT of order M of the vector u of size M and a linear equalization based on Ĥ of R is performed. From such linear equalization results estimated symbols ({circumflex over (X)}.sub.1, . . . , {circumflex over (X)}.sub.M)={circumflex over (X)} of the symbols X.sub.1, . . . , X.sub.M respectively transmitted through the subcarriers S.sub.1, . . . , S.sub.M. For example, estimated symbols ({circumflex over (X)}.sub.1, . . . , {circumflex over (X)}.sub.M) may be obtained by a minimum mean-squared error (MMSE) equalization, that is:

{circumflex over (X)}=WR

[0145] Where W is a diagonal matrix

[00075] $(\begin{matrix} W^{1} & 0 & 0 \\ 0 & ⋱ & 0 \\ 0 & 0 & W^{L} \end{matrix}) with W^{i} = \frac{\overline{{\hat{H}}^{.Math.}}}{{.Math. {\hat{H}}^{i} .Math.}^{2} + σ^{2}},$

with σ.sup.2 is the variance of the additive white Gaussian noise measured at the channel output.

[0146] It is then applied to the result of the linear equalization a subcarrier demapping module 4.7 and a parallel to serial module 4.8 at the output of which the N symbols emitted, including the N′ symbols, are retrieved.

[0147] Referring to FIG. 5 there is shown a flowchart representing the steps of radio signal processing according to the invention.

[0148] At step S11 a RS pattern stored in the memory unit 1.5 is selected. The selection may be either static or dynamically. When the RS pattern is dynamically selected, the transmitter 1.1 may change, for example for each OFDM symbol or for a number of OFDM symbols, the RS pattern used for the insertion of RS. This selection may be done according to feedbacks from the receiver 1.2 through a control channel. In the case of a dynamic selection of the RS pattern the transmitter may choose another configuration upon those saved in the MEMO_trans 1.5. Indeed, several configurations may be stored in the MEMO_trans 1.5, those configurations can be ordered according to the number of reference signals (Σ K.sup.i) and/or the number of groups of RS the RS pattern provides. A RS pattern may be defined by the number Σ K.sup.i of reference signals, by the number of groups (L) of RS or by the positions of the RS in the frequency domain.

[0149] The transmitter 1.1 may select a RS pattern based on the communication configuration (subcarrier spacing configuration, carrier frequency range, modulation and coding scheme, carrier frequency, resource allocation unit) and radio channel characteristics (strong phase noise variation, strong sensitivity to the frequency) of the transmission.

[0150] At step S12 the subcarrier mapping module 2.2 and the RS insertion module 2.3 are configured according to the RS pattern stored in the memory unit 1.5 and used for the transmission. Therefofe, the subcarrier mapping module 2.2 is configured to map the N′ symbols at its inputs on subcarriers that will not be occupied, according to the RS pattern, by the Σ K.sup.i reference signals.

[0151] At step S13 the RS insertion module 2.3, inserts the reference signals on positions defined by the RS pattern, that is, on the subcarriers ∪.sub.i=1.sup.L{S.sub.qi+1, . . . , S.sub.qi+K.sub.i} as previously described. The other subcarriers are occupied by the N′ symbols to be transmitted and by zeros according to the scheme of the subcarrier mapping.

[0152] At step S14 the signal is processed, that is on the M symbols X=(X.sub.1, . . . X.sub.M) is applied an OFDM scheme (IDFT module 2.4, CP module 2.5 and DAC module 2.6).

[0153] At step S15 the signal is emitted by Tx 2.0.

[0154] Referring to FIG. 6 there is shown a flowchart representing the steps of radio signal decoding according to the invention.

[0155] At step S21 the RS extractor module 4.4, the channel and phase noise estimation module 4.5 and the equalization module 4.6 are configured according to the configuration of the RS insertion module 2.3. For this purpose the receiver 1.2 may receive, for example from the transmitter 1.1, the RS pattern used for the transmission. The same RS pattern stored in the MEMO_trans 1.5 may be stored in the MEMO_recei 1.8. The transmitter 1.1 can optionally send control information to the receiver 1.2 through a control channel, this control information pointing the RS pattern selected for the transmission.

[0156] At step. S22 the RS extraction module 4.4 extracts parts of the symbols Y.sub.1, . . . , Y.sub.M outputted by the DFT module 4.3. More specifically, the RS extraction module 4.4 extracts the symbols ∪.sub.i=1.sup.L{Y.sub.n.sub.min.sub.i, . . . , Y.sub.n.sub.max.sub.i}.

[0157] At step S23 channel estimation and phase noise estimation is performed based on the symbols extracted as previously described.

[0158] At step S24 the symbols Y.sub.1, . . . , Y.sub.M outputted by the DFT module 4.3 are processed by the equalization module 4.6 to obtain the estimated symbols ({circumflex over (X)}.sub.1, . . . , {circumflex over (X)}.sub.M)={circumflex over (X)} of the symbols X.sub.1 , . . . , X.sub.M respectively transmitted through the subcarriers S.sub.1, . . . , S.sub.M. This is done according to the channel estimation and to the phase noise estimation computed by the channel and phase noise estimation module 4.5 as previously described. The estimated symbols ({circumflex over (X)}.sub.1, . . . , {circumflex over (X)}.sub.M) are then processed through the subcarrier demapping module 4.7 and the parallel to serial module 4.8 to retrieve the N′ symbols previous processed by the transmitter 1.1.

CIRCULAR PILOT SEQUENCES FOR JOINT CHANNEL AND PHASE NOISE ESTIMATION

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