Forward converter having a primary-side current sense circuit

Abstract

A load control device for controlling the amount of power delivered to an electrical load (e.g., an LED light source) includes first and second semiconductor switches, a transformer, a capacitor, a controller, and a current sense circuit operable to receive a sense voltage representative of a primary current conducted through a primary winding of the transformer. The primary winding is coupled in series with a semiconductor switch, while a secondary winding is adapted to be operatively coupled to the load. The capacitor is electrically coupled between the junction of the first and second semiconductor switches and the primary winding. The current sense circuit receives a sense voltage and averages the sense voltage when the first semiconductor switch is conductive, so as to generate a load current control signal that is representative of a real component of a load current conducted through the load.

Claims

1. A load control device for controlling an amount of power delivered to an electrical load, the load control device comprising: an isolated forward converter configured to receive a bus voltage and to conduct a load current through the electrical load, the isolated forward converter comprising: a transformer comprising a primary winding configured to conduct a primary current and a secondary winding configured to supply current to the electrical load; first and second semiconductor switches electrically coupled in series, the first and second semiconductor switches configured to generate a primary voltage across the primary winding of the transformer to cause the transformer to transfer power to the secondary winding when either of the first and second semiconductor switches is conductive; and an energy-storage inductor operatively coupled in series with the secondary winding of the transformer, the energy-storage inductor comprising a partially-gapped magnetic core set; a current sense circuit configured to receive a sense voltage representative of a magnitude of the primary current conducted through the primary winding of the transformer, the current sense circuit configured to generate a load current signal that is representative of a real component of the primary current by averaging the sense voltage when either of the first and second semiconductor switches is conductive; and a controller configured to control the first and second semiconductor switches to generate the primary voltage across the primary winding of the transformer and control a load current conducted through the electrical load in response to the load current signal.

2. The load control device of claim 1, wherein the current sense circuit comprises an averaging circuit configured to generate the load current signal when either of the first and second semiconductor switches is conductive, the controller configured to control the current sense circuit to provide the sense voltage to the averaging circuit when either of the first and second semiconductor switches is conductive.

3. The load control device of claim 2, wherein the controller is configured to generate a first drive signal for periodically rendering the first semiconductor switch conductive for an on time, and a second drive signal for periodically rendering the second semiconductor switch conductive for the on time.

4. The load control device of claim 3, wherein the controller is configured to control the current sense circuit to provide the sense voltage to the averaging circuit for the on time plus an additional amount of time when a target amount of power to be delivered to the electrical load is less than a threshold amount.

5. The load control device of claim 4, wherein the additional amount of time is a function of the target amount of power.

6. The load control device of claim 4, wherein the additional amount of time increases linearly as the target amount of power decreases.

7. The load control device of claim 3, wherein the controller is configured to control the current sense circuit to provide the sense voltage to the averaging circuit for the on time when the target amount of power to be delivered to the electrical load is greater than a threshold amount.

8. The load control device of claim 2, wherein the current sense circuit comprises a third semiconductor switch configured to disconnect the sense voltage from the averaging circuit, the controller configured to control the third semiconductor switch to provide the sense voltage to the averaging circuit when either of the first and second semiconductor switches is conductive.

9. The load control device of claim 8, wherein the sense voltage is coupled to the averaging circuit through two series-connected resistors, the third semiconductor switch coupled between the junction of the two resistors and a circuit common to allow the sense voltage to be provided to the averaging circuit when the third semiconductor switch is non-conductive.

10. The load control device of claim 2, wherein the controller is configured to generate a control signal for causing the current sense circuit to provide the sense voltage to the averaging circuit, the controller further configured to sample the load current signal when the current sense circuit is being controlled to provide the sense voltage to the averaging circuit.

11. The load control device of claim 1, further comprising: a capacitor electrically coupled between the junction of the first and second semiconductor switches and the primary winding of the transformer to cause the primary voltage across the primary winding to have a positive polarity when the first semiconductor switch is conductive and a negative polarity when the second semiconductor switch is conductive.

12. The load control device of claim 1, wherein the isolated forward converter further comprises a sense resistor coupled in series with the primary winding of the transformer, the sense resistor configured to conduct the primary current and produce the sense voltage.

13. A light-emitting diode (LED) driver for controlling an intensity of an LED light source, the LED driver comprising: an isolated forward converter configured to receive a bus voltage and to conduct a load current through the LED light source, the isolated forward converter comprising: a transformer comprising a primary winding configured to conduct a primary current and a secondary winding configured to conduct the load current through the LED light source; first and second semiconductor switches electrically coupled in series, the first and second semiconductor switches configured to generate a primary voltage across the primary winding of the transformer to cause the transformer to transfer power to the secondary winding when either of the first and second semiconductor switches is conductive; and an energy-storage inductor operatively coupled in series with the secondary winding of the transformer, the energy-storage inductor comprising a partially-gapped magnetic core set; a current sense circuit configured to receive a sense voltage representative of a magnitude of the primary current conducted through the primary winding of the transformer, the current sense circuit configured to generate a load current signal that is representative of a real component of the primary current by averaging the sense voltage when either of the first and second semiconductor switches is conductive; and a controller configured to control the first and second semiconductor switches to generate the primary voltage across the primary winding of the transformer and control the intensity of the LED light source by controlling a magnitude of a load current conducted through the LED light source in response to the load current signal.

14. The LED driver of claim 13, wherein the controller is configured to render each of the first and second semiconductor switch conductive for an on time, and average the sense voltage for the on time plus an additional amount of time when a target amount of power to be delivered to the LED light source is less than a threshold amount.

15. The LED driver of claim 14, wherein the additional amount of time is a function of the target amount of power.

16. The LED driver of claim 14, wherein the additional amount of time increases linearly as the target amount of power decreases.

17. The LED driver of claim 14, wherein the controller is configured to cause the averaging circuit to average the sense voltage for the on time when the target amount of power to be delivered to the LED light source is greater than a threshold amount.

18. The LED driver of claim 13, wherein the current sense circuit comprises an averaging circuit configured to generate the load current signal when either of the first and second semiconductor switches is conductive, the current sense circuit further comprising a third semiconductor switch configured to disconnect the sense voltage from the averaging circuit, the controller configured to control the third semiconductor switch to provide the sense voltage to the averaging circuit when either of the first and second semiconductor switches is conductive.

19. The LED driver of claim 18, wherein the sense voltage is coupled to the averaging circuit through two series-connected resistors, the third semiconductor switch coupled between the junction of the two resistors and a circuit common to allow the sense voltage to be provided to the averaging circuit when the third semiconductor switch is non-conductive.

20. The LED driver of claim 13, further comprising: a capacitor electrically coupled between the junction of the first and second semiconductor switches and the primary winding of the transformer to cause the primary voltage across the primary winding to have a positive polarity when the first semiconductor switch is conductive and a negative polarity when the second semiconductor switch is conductive.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) FIG. 1 is a simplified block diagram of a light-emitting diode (LED) driver for controlling the intensity of an LED light source.

(2) FIG. 2 is a simplified schematic diagram of an isolated forward converter and a current sense circuit of an LED driver.

(3) FIG. 3 is an example diagram illustrating a magnetic core set of an energy-storage inductor of a forward converter.

(4) FIG. 4 shows example waveforms illustrating the operation of a forward converter and a current sense circuit when the intensity of an LED light source is near a high-end intensity.

(5) FIG. 5 shows example waveforms illustrating the operation of a forward converter and a current sense circuit when the intensity of an LED light source is near a low-end intensity.

(6) FIG. 6 is an example plot of a relationship between an offset time and a target intensity of an LED driver.

(7) FIG. 7 is a simplified flowchart of a control procedure executed periodically by a controller of an LED driver.

(8) FIG. 8 is an example plot of a relationship between the offset time and the target intensity of an LED driver.

(9) FIG. 9 shows an example waveform of a load current conducted through an LED light source when a target current of an LED driver is at a steady-state value.

(10) FIG. 10 shows an example waveform of the load current conducted through the LED light source when the target current of the LED driver is being increased with respect to time.

(11) FIG. 11 shows example waveforms of a ramp signal of an LED driver and a load current conducted through an LED light source when the ramp signal is added to a target current.

DETAILED DESCRIPTION

(12) FIG. 1 is a simplified block diagram of a light-emitting diode (LED) driver 100 for controlling the intensity of an LED light source 102 (e.g., an LED light engine). The LED light source 102 is shown as a plurality of LEDs connected in series but may comprise a single LED or a plurality of LEDs connected in parallel or a suitable combination thereof, depending on the particular lighting system. In addition, the LED light source 102 may alternatively comprise one or more organic light-emitting diodes (OLEDs). The LED driver 100 comprises a hot terminal H and a neutral terminal that are adapted to be coupled to an alternating-current (AC) power source (not shown).

(13) The LED driver 100 comprises a radio-frequency (RFI) filter circuit 110 for minimizing the noise provided on the AC mains and a rectifier circuit 120 for generating a rectified voltage V.sub.RECT. The LED driver 100 further comprises a boost converter 130, which receives the rectified voltage V.sub.RECT and generates a boosted direct-current (DC) bus voltage V.sub.BUS across a bus capacitor CBUS. The boost converter 130 may alternatively comprise any suitable power converter circuit for generating an appropriate bus voltage, such as, for example, a flyback converter, a single-ended primary-inductor converter (SEPIC), a Ćuk converter, or other suitable power converter circuit. The boost converter 120 may also operate as a power factor correction (PFC) circuit to adjust the power factor of the LED driver 100 toward a power factor of one. The LED driver 100 also comprises an isolated, half-bridge forward converter 140, which receives the bus voltage V.sub.BUS and controls the amount of power delivered to the LED light source 102 so as to control the intensity of the LED light source between a low-end (i.e., minimum) intensity L.sub.LE (e.g., approximately 1-5%) and a high-end (i.e., maximum) intensity L.sub.HE (e.g., approximately 100%).

(14) The LED driver 100 further comprises a control circuit, e.g., a controller 150, for controlling the operation of the boost converter 130 and the forward converter 140. The controller 150 may comprise, for example, a digital controller or any other suitable processing device, such as, for example, a microcontroller, a programmable logic device (PLD), a microprocessor, an application specific integrated circuit (ASIC), or a field-programmable gate array (FPGA). The controller 150 generates a bus voltage control signal V.sub.BUS-CNTL, which is provided to the boost converter 130 for adjusting the magnitude of the bus voltage V.sub.BUS. The controller 150 receives from the boost converter 130 a bus voltage feedback control signals V.sub.BUS-FB, which is representative of the magnitude of the bus voltage V.sub.BUS.

(15) The controller 150 also generates drive control signals V.sub.DRIVE1, V.sub.DRIVE2, which are provided to the forward converter 140 for adjusting the magnitude of a load voltage V.sub.LOAD generated across the LED light source 102 and the magnitude of a load current I.sub.LOAD conducted through the LED light source to thus control the intensity of the LED light source to a target intensity L.sub.TRGT. The LED driver 100 further comprises a current sense circuit 160, which is responsive to a sense voltage V.sub.SENSE that is generated by the forward converter 140 and is representative of the magnitude of the load current I.sub.LOAD. The current sense circuit 160 is responsive to a signal-chopper control signal V.sub.CHOP (which is received from the controller 150) and generates a load current feedback signal V.sub.I-LOAD (which is a DC voltage representative of the magnitude of the load current I.sub.LOAD). The controller 150 receives the load current feedback signal V.sub.I-LOAD from the current sense circuit 160 and controls the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 to adjust the magnitude of the load current I.sub.LOAD to a target load current I.sub.TRGT to thus control the intensity of the LED light source 102 to the target intensity L.sub.TRGT. The target load current I.sub.TRGT may be adjusted between a minimum load current I.sub.MIN and a maximum load current I.sub.MAX.

(16) The controller 150 is coupled to a memory 170 for storing the operational characteristics of the LED driver 100 (e.g., the target intensity L.sub.TRGT, the low-end intensity L.sub.LE, the high-end intensity L.sub.HE, etc.). The LED driver 100 may also comprise a communication circuit 180, which may be coupled to, for example, a wired communication link or a wireless communication link, such as a radio-frequency (RF) communication link or an infrared (IR) communication link. The controller 150 may be operable to update the target intensity L.sub.TRGT of the LED light source 102 or the operational characteristics stored in the memory 170 in response to digital messages received via the communication circuit 180. Alternatively, the LED driver 100 could be operable to receive a phase-control signal from a dimmer switch for determining the target intensity L.sub.TRGT for the LED light source 102. The LED driver 100 further comprises a power supply 190, which receives the rectified voltage V.sub.RECT and generates a direct-current (DC) supply voltage V.sub.CC for powering the circuitry of the LED driver.

(17) FIG. 2 is a simplified schematic diagram of a forward converter 240 and a current sense circuit 260, e.g., the forward converter 140 and the current sense circuit 160 of the LED driver 100 shown in FIG. 1. The forward converter 240 comprises a half-bridge inverter circuit having two field effect transistors (FETs) Q210, Q212 for generating a high-frequency inverter voltage V.sub.INV from the bus voltage V.sub.BUS. The FETs Q210, Q212 are rendered conductive and non-conductive in response to the drive control signals V.sub.DRIVE1, V.sub.DRIVE2, which are received from a controller (e.g., the controller 150). The drive control signals V.sub.DRIVE1, V.sub.DRIVE2 are coupled to the gates of the respective FETs Q210, Q212 via a gate drive circuit 214 (e.g., part number L6382DTR, manufactured by ST Microelectronics). The controller generates the inverter voltage V.sub.INV at a constant operating frequency f.sub.OP (e.g., approximately 60-65 kHz) and thus a constant operating period T.sub.OP. However, the operating frequency f.sub.OP may be adjusted under certain operating conditions. The controller adjusts the duty cycle DC of the inverter voltage V.sub.INV to adjust the magnitude of the load current I.sub.LOAD and thus the intensity of an LED light source 202. The forward converter 240 may be characterized by a turn-on time T.sub.TURN-ON from when the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 are driven high until the respective FET Q210, Q212 is rendered conductive, and a turn-off time T.sub.TURN-OFF from when the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 are driven low until the respective FET Q210, Q212 is rendered non-conductive.

(18) The inverter voltage V.sub.INV is coupled to the primary winding of a transformer 220 through a DC-blocking capacitor C216 (e.g., having a capacitance of approximately 0.047 μF), such that a primary voltage V.sub.PRI is generated across the primary winding. The transformer 220 is characterized by a turns ratio n.sub.TURNS (i.e., N.sub.1/N.sub.2) of approximately 115:29. The sense voltage V.sub.SENSE is generated across a sense resistor R222, which is coupled series with the primary winding of the transformer 220. The FETs Q210, Q212 and the primary winding of the transformer 220 are characterized by parasitic capacitances C.sub.P1, C.sub.P2, C.sub.P3.

(19) The secondary winding of the transformer 220 generates a secondary voltage, which is coupled to the AC terminals of a full-wave diode rectifier bridge 224 for rectifying the secondary voltage generated across the secondary winding. The positive DC terminal of the rectifier bridge 224 is coupled to the LED light source 202 through an output energy-storage inductor L226 (e.g., having an inductance of approximately 10 mH), such that the load voltage V.sub.LOAD is generated across an output capacitor C228 (e.g., having a capacitance of approximately 3 μF).

(20) FIG. 3 is an example diagram illustrating a magnetic core set 290 of an energy-storage inductor (e.g., the output energy-storage inductor L226 of the forward converter 240 shown in FIG. 2). The magnetic core set 290 comprises two E-cores 292A, 292B, and may comprise part number PC40EE16-Z, manufactured by TDK Corporation. The E-cores 292A have respective outer legs 294A, 294B and inner legs 296A, 296B. Each inner leg 296A, 296B may have a width w.sub.LEG (e.g., approximately 4 mm). The inner leg 296A of the first E-core 292A has a partial gap 298A (i.e., the magnetic core set 290 is partially gapped) such that the inner legs 296A, 296B are spaced apart by a gap distance d.sub.GAP (e.g., approximately 0.5 mm). The partial gap 298A may extend for a gap width w.sub.GAP, e.g., approximately 2.8 mm, such that the gap extends for approximately 70% of the leg width w.sub.LEG of the inner leg 296A. Alternatively, both of the inner legs 296A, 296B could comprise partial gaps. The partially-gapped magnetic core set 290 shown in FIG. 3 allows the output energy-storage inductor L226 of the forward converter 240 shown in FIG. 2 to maintain continuous current at low load conditions (e.g., near the low-end intensity L.sub.LE).

(21) FIG. 4 shows example waveforms illustrating the operation of a forward converter and a current sense circuit, e.g., the forward converter 240 and the current sense circuit 260 shown in FIG. 2. The controller drives the respective drive control signals V.sub.DRIVE1, V.sub.DRIVE2 high to approximately the supply voltage V.sub.CC to render the respective FETs Q210, Q212 conductive for an on-time T.sub.ON at different times (i.e., the FETs Q210, Q212 are not conductive at the same time). When the high-side FET Q210 is conductive, the primary winding of the transformer 220 conducts a primary current I.sub.PRI to circuit common through the capacitor C216 and sense resistor R222. Immediately after the high-side FET Q210 is rendered conductive (at time t.sub.1 in FIG. 4), the primary current I.sub.PRI conducts a short high-magnitude pulse of current due to the parasitic capacitance C.sub.P3 of the transformer 220 as shown in FIG. 4. While the high-side FET Q210 is conductive, the capacitor C216 charges such that a voltage having a magnitude of approximately half of the magnitude of the bus voltage V.sub.BUS is developed across the capacitor. Accordingly, the magnitude of the primary voltage V.sub.PRI across the primary winding of the transformer 220 is approximately equal to approximately half of the magnitude of the bus voltage V.sub.BUS. When the low-side FET Q212 is conductive, the primary winding of the transformer 220 conducts the primary current I.sub.PRI in an opposite direction and the capacitor C216 is coupled across the primary winding, such that the primary voltage V.sub.PRI has a negative polarity with a magnitude equal to approximately half of the magnitude of the bus voltage V.sub.BUS.

(22) When either of the high-side and low-side FETs Q210, Q212 are conductive, the magnitude of an output inductor current I.sub.L conducted by the output inductor L226 and the magnitude of the load voltage V.sub.LOAD across the LED light source 202 both increase with respect to time. The magnitude of the primary current I.sub.PRI also increases with respect to time while the FETs Q210, Q212 are conductive (after the initial current spike). When the FETs Q210, Q212 are non-conductive, the output inductor current I.sub.L and the load voltage V.sub.LOAD both decrease in magnitude with respective to time. The output inductor current I.sub.L is characterized by a peak magnitude I.sub.L-PK and an average magnitude I.sub.L-AVG as shown in FIG. 4. The controller increases and decreases the on times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 (and the duty cycle DC of the inverter voltage V.sub.INV) to respectively increase and decrease the average magnitude I.sub.L-AVG of the output inductor current I.sub.L and thus respectively increase and decrease the intensity of the LED light source 102.

(23) When the FETs Q210, Q212 are rendered non-conductive, the magnitude of the primary current I.sub.PRI drops toward zero amps (e.g., as shown at time t.sub.2 in FIG. 4 when the high-side FET Q210 is rendered non-conductive). However, current may continue to flow through the primary winding of the transformer 220 due to the magnetizing inductance L.sub.MAG of the transformer (which conducts a magnetizing current I.sub.MAG). In addition, when the target intensity L.sub.TRGT of the LED light source 102 is near the low-end intensity L.sub.LE, the magnitude of the primary current I.sub.PRI oscillates after either of the FETs Q210, Q212 is rendered non-conductive due to the parasitic capacitances C.sub.P1, C.sub.P2 of the FETs, the parasitic capacitance C.sub.P3 of the primary winding of the transformer 220, and any other parasitic capacitances of the circuit, such as, parasitic capacitances of the printed circuit board on which the forward converter 240 is mounted.

(24) The real component of the primary current I.sub.PRI is representative of the magnitude of the secondary current I.sub.SEC and thus the intensity of the LED light source 202. However, the magnetizing current I.sub.MAG (i.e., the reactive component of the primary current I.sub.PRI) also flows through the sense resistor R222. The magnetizing current I.sub.MAG changes from negative to positive polarity when the high-side FET Q210 is conductive, changes from positive to negative polarity when the low-side FET Q212 is conductive, and remains constant when the magnitude of the primary voltage V.sub.PRI is zero volts, as shown in FIG. 4. The magnetizing current I.sub.MAG has a maximum magnitude defined by the following equation:

(25) $\begin{array}{cc}{I}_{\mathrm{MAG}-\mathrm{MAX}}=\frac{V_{\mathrm{BUS}}.Math.T_{HC}}{4.Math.L_{MAG}},& \left(\mathrm{Equation}1\right)\end{array}$
where T.sub.HC is the half-cycle period of the inverter voltage V.sub.INV, i.e., T.sub.HC=T.sub.OP/2. As shown in FIG. 4, the areas 250, 252 are approximately equal, such that the average value of the magnitude of the magnetizing current I.sub.MAG when the magnitude of the primary voltage V.sub.PRI is greater than approximately zero volts.

(26) The current sense circuit 260 averages the primary current I.sub.PRI during the positive cycles of the inverter voltage V.sub.INV, i.e., when the high-side FET Q210 is conductive. The load current feedback signal V.sub.I-LOAD generated by the current sense circuit 260 has a DC magnitude that is the average value of the primary current I.sub.PRI when the high-side FET Q210 is conductive. Because the average value of the magnitude of the magnetizing current I.sub.MAG is approximately zero when the high-side FET Q210 is conductive, the load current feedback signal V.sub.I-LOAD generated by the current sense circuit is representative of only the real component of the primary current I.sub.PRI.

(27) The current sense circuit 260 comprises an averaging circuit for producing the load current feedback signal V.sub.I-LOAD. The averaging circuit may comprise a low-pass filter having a capacitor C230 (e.g., having a capacitance of approximately 0.066 uF) and a resistor R232 (e.g., having a resistance of approximately 3.32 kΩ). The low-pass filter receives the sense voltage V.sub.SENSE via a resistor R234 (e.g., having resistances of approximately 1 kΩ). The current sense circuit 160 further comprises a transistor Q236 (e.g., a FET as shown in FIG. 2) coupled between the junction of the resistors R232, R234 and circuit common. The gate of the transistor Q236 is coupled to circuit common through a resistor R238 (e.g., having a resistance of approximately 22 kΩ) and receives the signal-chopper control signal V.sub.CHOP from the controller.

(28) When the high-side FET Q210 is rendered conductive, the controller drives the signal-chopper control signal V.sub.CHOP low toward circuit common to render the transistor Q236 non-conductive for a signal-chopper time T.sub.CHOP, which is approximately equal to the on time T.sub.ON of the high-side FET Q210 as shown in FIG. 4. The capacitor C230 is able to charge from the sense voltage V.sub.SENSE through the resistors R232, R234 while the signal-chopper control signal V.sub.CHOP is low, such that the magnitude of the load current feedback signal V.sub.I-LOAD is the average value of the primary current I.sub.PRI and is thus representative of the real component of the primary current during the time when the high-side FET Q210 is conductive. When the high-side FET Q210 is not conductive, the controller 150 drives the signal-chopper control signal V.sub.CHOP high to render the transistor Q236 non-conductive. Accordingly, the controller is able to accurately determine the average magnitude of the load current I.sub.LOAD from the magnitude of the load current feedback signal V.sub.I-LOAD since the effects of the magnetizing current I.sub.MAG and the oscillations of the primary current I.sub.PRI on the magnitude of the load current feedback signal V.sub.I-LOAD are reduced or eliminated completely.

(29) As the target intensity L.sub.TRGT of the LED light source 202 is decreased toward the low-end intensity L.sub.LE (and the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 get smaller), the parasitics of the forward converter 140 (i.e., the parasitic capacitances C.sub.P1, C.sub.P2 of the FETs, the parasitic capacitance C.sub.P3 of the primary winding of the transformer 220, and other parasitic capacitances of the circuit) can cause the magnitude of the primary voltage V.sub.PRI to slowly decrease towards zero volts after the FETs Q210, Q212 are rendered non-conductive.

(30) FIG. 5 shows example waveforms illustrating the operation of a forward converter and a current sense circuit (e.g., the forward converter 240 and the current sense circuit 260) when the target intensity L.sub.TRGT is near the low-end intensity L.sub.LE. The gradual drop off in the magnitude of the primary voltage V.sub.PRI allows the primary winding to continue to conduct the primary current I.sub.PRI, such that the transformer 220 continues to deliver power to the secondary winding after the FETs Q210, Q212 are rendered non-conductive as shown in FIG. 5. In addition, the magnetizing current I.sub.MAG continues to increase in magnitude. Accordingly, the controller 150 increases the signal-chopper time T.sub.CHOP (during which the signal-chopper control signal V.sub.CHOP is low) by an offset time T.sub.OS when the target intensity L.sub.TRGT of the LED light source 202 is near the low-end intensity L.sub.LE. The controller may adjust the value of the offset time T.sub.OS as a function of the target intensity L.sub.TRGT of the LED light source 202 as shown in FIG. 6. For example, the controller may adjust the value of the offset time T.sub.OS linearly with respect to the target intensity L.sub.TRGT when the target intensity L.sub.TRGT is below a threshold intensity L.sub.TH (e.g., approximately 10%), as shown in FIG. 5.

(31) FIG. 7 is a simplified flowchart of a control procedure 300 executed periodically by a controller (e.g., the controller 150 of the LED driver 100 shown in FIG. 1 and/or the controller controlling the forward converter 240 and the current sense circuit 260 shown in FIG. 2). The controller may execute the control procedure 300, for example, at the operating period T.sub.OP of the inverter voltage V.sub.INV of the forward converter 240. First, the controller determines the appropriate on time T.sub.ON for the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 in response to the target intensity L.sub.TRGT and the load current feedback signal V.sub.I-LOAD at step 310. If the controller should presently control the high-side FET Q210 at step 312, the controller drives the first drive control signal V.sub.DRIVE1 high to approximately the supply voltage V.sub.CC for the on-time T.sub.ON at step 314. If the target intensity L.sub.TRGT is greater than or equal to the threshold intensity L.sub.TH at step 316, the controller 150 sets the signal-chopper time T.sub.CHOP equal to the on-time T.sub.ON at step 318. If the target intensity L.sub.TRGT is less than the threshold intensity L.sub.TH at step 316, the controller determines the offset time T.sub.OS in response to the target intensity L.sub.TRGT at step 320 (e.g., using the relationship shown in FIG. 6), and sets the signal-chopper time T.sub.CHOP equal to the sum of the on-time T.sub.ON and the offset time T.sub.OS at step 322.

(32) Next, the controller drives the signal-chopper control signal V.sub.CHOP low towards circuit common for the signal-chopper time T.sub.CHOP at step 324. The controller then samples the averaged load current feedback signal V.sub.I-LOAD at step 326 and calculates the magnitude of the load current I.sub.LOAD using the sampled value at step 328, for example, using the following equation:

(33) $\begin{array}{cc}{I}_{\mathrm{LOAD}}=\frac{n_{\mathrm{TURNS}}.Math.V_{I-LOAD}.Math.T_{\mathrm{HC}}}{R_{SENSE}.Math.\left(T_{C\mathrm{HOP}}-T_{\mathrm{DELAY}}\right)},& \left(\mathrm{Equation}2\right)\end{array}$
where T.sub.DELAY is the total delay time due to the turn-on time and the turn-off time of the FETs Q210, Q212, e.g., T.sub.DELAY=T.sub.TURN-ON−T.sub.TURN-OFF, which may be equal to approximately 200 μsec. Finally, the control procedure 300 exits after the magnitude of the load current I.sub.LOAD has been calculated. If the controller should presently control the low-side FET Q210 at step 312, the controller drives the second drive control signal V.sub.DRIVE2 high to approximately the supply voltage V.sub.CC for the on-time T.sub.ON at step 330, and the control procedure 300 exits without the controller driving the signal-chopper control signal V.sub.CHOP low.

(34) Alternatively, the controller can use a different relationship to determine the offset time T.sub.OS throughout the entire dimming range of the LED light source (i.e., from the low-end intensity L.sub.LE to the high-end intensity Um), as shown in FIG. 8. For example, the controller could use the following equation:

(35) $\begin{array}{cc}{T}_{\mathrm{OS}}=\frac{\frac{V_{\mathrm{BUS}}}{4}.Math.C_{\mathrm{PARASITIC}}}{\begin{array}{c}\frac{T_{\mathrm{ON}}+T_{\mathrm{OS}-\mathrm{PREV}}}{T_{\mathrm{HC}}}.Math.I_{\mathrm{MAG}-\mathrm{MAX}}+\\ \frac{K_{\mathrm{RIPPLE}}}{n_{\mathrm{TURNS}}}.Math.I_{\mathrm{LOAD}}\end{array}},& \left(\mathrm{Equation}3\right)\end{array}$
where T.sub.OS-PREV is the previous value of the offset time, K.sub.RIPPLE is the dynamic ripple ratio of the output inductor current I.sub.L (which is a function of the load current I.sub.LOAD) i.e.,
K.sub.RIPPLE=I.sub.L-PK/I.sub.L-AVG,  (Equation 4)
and C.sub.PARASITIC is the total parasitic capacitance between the junction of the FETs Q210, Q212 and circuit common.

(36) As previously mentioned, the controller increases and decreases the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 for controlling the FETs Q210, Q212 of the forward converter 140 to respectively increase and decrease the intensity of the LED light source. Due to hardware limitations, the controller may be operable to adjust the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 by a minimum time step T.sub.STEP, which results in a corresponding step I.sub.STEP in the load current I.sub.LOAD. Near the high-end intensity L.sub.HE, this step I.sub.STEP in the load current I.sub.LOAD may be rather large (e.g., approximately 70 mA). Since it is desirable to adjust the load current I.sub.LOAD by smaller amounts, the controller is operable to “dither” the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2, e.g., change the on-times between two values that result in the magnitude of the load current being controlled to DC currents on either side of the target current I.sub.TRGT.

(37) FIG. 9 shows an example waveform of a load current conducted through an LED light source (e.g., the load current I.sub.LOAD). For example, the load current I.sub.LOAD shown in FIG. 9 may be conducted through the LED light source when the target current I.sub.TRGT is at a steady-state value of approximately 390 mA. A controller (e.g., the controller 150 of the LED driver 100 shown in FIG. 1 and/or the controller controlling the forward converter 240 and the current sense circuit 260 shown in FIG. 2) may control a forward converter (e.g., the forward converter 140, 240) to conduct the load current I.sub.LOAD shown in FIG. 9 through the LED light source. The controller adjusts the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 to control the magnitude of the load current I.sub.LOAD to between two DC currents I.sub.L-1, I.sub.L-2 that are separated by the step I.sub.STEP (e.g., approximately 350 mA and 420 mA, respectively). The load current I.sub.LOAD is characterized by a dithering frequency f.sub.DITHER (e.g., approximately 2 kHz) and a dithering period T.sub.DITHER as shown in FIG. 9. For example, a duty cycle DC.sub.DITHER of the load current I.sub.LOAD may be approximately 57%, such that the average magnitude of the load current I.sub.LOAD is approximately equal to the target current I.sub.TRGT (e.g., 390 mA for the example of FIG. 9).

(38) FIG. 10 shows an example waveform of the load current I.sub.LOAD when the target current I.sub.TRGT is being increased with respect to time. As shown in FIG. 10, the controller 150 is able to adjust the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 to control the magnitude of the load current I.sub.LOAD to between two DC currents I.sub.L-1, I.sub.L-2 that are separated by the step I.sub.STEP. The duty cycle DC.sub.DITHER of the load current I.sub.LOAD increases as the target current I.sub.TRGT increases. At some point, the controller is able to control the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 to achieve the desired target current I.sub.TRGT without dithering the on-times, thus resulting in a constant section 400 of the load current I.sub.LOAD. As the target current I.sub.TRGT continues to increase after the constant section 400, the controller is able to control the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2 to dither the magnitude of the load current I.sub.LOAD between the DC current I.sub.L-2 and a larger DC current I.sub.L-3.

(39) However, the constant section 400 of the load current I.sub.LOAD as shown in FIG. 10 may cause the human eye to detect a visible step in the adjustment of the intensity of the LED light source. Therefore, when the controller is actively adjusting the intensity of the LED light source, the controller is operable to add a periodic supplemental signal (e.g., a ramp signal I.sub.RAMP or sawtooth waveform) to the target current I.sub.TRGT. FIG. 11 shows example waveforms of the ramp signal I.sub.RAMP and the resulting load current I.sub.LOAD when the ramp signal is added to the target current I.sub.TRGT. Note that these waveforms are not to scale and the ramp signal I.sub.RAMP is a digital waveform. The ramp signal I.sub.RAMP is characterized by a ramp frequency f.sub.RAMP (e.g., approximately 238 Hz) and a ramp period TRAMP. The ramp signal I.sub.RAMP may have, for example, a maximum ramp signal magnitude I.sub.RAMP-MAX of approximately 150 mA. The ramp signal I.sub.RAMP may increase with respect to time in, for example, approximately 35 steps across the length of the ramp period TRAMP. When the controller adds the ramp signal I.sub.RAMP to the target current I.sub.TRGT to control the on-times T.sub.ON of the drive control signals V.sub.DRIVE1, V.sub.DRIVE2, the resulting load current I.sub.LOAD has a varying magnitude as shown in FIG. 11. As a result, the perception to the human eye of the visible steps in the intensity of the LED light source as the controller is actively adjusting the intensity of the LED light source is reduced.

(40) When the target current I.sub.TRGT returns to a steady-state value, the controller may stop adding the ramp signal I.sub.RAMP to the target current I.sub.TRGT. For example, the controller may decrease the magnitude of the ramp signal I.sub.RAMP from the maximum ramp signal magnitude I.sub.RAMP-MAX to zero across a period of time after the target current I.sub.TRGT has reached a steady-state value.

(41) While FIG. 11 shows the ramp signal I.sub.RAMP (i.e., a sawtooth waveform) that is added to the target current I.sub.TRGT, other periodic waveforms could be used.

(42) Although the present disclosure has been described in relation to particular embodiments thereof, many other variations and modifications and other uses will become apparent to those skilled in the art. It is preferred, therefore, that the present disclosure be limited not by the specific disclosure herein, but only by the appended claims.