Abstract
A filter circuit comprising: a signal path for carrying a signal from an input to an output; the signal path comprising a first reactive component; a first node on the signal path; a first series resonant circuit comprising at least a second reactive component in series with a third reactive component, the first series resonant circuit being connected between the first node and a ground; an active circuit; the active circuit comprising a voltage controlled current source (VCCS) arranged to change the current flow through the second reactive component in dependence on a voltage sensed (or measured) on the signal path. The first series resonant circuit forms a single harmonic trap with a notch frequency defined by the component values of its reactive components. The effectiveness of the series resonant circuit is dependent upon the strength with which it draws current from the signal path at its resonant frequency.
Claims
1. A filter circuit comprising: a signal path for carrying a signal from an input to an output; the signal path comprising a first reactive component; a first node on the signal path; a first series resonant circuit comprising at least a second reactive component in series with a third reactive component, the first series resonant circuit being connected between the first node and a ground; an active circuit; the active circuit comprising a voltage controlled current source arranged to change the current flow through the second reactive component in dependence on a voltage sensed on the signal path.
2. A circuit as claimed in claim 1, wherein the second reactive component is an inductive component.
3. A circuit as claimed in claim 1, wherein the first reactive component is a capacitive component.
4. A circuit as claimed in claim 1, wherein the active circuit is arranged to change the current flow through the second reactive component in dependence on a voltage difference sensed across the first reactive component.
5. A circuit as claimed in claim 1, further comprising: a second node on the signal path; a second series resonant circuit comprising at least a fourth reactive component in series with a fifth reactive component, the second resonant circuit being connected between the second node and said ground; and an active circuit comprising a voltage controlled current source arranged to change the current flow through the fourth reactive component in dependence on a voltage sensed on the signal path.
6. A circuit as claimed in claim 5, wherein the fourth reactive component is an inductive component.
7. A circuit as claimed in any claim 5, wherein the signal path comprises at least one reactive component between the first node and the second node.
8. A circuit as claimed in claim 7, wherein the active circuit is arranged to change the current flow through the fourth reactive component in dependence on the voltage difference across the reactive component between the first node and the second node.
9. A circuit as claimed in claim 5, wherein the signal path further comprises a sixth reactive component and a seventh reactive component in series with the first reactive component; and wherein the first node is located between the first reactive component and the sixth reactive component and the second node is located between the sixth reactive component and the seventh reactive component.
10. A circuit as claimed in claim 1, further comprising a variable capacitor or capacitor bank connected between ground and a node between the second reactive component and the third reactive component.
11. A circuit as claimed in claim 1, the active circuit comprises: a current source arranged to draw current through a first amplifying element and thereby through the second reactive component, the first amplifying element being driven by the second node.
12. A circuit as claimed in claim 11, wherein the current source of the active circuit is arranged to draw current through a second amplifying element and thereby through the fourth reactive component, the second amplifying element being driven by the first node.
13. A circuit as claimed in claim 12, wherein the active circuit is a differential common-source amplifier.
14. A differential circuit having a first positive arm and a second negative arm; wherein the first positive arm is a circuit according to claim 1, wherein the second negative arm is a circuit according to claim 1, and wherein the first series resonant circuit of the first positive arm and the first series resonant circuit of the second negative arm are connected together to form an AC ground.
15. A differential circuit as claimed in claim 14, wherein the first positive arm is connected to the second positive arm by a centre-tapped inductor with the AC ground formed at its centre tap, and wherein the centre-tapped inductor forms part of the first series resonant circuits of the first positive arm and the second negative arm.
16. A method of filtering a signal comprising: passing the signal along a signal path from an input to an output through a first reactive component; from a first node on the signal path, shorting the signal to ground through a first series resonant circuit at the resonant frequency of the first series resonant circuit, the first series resonant circuit comprising a second reactive component in series with a third reactive component; and using an active circuit comprising a voltage controlled current source to draw current through the second reactive component, the active circuit changing the current flow through the second reactive component in dependence on a voltage sensed on the signal path.
Description
[0045] Certain preferred embodiments of the invention will now be described, by way of example only, and with reference to the accompanying drawings in which:
[0046] FIG. 1 illustrates a typical direct sampling receiver front-end;
[0047] FIG. 2 illustrates a fifth order high pass filter;
[0048] FIG. 3 illustrates a series resonant circuit;
[0049] FIG. 4 illustrates a single-harmonic trap filter arrangement;
[0050] FIGS. 5a and 5b show embodiments of a single-harmonic trap filter with active circuit;
[0051] FIGS. 6a and 6b show embodiments of a single-harmonic trap filter with active circuit driven by a voltage difference on the signal path;
[0052] FIG. 7 shows a high pass filter with two series resonant circuits;
[0053] FIG. 8 is a graph showing the phase of signals at nodes A and B of FIG. 7;
[0054] FIG. 9 shows a differential form of the filter of FIG. 7;
[0055] FIG. 10 shows another differential filter with tunability;
[0056] FIG. 11 shows a differential filter using centre-tapped inductors;
[0057] FIG. 12 shows a differential filter with tenability, using centre-tapped inductors;
[0058] FIGS. 13a and 13b show how to connect active circuits to differential filters;
[0059] FIG. 14 shows the active circuits of FIGS. 13a and 13b in more detail;
[0060] FIG. 15 is a graph of forward and reverse transmission coefficients for a filter according to FIG. 14; and
[0061] FIG. 16 is a graph comparing the forward transmission coefficients for the filter of FIG. 14 against a basic single harmonic trap filter.
[0062] FIG. 1 depicts a typical direct sampling receiver front-end 100 for a wideband receiver operating for example in the 6 to 8.5 GHz band. Antenna 101 receives a RF signal and passes it to high-pass filter 102 which rejects signals below about 6 GHz, with a high rejection notch at around 5.1 to 5.8 GHz (although it will be appreciated that these numbers are provided purely by way of example). The output of high-pass filter 102 feeds to the input of low-noise amplifier 103 which provides gain for the signal of interest across the operating band of 6 to 8.5 GHz. The output of low-noise amplifier 103 is then fed to an analog-to-digital converter (ADC) 104 that finally digitises the RF signal.
[0063] FIG. 2. shows a basic fifth order high pass LC-ladder filter 200. The (forward) signal path runs from RF.sub.i to RF.sub.o. For example, an antenna may be connected to RF.sub.i and the filtered signal from RF may be fed to further processing circuits. The filter 200 has three capacitors C.sub.1, C.sub.2, C.sub.3 in the signal path and two inductors L.sub.1, L.sub.2 connecting the signal path to ground. The reactance values of C.sub.1, C.sub.2, C.sub.3, L.sub.1 and L.sub.2 determine the properties of the filter such as its order, cut-off frequency, stopband frequency, and stopband rejection.
[0064] FIG. 3 shows a basic series resonant circuit 300 connecting a node A to ground through an inductor L.sub.1 in series with a capacitor C.sub.1. At the resonant frequency (⅟(2π(L.sub.1C.sub.1).sup.½)) the inductor (X.sub.L) and capacitor (X.sub.C) reactances cancel out (i.e., X.sub.L-X.sub.C=0), so that the series resonant circuit 300 forms a short circuit to ground at that frequency (i.e., the circuit impedance is minimum and entirely resistive). Moreover, at resonance, the voltage across the capacitor (V.sub.C) is equal to, but in anti-phase to the voltage across the inductor (V.sub.L), and the circuit current is at maximum. Below the resonance, the circuit is capacitive (X.sub.C > X.sub.L), and above the resonance, the circuit is inductive (X.sub.L >X.sub.C).
[0065] FIG. 4 shows a basic single-harmonic trap (SHT) filter 400 having a signal path from RF.sub.i to RF and with the series resonant circuit 300 of FIG. 3 connected at node A at RF. The HPF function comprises capacitor C.sub.2 in the signal path and inductor L.sub.1. Capacitor C.sub.2 is the only reactive component in the signal path, i.e., between the signal input, RF.sub.i and the signal output, RF. Therefore, in the passband C.sub.2 provides the direct impedance seen by the signal passing along the signal path. By appropriate selection of L.sub.1 and C.sub.1, the resonant frequency of these two reactive components can be used to form a notch in the transfer function (at the resonant frequency) by providing a short circuit/low impedance path to ground (at that resonant frequency). With C.sub.1 resonating with L.sub.1 (in the stopband), said filter is a SHT HPF. Frequency components of the input signal at the resonant frequency at node A or output will see a short circuit to ground, and therefore current can flow to ground at this resonant frequency and the amplitude of the output signal at this resonant frequency will be greatly diminished. At other frequencies (either side of the resonant frequency of the series resonant circuit) the signal will see an impedance between the signal path and ground and therefore the signal at these frequencies will be significantly less reduced in amplitude and instead passes to the signal path output RF.sub.o.
[0066] FIGS. 5a and 5b show the use of an active circuit to further mitigate the unwanted signal (the interferer). The filters 500a and 500b in FIG. 5a and FIG. 5b have active circuits 501a and 501b (labelled “g.sub.m and g.sub.m*” in the figures) that sense the signal on the signal path upstream of the capacitor C.sub.2 (i.e. between RF.sub.i and C.sub.2). The active circuit 501a, 501b draws a constant dc current through the inductor L.sub.1, but varies the magnitude of this current based on the sensed voltage on the signal path upstream of C.sub.2. As the signal upstream of C.sub.2 contains both the wanted signal (the signal to be passed by the filter) and the interferer (the signal to be rejected by the filter), the active circuit 501a, 501b varies the current in part based on the magnitude of the interferer and applies it to the node between L.sub.1 and C.sub.1. As the magnitude of the current is thereby increased at the resonant frequency, the amount of unwanted signal that is tapped off from the signal path is increased. Thus, the effectiveness of the series resonant circuit is increased by the active circuit so that the overall removal of the unwanted signal (interferer) by the filter 500a, 500b is improved.
[0067] Care needs to be taken to ensure that the active circuits 501a, 501b operate in phase with the node at which the series resonant circuit is connected so as to enhance the signal removal, i.e. the filter effectiveness. This will depend on the phase of the signal at which the active circuits 501a, 501b sense or measure the voltage on the signal path and the order of the reactive components in the series resonant circuit. FIG. 5a shows the case where the signal is sensed upstream of a capacitor C.sub.2 and where the inductor L.sub.1 is connected directly to the signal path while the capacitor C.sub.1 connects to ground. The active circuit 501a is connected to the node between the inductor L.sub.1 and the capacitor C.sub.1. At resonance, there is a phase change of approximately 180 degrees across C.sub.2, and also a phase change of approximately 180 degrees across L.sub.1. Therefore the phase at the node where the active circuit 501a is connected is inphase with the node at which the active circuit senses the signal path. Thus, the active circuit 501a in FIG. 5a is a non-inverting active circuit (denoted by the ‘+’ at its input and the ‘+’ at its output) such that its output is non-inverted with respect to its input. FIG. 5b shows the opposite situation in which the capacitor C.sub.1 is connected directly to the signal path while the inductor L.sub.1 connects to ground. Here there is a 180-degree phase change between the node where the active circuit 501b is connected to the series resonant circuit and the node at which the active circuit senses the signal path. Therefore in FIG. 5b the active circuit 501b is an inverting active circuit (denoted by the ‘+’ at its input and the ‘-’ at its output, and the label g.sub.m*) such that its output is inverted with respect to its input to compensate for this phase change.
[0068] FIGS. 6a and 6b are similar to FIGS. 5a and 5b except that instead of the active circuits 601a, 601b merely sensing the voltage on the signal path it senses the voltage drop across a reactive component on the signal path (in this case the capacitor C.sub.2). As the capacitor C.sub.2 can be chosen to provide an impedance at the frequency of the unwanted (interferer) signal, the amplitude of the interferer will drop across the capacitor C.sub.2 while the amplitude of the desired signal does not drop (or at least drops significantly less). Thus, as the active circuits 601a, 601b are driven by a difference signal which is predominantly at the frequency of the interferer, the active circuits 601a, 601b control the current draw through the series resonant circuit in relation to the interferer without any significant component of other frequencies such as those of the desired signal. This makes for a more efficient active circuit, thus further reducing the amplitude of the unwanted signal on the signal path. As with FIGS. 5a and 5b, FIG. 6a shows a non-inverting active circuit 601a (g.sub.m) acting on a series resonant circuit with the inductor L.sub.1 connected directly to the signal path, while FIG. 6b shows an inverting active circuit 601b (g.sub.m*) acting on a series resonant circuit with the capacitor C.sub.1 connected directly to the signal path.
[0069] FIG. 7 shows a 5.sup.th order LC ladder high pass filter 700 with two series resonant circuits tapping off from the signal path. The signal path from RF.sub.i to RF.sub.o has three capacitive components C.sub.3, C.sub.2, C.sub.1 with a first node formed between the first two capacitive components (node A between C.sub.3 and C.sub.2) and a second node formed between the last two capacitive components (node B between C.sub.2 and C.sub.1). A first series resonant circuit 702 is connected between node A and ground and is formed from inductive component L.sub.2 and capacitive component C.sub.5. A second series resonant circuit 703 is connected between node B and ground and is formed from inductive component L.sub.1 and capacitive component C.sub.4. No active circuits are shown in this arrangement for simplicity, but they can be connected as shown in FIGS. 5a or 6a so as to increase the signal reduction at resonance as discussed above. It will be appreciated that this filter architecture can readily be extended to higher order filters by adding further inductive components and capacitive components as per the usual structure of LC ladder filters. Further series resonant circuits could also be added if desired, although it is not essential to provide a series resonant circuit at every intermediate node of the filter. Indeed, the architecture of FIG. 7 could have only one of the series resonant circuits (L.sub.2/C.sub.5 or L.sub.1/C.sub.4) and still provide an improved filtered signal at RF. It will also be appreciated that the first series resonant circuit 702 and the second series resonant circuit 703 may have different resonant frequencies so as to provide two notches in the filter output at RF or they may have the same resonant frequency to provide a single deep notch in the filter output at RF.
[0070] FIG. 8 shows the phase of signals at nodes A and B of the filter 700 of FIG. 7 across a range of frequencies. At node A, it can be seen that the phase of the signal undergoes an approximately 180-degree phase shift at around 5.3 GHz, this being the resonant frequency of the first series resonant circuit 702 (L.sub.2/C.sub.5) attached at node A. At node B, it can be seen that the phase of the signal undergoes an approximately 180-degree phase shift at around 4.8 GHz, this being the resonant frequency of the second series resonant circuit 703 (L.sub.1/C.sub.4) attached at node B. A further phase shift occurs at 5.3 GHz (i.e. the resonant frequency of the first series resonant circuit 702) due to the effect of the first series resonant circuit 702 at node A.
[0071] FIG. 9 shows a differential filter circuit 900, this being a differential form of the circuit 700 of FIG. 7. The differential circuit 900 has a positive arm 906 including a positive signal path from RF.sub.i,+ to RF.sub.o,+ and a negative arm 907 including a negative signal path from RF.sub.i,- to RF.sub.o,-. The positive arm 906 and the negative arm 907 are identical and are connected together via their respective series resonant circuits 902, 904 and 903, 905 to form a virtual ground 908 between them at the point of connection (i.e. symmetrically between the two signal paths). The virtual ground 908 may be connected to an analog ground if desired or it may be left as a floating ground. As can be seen in FIG. 9, the first series resonant circuit 902 of the positive signal arm 906 is connected to the first series resonant circuit 904 of the negative signal arm 907. Similarly, the second series resonant circuit 903 of the positive signal arm 906 is connected to the second series resonant circuit 905 of the negative signal arm 907. Again, no active circuits are shown here for simplicity, but they can readily be added as discussed below.
[0072] FIG. 10 shows another differential filter circuit 1000 similar to that of FIG. 9, but with the addition of variable capacitors 1009 connected in parallel with each of the capacitive elements C.sub.4, C.sub.5 of the first and second series resonant circuits 1002, 1003, 1004, 1005 (of both the positive and negative signal arms 1006, 1007). These variable capacitors 1009 provide for tuning of the first and second series resonant circuits 1002, 1003, 1004, 1005 so as to adjust the resonant frequencies and thereby adjust the frequency of the notch in the filter output RF.sub.o,+, RF.sub.o,-. The variable capacitors 1009 may be switchable capacitor banks in some examples, or may be continuously variable capacitors in other examples.
[0073] FIG. 11 is similar to FIG. 9, but shows a further improvement that can be made in differential embodiments. In this example, the differential filter 1100 has the capacitors C.sub.4, C.sub.5 of the series resonant circuits 1102, 1103, 1104, 1105 connected directly to the respective signal paths and combines the inductors L.sub.1, L.sub.2 of the series resonant circuits 1102, 1103, 1104, 1105 into centre-tapped inductors 1110. For example, the inductive element of the first series resonant circuit 1102 of the positive signal arm 1106 and the inductive element of the first series resonant circuit 1104 of the negative signal arm 1107 are combined into a single centre-tapped inductor 1110 with inductance of 2L.sub.2. Similarly, the inductors of the second resonant circuits 1103, 1105 (positive and negative) are combined into a single centre-tapped inductor 1110 with inductance 2L.sub.1. The use of centre-tapped inductors 1110 reduces the die area required for the circuit when fabricated on chip.
[0074] FIG. 12 shows a differential circuit 1200 with centre-tapped inductors 1210 like that of FIG. 11 and also with variable capacitors 1209 as in FIG. 10. Again, no active circuits are shown in FIGS. 11 or 12 for clarity but can easily be added as discussed below.
[0075] FIG. 13a shows the differential circuit 1300a similar to that of FIG. 9, but with active circuits 1301a, 1311a added. The active circuit 1301a on the positive signal arm 1306 of the differential circuit 1300 is connected with its input taken across the second capacitor C.sub.2 on the signal path. As described above in relation to FIGS. 6a and 6b, taking the difference signal across a reactive element of the signal path means that the active circuit 1301a is driven predominantly by the interferer signal as that experiences the greatest amplitude reduction across the reactive element. In FIG. 13a the active circuit 1301a of the positive signal arm 1306 is configured to have a differential output with one output driving the first series resonant circuit 1302 of the positive signal arm 1306 and the other output driving the second series resonant circuit 1303 of the positive signal arm 1306.
[0076] As the interferer degrades downstream, the second resonant circuit 1303 will be primarily affected. It is thus appropriate in this arrangement to have the first series resonant circuit 1302 formed by L.sub.2 and C.sub.5 driven by an inverting output (indicated by “-”) of the active circuit 1301a and the second series resonant circuit 1303 formed by L.sub.1 and C.sub.4 driven by a non-inverting output (indicated by “+”) of the active circuit 1301a. With this arrangement, the first series resonant circuit 1302 is driven in a feedback arrangement and the second series resonant circuit 1303 is driven in a feedforward arrangement. It may be noted that the feedback arrangement shown in FIG. 13a with the inverting output of active circuit 1301a actually adds a small amount of interferer signal back onto the signal path. However, with the active circuit 1301a designed for the feedforward interferer removal through the second series resonant circuit 1303, the amount added back on at the first series resonant circuit is insignificant. To remove signal from the first series resonant circuit 1302 as well as from the second series resonant circuit 1303, the active circuit 1301a would need to provide a second non-inverting output. However, it is more convenient from a circuit perspective to use a differential active circuit 1301a with one inverting output and one non-inverting output. The inverting (“-”) output from the active circuit 1301a could be connected to an ac ground or to Vdd in order to avoid adding unwanted signal back onto the first series resonant circuit 1302. However, connecting the inverting output to the first series resonant circuit 1302 has an insignificant impact on the performance of the whole circuit and there is a significant advantage in terms of load balancing by connecting the inverting and non-inverting outputs of the differential active circuit 1301a each to a series resonant circuit. The same arrangement is shown with the active circuit 1311a driving the first series resonant circuit 1304 and second series resonant circuit 1305 of the negative signal arm 1307. It will be appreciated that in other examples, if the phases are different, or if the active circuit 1301a, 1311a senses the signal on the signal path at other points along the signal path, the phases may be different. Also, more than one active circuit 1301a, 1311a may be used to drive the two series resonant circuits 1302, 1303 or 1304, 1305. In each case the phases should be considered, and the active circuit 1301a, 1311a appropriately chosen so as to drive the series resonant circuits 1302, 1303, 1304, 1305 appropriately.
[0077] FIG. 13b shows the differential circuit 1300b similar to that of FIG. 13a, but with active circuits 1312b, 1313b added. The active circuit 1312b on the positive signal arm 1306 of the differential circuit 1300b is connected with its input taken across the second capacitor C.sub.3 on the signal path. As described above in relation to FIGS. 6a and 6b, taking the difference signal across a reactive element of the signal path means that the active circuit 1312b is driven predominantly by the interferer signal as that experiences the greatest amplitude reduction across the reactive element. The active circuit 1312b of the positive signal arm 1306 is configured to have a differential output with one output driving the first series resonant circuit 1302 of the positive signal arm 1306 and the other output not connected (connected to ground). The same arrangement is shown on the negative signal arm 1307 with the active circuit 1313b configured to have a differential output with one output driving the first series resonant circuit 1304 of the negative signal arm 1307 and the other output not connected (connected to ground).
[0078] As in FIG. 13a, it is thus appropriate in this arrangement to have the first series resonant circuit 1302 formed by L.sub.2 and C.sub.5 driven by a non-inverting output (“+”) of the active circuit 1312b and the second series resonant circuit 1303 formed by L.sub.1 and C.sub.4 driven by a non-inverting output (“+”) of the active circuit 1301b. As was discussed above with reference to FIG. 13a, the inverting output (“-”) of the active circuit 1301a is non-ideal from the perspective of removing the interferer signal, but the effect is minimal, while the load balancing advantages are significant. In the arrangement of FIG. 13b, the non-inverting output of the active circuit 1312b dominates over the inverting output of the active circuit 1301b, i.e. gm.sub.1 has much more effect at this node than gm.sub.2. In alternative embodiments (e.g. where there is no load balancing issue), the inverting output of active circuit 1301b could be connected to ground or Vdd instead.
[0079] It will be appreciated that in other examples, if the phases are different, or if the active circuit 1301b, 1311b, 1312b, 1313b senses the signal on the signal path at other points along the signal path, the phases may be different. Also, more than one active circuit may be used to drive the two series resonant circuits 1302, 1303 or 1304, 1305. In each case the phases should be considered, and the active circuit 1301b, 1312b appropriately chosen so as to drive the series resonant circuit 1302, 1303 appropriately. The active circuits 1311b, 1313b for the negative signal arm 1307 operate in exactly the same manner as the active circuits 1301b, 1312b for the positive signal arm 1306.
[0080] FIG. 14 is similar to FIG. 13a, but illustrates a filter 1400 with one particular embodiment of the active circuits 1401, 1411. In this case, each active circuit 1401, 1411 is arranged as a differential amplifier comprising two amplifier elements (e.g. FETs) M.sub.1, M.sub.2 and M.sub.3, M.sub.4, respectively. The amplifier elements M.sub.1, M.sub.2 in 1401 are connected to a current source I.sub.1 and are driven (through connections to their gates) from nodes A and B as illustrated. The amplifier outputs (the drains of M.sub.1, M.sub.2) are connected to the series resonant circuits 1402, 1403 at nodes E and F which in this example are the nodes at the intersection of the inductive element and the capacitive element of the respective series resonant circuit 1402, 1403. The gates of amplifier elements M.sub.3, M.sub.4 in the active circuit 1411 of the negative signal arm 1407 are driven by nodes C and D on the negative signal path and with the drains of M.sub.3, M.sub.4 connected to nodes G and H being the intersections of the inductive element and the capacitive element on each of the first and second series resonant circuits 1404, 1405 of the negative signal arm 1407.
[0081] The required g.sub.m is proportional to the interferer strength. As the signal plus interferer propagates from left to right, the chosen value of g.sub.m (small) really only affects the interference signal (small) at the output. Nonetheless, it is still preferable to load each half of the differential amplifier with the same/similar load, and that is why the drain of M.sub.1 is connected to node E.
[0082] With this arrangement, the active circuits 1401, 1411 provide feed-forward cancellation to the second series resonant circuit 1403, 1405 on each signal arm. As the active circuits 1401, 1411 sense the signal across a reactive component on the signal arm, the active circuits 1401, 1411 are driven by a low input (approximately 0 V) for the signal of interest (the signal that is to be passed by the filter 1400) and is driven by a high input (the voltage drop across the reactive component) for the interferer signal (that is to be rejected by the filter 1400). Away from the resonant frequencies of the series resonant circuits 1402, 1403, 1404, 1405, the series resonant circuits 1402, 1403, 1404, 1405 have little impact on the signal path, and thus, the filter 1400 has low insertion loss. At (or close to) the resonant frequencies of the series resonant circuits 1402, 1403, 1404, 1405 the series resonant circuits themselves provide a short to ground that removes unwanted signal components from the signal path and the active circuit 1401, 1411 enhances these series resonant circuits 1402, 1403, 1404, 1405 further to increase the removal of said unwanted signal components.
[0083] FIG. 15 shows the frequency response (forward transmission coefficient, S.sub.21) of the filter circuit 1400 of FIG. 14 with the series resonant circuits tuned to approximately 4.8 GHz and 5.3 GHz. The ideal filter response is shown in which two distinct notches in the filter response are visible, one at each resonant frequency. The notches are sufficiently close together that they essentially form a single wide-band notch. FIG. 15 also shows the reverse transmission coefficient, S.sub.12 in dB.
[0084] FIG. 16 shows a comparison of the frequency response of a single harmonic trap (single parallel resonant circuit in the signal path) HPF with resonant frequency at approximately 5.3 GHz compared with the frequency response of a dual harmonic trap (two series resonant circuits such as those of FIG. 14) HPF.
[0085] It will be appreciated that many variations of the above embodiments may be made without departing from the scope of the invention which is defined by the appended claims.
