PHASE-CODED FREQUENCY MODULATED CONTINUOUS WAVE (FMCW) RADAR SYSTEM METHOD AND ARCHITECTURE

Abstract

A method of phase-code alignment of one or more phase-coded frequency-modulated continuous wave (PC-FMCW) signals received at a PC-FMCW radar system, is disclosure. The method may include monitoring the amplitude of a phase-coded received signal and whenever a change in the square amplitude of the phase-coded received signal is detected, the signal may be multiplied by a delayed version of the phase code to generate an uncoded signal.

Claims

1. A method of phase-code alignment, the method comprising: receiving one or more phase-coded frequency modulated continuous wave (PC-FMCW) signals from a PC-FMCW radar system; monitoring an amplitude of the one or more PC-FMCW received signals; and upon detecting a change in a square amplitude of the one or more PC-FMCW received signals, multiplying the one or more PC-FMCW received signals by a delayed version of a phase code of the one or more PC-FMCW received signals to generate an uncoded signal.

2. The method of claim 1, further comprising: detecting a first change in square amplitude on a first received signal at a first time delay with respect to a transmitted signal and having a first frequency, and multiplying the first signal by a delayed version of the phase code, and wherein subsequent changes in square amplitude represent receipt of subsequent received signals having respective subsequent time delays with respect to the transmitted signal and subsequent frequencies.

3. The method of claim 1, wherein the multiplication of the signal by a delayed version of the phase code is according to the equation: $\boxed{\begin{array}{c}\left[{\displaystyle \underset{p=1}{\overset{P}{.Math.}}{\beta }_p\cos \left(2\pi f_{p}t-{\tau }_p\right)\cdot C\left(t-{\tau }_p\right)\cdot rect\left(\frac{t-T_c/2}{T_c}\right)}\right]\\ \times \left[{\displaystyle \underset{q=1}{\overset{P}{.Math.}}C*\left(t-{\tau }_q\right)\cdot rect\left(\frac{t-T_c/2}{T_c}\right)}\right]\end{array}}$ .

4. The method of claim 1, wherein the phase code is a code at least one of a Hadamard-Welsch code type, a Zadoff-Chu code type, a Kasami code type, or a Gold code type.

5. (canceled)

6. A method of detecting an object with a phase-coded frequency-modulated continuous wave (PC-FMCW) radar system, the method comprising: generating an initial signal in a synthesizer; phase-coding the initial signal in an encoder to provide a coded signal; generating a transmission signal by modulating a carrier signal with the coded signal; transmitting the coded signal; receiving a reflected signal resulting from the transmitted signal reflecting off an object; decoding the reflected signal; and determining a range of the object from the decoded signal; wherein the step of decoding the reflected signal further comprises performing phase-code alignment of the reflected signal.

7. The method of claim 6, wherein the PC-FMCW comprises an aircraft radar system.

8. The method of claim 7, wherein the code signal is selected to identify a particular aircraft.

9. The method of claim 7, wherein the code signal is selected to identify a particular airline.

10. (canceled)

11. (canceled)

12. A phase-coded frequency-modulated continuous wave (PC-FMCW) radar system comprising: a synthesizer for generating an initial signal; an encoder for phase-coding the initial signal; a local oscillator configured for modulating the phase-coded signal to provide a PC-FMCW signal; a transmitter antenna configured for transmitting the PC-FMCW signal as a transmitted signal; a receiver for receiving a reflected phase-coded signal resulting from the transmitted signal reflecting from an object; a system configured for aligning a phase code of the reflected phase-coded signals; a decoder for decoding the code-aligned signals; and a means for analysing the decoded signals to determine the range of the object.

13. The system of claim 12, where the means for analysing includes a means for performing Fast Fourier Transform, FFT, of the decoded signals.

14. The system of claim 11, wherein the system comprises an aircraft radar system.

15. The system of claim 11, wherein the decoder operates using an algorithm implement in software.

Description

BRIEF DESCRIPTION

[0014] Examples of the radar system according to this disclosure will now be described in more detail with reference to the drawings. It should be noted, that these are examples only, and that variations are possible within the scope of the claims.

[0015] FIG. 1 shows a schematic PC-FMCW radar architecture.

[0016] FIG. 2 shows, schematically, the coder and decoder of FIG. 1.

[0017] FIG. 3 shows an example of phase coding.

[0018] FIG. 4 shows an example of PC-FMCW received signals.

[0019] FIGS. 5(a) to 5(d) illustrate code misalignment in received PC-FMCW signals.

[0020] FIGS. 6(a) to 6(c) illustrate a known way of phase code alignment.

[0021] FIG. 7 shows a decoder for use with a system according to the disclosure.

[0022] FIG. 8 is shown to explain the principle of phase code alignment according to the disclosure.

[0023] FIG. 9 shows operation of decoding according to the disclosure.

DETAILED DESCRIPTION

[0024] The general concept of PC-FMCW radars will first be described, with reference to FIG. 1.

[0025] The typical FMCW radar architecture comprises a signal synthesizer 1 to generate a continuous wave signal which is frequency modulated using a local oscillator 2 and a first mixer 3. The frequency of the transmit signal increases over time. The FM signal is amplified by a high power amplifier 4 and transmitted by a transmitter antenna 5. The transmitted signal, called chirp, is described mathematically as:

[00001]$\begin{array}{cc}{x}_T\left(t\right)=A\left(t\right)\cdot rect\left(\frac{t-T_c/2}{T_c}\right)\cos \left(\varphi \left(t\right)\right)& \text{­­­(1)}\end{array}$

where A(t) is the envelope of the transmitted signal, Φ(t) is the phase of the signal and rect(.Math.) indicates a unitary pulse centred in T.sub.c/2 with width T.sub.c, where T.sub.c represents the duration of the chirp.

[0026] Since the chirp frequency increase linearly with time, the phase Φ(t) is:

[00002]$\begin{array}{cc}\varphi \left(t\right)=2\pi {\displaystyle {\int }_0^{t}f\left(t\right)dt}=2\pi \left(f_{c}t+\frac{1}{2}k{t}^2\right)+{\varphi }_0& \text{­­­(2)}\end{array}$

f.sub.c being the starting frequency of the chirp and k = B.sub.c/T.sub.c being the chirp slope, i.e., the ratio between the chirp bandwidth B.sub.c and the chirp duration T.sub.c.

[0027] When the transmitted signal meets an object in range of the radar, the signal is reflected or echoed by the object and detected at a receiver antenna 6 of the radar system. The received signal is amplified, here by a low noise amplifier 7. The received signal, after de-chirping, which is the beat signal, is represented as:

[00003]$\begin{array}{cc}\begin{array}{c}{x}_B\left(t\right)=\frac{A\left(t\right)A\left(t-\tau \right)}{2}\cdot rect\left(\frac{t-\tau -T_c/2}{T_c}\right)\\ \cdot \cos \left(\varphi \left(t\right)-\varphi \left(t-\tau \right)\right)\end{array}& \text{­­­(3)}\end{array}$

τ being the echo delay, that depends by the target range r and by the target speed v as:

[00004]$\begin{array}{cc}\tau =\frac{2\left(r+vt\right)}{c_0}& \text{­­­(4)}\end{array}$

[0028] To determine the distance of the object, the received signal is compared with the original signal at mixer 8 and the beat signal is then filtered by low pass filter 9 to eliminate high frequency components and is sampled by analog-to-digital converter (ADC) 10. The digitized beat signal can then be processed to analyse the object.

[0029] The reflected signals will each be received with a respective time delay with respect to the transmitted chirp. The beat signals can then undergo a FFT to provide a beat signal spectrum showing power versus frequency for each received beat signal and the range is proportional to the frequency.

[0030] As mentioned above, FMCW is subject to interference where radars are operating close to each other. This can lead to false target detection and also increased noise.

[0031] To address this problem, phase-coded FMCW (PC-FMCW) systems were developed, adding a coder 11 and decoder 12 to the typical FMCW architecture to allow radars to identify their own signals. Phase coding performs a spreading of the transmitted signal and a received signal can only be reconstructed using the proper phase code. Uncoded or differently coded signals at the receiver will appear as noise.

[0032] Coding techniques will be described further below, but in simple terms, the generated signal from the synthesizer 1 is encoded at the coder 11. The encoded signal is then mixed with the local oscillator 2 signal at the first mixer 3, is amplified and transmitted as described above.

[0033] The received signal is decoded in that it is mixed at second mixer 8 with the uncoded transmit signal, via a further mixer 13 which preserves the original phase coding for each beat frequency after dechirping.

[0034] Because of the propagation time of the reflected signal, the signal envelopes of the transmitted signal and the received signal will not match and so the codes are not aligned. It is not, therefore, possible to successfully decode the signal unless some processing occurs to align the signals and codes. Again, this will be discussed further below.

[0035] Returning to the coder 11, and with reference to FIG. 2, the coder 11 multiplies 14 the synthesizer output (the ‘chirp’) with a phase code 15. The phase code has the same duration as the chirp and should be selected to have good auto-correlation properties, e.g. a Kasami code.

[0036] At the decoder 12, the phase-code is aligned 16 with the acquired signal and is then decoded based on the digitised received signal to perform target range and/or speed estimation 17. The results may be displayed on a monitor or other imager (not shown).

[0037] Describing the phase coding now in more detail, a phase coded waveform divides the chirp into a number L.sub.c of time segments or code parts of duration T.sub.code, where L.sub.c defines the length of the code.

[0038] The encoding of the transmit signal represented as above, for a single period T.sub.c provides a phase-coded waveform represented as:

[00005]$\begin{array}{cc}{x}_{TC}\left(t\right)=C\left(t\right)\cdot rect\left(\frac{t-T_c/2}{T_c}\right)\cos \left(\varphi \left(t\right)\right)& \text{­­­(5)}\end{array}$

C(t) being a phase code of length L.sub.c, defined as:

[00006]$\begin{array}{cc}C\left(t\right)={\displaystyle \underset{n=1}{\overset{L_c}{.Math.}}{e}^{j{\varphi }_n}rect\left(\frac{t-\left(n-1/2\right)T_{code}}{T_{code}}\right)}& \text{­­­(6)}\end{array}$

where Φ.sub.n = Φ(n) ∈[0, 2π]; n = 1, ...,L.sub.c is the code phase sequence and T.sub.code the time duration of an element of the sequence. For a binary-code, Φ.sub.n = {0, π} so the code sequence can assume only the values 1 and -1, which cause changes in the phase of the signal every T.sub.code time. As the code changes, phase discontinuities occur.

[0039] This is represented in FIG. 3 where x(t) is the initial signal, C(t) is the code applied to the signal and x.sub.c(t) shows the phase encoded signal where phase shifts can be seen when the code changes between -1 and 1. FIG. 4 shows three different phase coded FMCW received signals. Each signal has a different frequency but is encoded with the same code.

[0040] If the waveform is encoded using a bipolar phase code - i.e. where the codes are -1 and +1, then the envelope of the transmitted signal A(t) = C(t)and the envelope of the received signal B(t) = C(t - τ). Therefore, the beat signal after dechirping at a low-pass filter can be represented as:

[00007]$\begin{array}{cc}{x}_{BC}\left(t\right)=\frac{C\left(t\right)C\left(t-\tau \right)}{2}\cdot rect\left(\frac{t-\tau -T_c/2}{T_c}\right)\cdot \cos \left(\varphi \left(t\right)-\varphi \left(t-\tau \right)\right)& \text{­­­(7)}\end{array}$

[0041] There is, therefore, a delay between the transmitted signal envelope and the received signal envelope. The code misalignment is shown, by way of example, in FIGS. 5(a) to 5(d) where FIG. 5(a) shows the transmitted signal and FIG. 5(b) shows the encoded transmitted signal along a time line. FIGS. 5(c) and 5(d), respectively, show the received signal and received encoded signal and the misalignment in the codes. These envelopes need to be aligned if the received signal is to be decoded.

[0042] Uysal (referred to above) uses a time domain group delay filter, implemented either in hardware or software, to align the envelopes of each beat frequency. The approach used by Uysal is described with reference to FIGS. 6(a) to 6(c). The received signals at each frequency are each received at a different time as shown in FIG. 6(a). The group delay filter determines, after receipt of the latest signal, the delay between each signal and applies a filter function to each signal to align them in time as shown in FIG. 6(b). Each beat signal is delayed proportionally to its frequency as:

[00008]$\begin{array}{cc}{\tau }_D=\frac{1}{k}\left(f_{b,max}-f_b\right)& \text{­­­(8)}\end{array}$

where f.sub.b is the frequency of the beat signal computed as:

[00009]$\begin{array}{cc}{f}_b=\frac{2r}{c_0}k& \text{­­­(9)}\end{array}$

f.sub.b,max is the maximal detectable frequency:

[00010]$\begin{array}{cc}{f}_b=\frac{2r_{max}}{c_0}k& \text{­­­(10)}\end{array}$

being k = B.sub.c/T.sub.c the sweep slope.

[0043] In other words, each beat signal is delayed proportionally to its frequency. Because this process needs to wait until the last beat signal has been received before it can determine the delay, processing is not in real time and information about closer targets is delayed to some extent. Both the hardware and software implementation of the group delay filtering increases the complexity of the radar system since additional processing needs to be added before the FFT operation and very high order filters are required. Furthermore, the signals do not provide any information about any interfering system.

[0044] The solution of the present disclosure provides a new way of phase-code alignment without the use of group delay filters.

[0045] According to the present disclosure, the decoder operates using an algorithm implemented in software. The algorithm operates by using the energy of the received signal to trigger the decoding envelope alignment. Thus, as soon as the first beat signal arrives at the receiver, decoding can begin.

[0046] When the first beat signal is received, it is sampled by the ADC and the coding is aligned with the received signal (rather than, as in the prior art, the waveforms all being aligned to a code).

[0047] The phase decoding involves multiplying the coded signal by the complex conjugate of the code as represented below:

[00011]$\begin{array}{cc}\begin{array}{l}{c}_n\cdot c_n^{\ast }=\left[e^{j{\varphi }_n}\cdot rect\left(\frac{t-\left(n-1/2\right)T_{code}}{T_{code}}\right)\right]\\ \cdot \left[e^{-j{\varphi }_n}\cdot rect\left(\frac{t-\left(n-1/2\right)T_{code}}{T_{code}}\right)\right]\end{array}& \text{­­­(11)}\end{array}$

[0048] Now, Equation (11) is equal to 1 because

[00012]$e^{-j{\varphi }_n}\cdot e^{j{\varphi }_n}=1$

and rect.sup.2(.Math.) = 1 since a rectangular function has unitary amplitude. Accordingly, multiplying the code by its complex conjugate restores the original signal.

[0049] As shown in FIG. 7, in the decoder 12, the code and the beat signal are applied to a decoder mixer 20 to provide the decoded beat signal.

[0050] The phase code alignment according to the disclosure uses an amplitude detector to trigger the alignment and decoding.

[0051] FIG. 9 shows the principle of the alignment process. Similar to what is shown in FIG. 4, three phase-coded FMCW signals are received at the receiver, each delayed in time. The first beat signal x.sub.C,1(t) is received at time τ.sub.1 which represents a first time delay with respect to the transmit signal. The first signal is at a first frequency. The second signal, at a second frequency arrives at a later time τ.sub.2, and the third signal, at a third frequency, arrives at a later time τ.sub.3. The square amplitude of the received signals - shown on line A(t) is used to trigger the decoding procedure.

[0052] As described further with reference to FIG. 9, the beat signal is analysed by an amplitude detector 30. An estimate of noise amplitude 40 may also be taken into account when detecting the signal amplitude. Based on the detected amplitude delays for the code are computed 50. Whenever a change in the square amplitude is detected, the signal is multiplied by a delayed version of the code 60 to generate a delay coded signal which is then decoded 70. This is represented mathematically as follows:

[00013]$\begin{array}{cc}\begin{array}{c}\left[{\displaystyle \underset{p=1}{\overset{P}{.Math.}}{\beta }_p\cos \left(2\pi f_{p}t-{\tau }_p\right)\cdot C\left(t-{\tau }_p\right)\cdot rect\left(\frac{t-T_c/2}{T_c}\right)}\right]\\ \times \left[{\displaystyle \underset{q=1}{\overset{P}{.Math.}}C*\left(t-{\tau }_q\right)\cdot rect\left(\frac{t-T_c/2}{T_c}\right)}\right]\end{array}& \text{­­­(12)}\end{array}$

[0053] Now, defining cos(2πƒ.sub.pt - τ.sub.p) = x.sub.p(ƒ.sub.p, τ.sub.p) and applying the decoding procedure, Equation (12) results to be the sum of two terms:

[00014]$\begin{array}{cc}{\displaystyle \underset{p=1}{\overset{P}{.Math.}}{\beta }_p\cdot x_p\left(f_p,{\tau }_p\right)}& \text{­­­(13a)}\end{array}$

[00015]$\begin{array}{cc}{\displaystyle \underset{\begin{array}{c}p,q=1\\ p\ne q\end{array}}{\overset{P}{.Math.}}{x}_p\left(f_p,{\tau }_p\right)\cdot C\left(t-{\tau }_p\right)\cdot C*\left(t-{\tau }_q\right)}& \text{­­­(13b)}\end{array}$

[0054] The first term describes the decoded signals (code is aligned), the second term includes both the multiplication between shifted versions of the same code as well as multiplication of the complex-conjugate of the code with other codes (i.e., the phase-coded interference). It follows the term in (13b) shall be minimized to reduce the impact of the interference as well as the noise produced by not-synchronized code. This can be done with codes with good auto-correlation proprieties, such as selected sequences of Hadamard-Welsh code, Zadoff-Chu code, Kasami or Gold codes.

[0055] The solution according to the disclosure can be fully implemented in software and does not require additional hardware and is, therefore, less costly and less complex.

[0056] Whilst the system according to the disclosure may provide advantages in many fields where radars are used, and particularly in fields where FMCW radars are used but which can be adversely affected by interference from other radars, the solution of this disclosure is believed to have particular advantages in radars used in a wide range of airborne platforms e.g. fixed-wing aircraft, helicopters, unmanned platforms etc. In addition to the phase coding allowing each aircraft to identify its own radar signals, and not to falsely detect signals from other aircraft as target detection signals, it is also envisaged that phase coding could be implemented by airlines or groups of aircraft being allocated their own specific and reserved code set. In this way, it is envisaged that, for example, an aircraft belonging to a given set or group can provide information e.g. weather information to aircraft in its own group. Decoding could happen in the aircraft but could also take place offline by a controller common to the group.