SYSTEM AND METHOD FOR CONTROLLING COMBINED RADIO SIGNALS

Abstract

A method for controlling a combined waveform, representing a combination of at least two signals having orthogonal frequency multiplexed signal components, comprising: receiving information defining the at least two signals; transforming the information defining each signal to a representation having orthogonal frequency multiplexed signal components, such that at least one signal has at least two alternate representations of the same information, and combining the transformed information using the at least two alternate representations, in at least two different ways, to define respectively different combinations; analyzing the respectively different combinations with respect to at least one criterion; and outputting a respective combined waveform or information defining the waveform, representing a selected combination of the transformed information from each of the at least two signals selected based on the analysis.

Claims

1. A communication method, comprising: receiving data to be communicated to a receiver; defining a multiplex signal comprising modulated subcarriers of a multiple subcarrier signal comprising: pilot signals comprising receiver calibration symbols; and information symbols modulated with portions of the data to be communicated; generating at least two alternate multiplex signal waveforms according to respective values of a parameter applied to all of the modulated subcarriers; analyzing the at least two alternate multiplex signal waveforms using a model of the receiver to determine receiver ability to demodulate the data, dependent on decoding of receiver calibration information, and selectively producing an output dependent on the analyzing.

2. The communication method according to claim 1, wherein the pilot signals are interleaved over time in different frequency subcarriers.

3. The communication method according to claim 1, wherein the pilot signals provide timing, channel estimation, and signal equalization at the receiver.

4. The communication method according to claim 1, wherein the modulated subcarriers each have a preamble, and a payload comprising a cyclic prefix and a data symbol.

5. The communication method according to claim 4, wherein the values of the parameter comprise values of a cyclic shift.

6. The communication method according to claim 1, wherein the information symbols are modulated with the data to be communicated by performing an inverse Fourier transform.

7. The communication method according to claim 1, wherein the analyzing comprises determining a peak to average power ratio of the at least two alternate multiplex signal waveforms.

8. The communication method according to claim 1, wherein the analyzing comprises generation of virtual pilot signals within the model of the receiver.

9. The communication method according to claim 1, wherein the multiplex signal has a bandwidth of at least 20 MHz.

10. The communication method according to claim 1, wherein the analyzing comprises determining a decoding error rate for the data to be communicated.

11. A communication system, comprising: an input port configured to receive data to be communicated to a receiver; a memory configured to store information representing a multiplex signal comprising modulated subcarriers of a multiple subcarrier signal comprising pilot signals comprising receiver calibration symbols, and information symbols modulated with portions of the data to be communicated; at least one automated processor configured to: generate at least two alternate multiplex signal waveforms according to respective values of a parameter applied to all of the modulated subcarriers; and analyze the at least two alternate multiplex signal waveforms using a model of the receiver to determine receiver ability to demodulate the data, dependent on decoding of receiver calibration information, and an output port configured to present an output selectively dependent on the analyzing.

12. The communication system according to claim 11, wherein the pilot signals are interleaved over time in different frequency subcarriers, and provide timing, channel estimation, and signal equalization at the receiver.

13. The communication system according to claim 11, wherein the modulated subcarriers each have a preamble, and a payload comprising a cyclic prefix and a data symbol.

14. The communication system according to claim 13, wherein the values of the parameter comprise values of a cyclic shift.

15. The communication system according to claim 11, wherein the information symbols are modulated with the data to be communicated by performing an inverse Fourier transform.

16. The communication system according to claim 11, wherein the at least one automated processor is further configured to determine a peak to average power ratio of the at least two alternate multiplex signal waveforms.

17. The communication system according to claim 11, wherein the at least one automated processor is further configured to generate virtual pilot signals within the model of the receiver.

18. The communication system according to claim 1, further comprising a power amplifier having a bandwidth of at least 20 MHz configured to transmit a radio frequency signal corresponding to one of the at least two alternate multiplex signal waveforms.

19. A communication system, comprising: a memory configured to store information representing a multiplex signal comprising modulated subcarriers of a multiple subcarrier signal comprising pilot signals comprising receiver calibration symbols, and information symbols modulated with data to be communicated to a receiver; at least one automated processor configured to analyze the at least two alternate multiplex signal waveforms using a model of the receiver to determine receiver ability to demodulate the data modulated in the at least two alternate multiplex signal waveforms, dependent on decoding of the pilot signals; and a power amplifier configured to amplify a radio frequency signal corresponding to one of the at least two alternate multiplex signal waveforms.

20. The communication system according to claim 19, wherein the information symbols are modulated on the information symbols using a frequency or wavelet domain transform.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

[0235] FIGS. 1A and 1B show typical behavior of an orthogonal frequency-domain multiplexed channel in the frequency and time domains, respectively.

[0236] FIG. 2 represents a time-frequency resource grid for an OFDM channel, showing typical locations of pilot symbols according to the protocol for LTE.

[0237] FIG. 3 shows the combination of two OFDM channels in a transmitter using a double-upconversion method.

[0238] FIG. 4 provides a simple block diagram showing how two OFDM channels may be combined, wherein the data bits of one OFDM channel may be cyclically shifted in order to reduce the peak-to-average power ratio (PAPR).

[0239] FIG. 5 shows a block diagram of an OFDM communication system that incorporates the shift-and-add algorithm in the transmitter and a pilot phase tracker in the receiver.

[0240] FIG. 6 shows a block diagram of an OFDM communication system that enables the SAA algorithm for a resource grid as in FIG. 2.

[0241] FIG. 7 shows a top-level flowchart for a generalized carrier aggregation method of the present invention.

[0242] FIG. 8 shows a block diagram of an OFDM communication system, whereby the receiver generates an equalized resource grid based on an array of pilot symbols as in FIG. 2.

[0243] FIG. 9 shows a block diagram that represents the process of equalizing the resource grid at the receiver using an array of pilot symbols as in FIG. 2.

[0244] FIG. 10 shows the structure of two OFDM channels, with cyclic shifting of the data for one channel in order to reduce the PAPR of the combined signal.

[0245] FIG. 11 provides a block diagram showing memory storage of multiple shifted replicas of data from an OFDM channel, with selection of one replica corresponding to minimizing the PAPR of the combined signal.

[0246] FIG. 12A shows a typical 64QAM constellation diagram for a simulated OFDM received signal without added noise.

[0247] FIG. 12B shows a 64QAM constellation diagram for a simulated OFDM received signal with noise added.

[0248] FIG. 13A shows a probability plot for PAPR of the carrier aggregation of simulated OFDM signals, showing reduced PAPR for the method of the invention.

[0249] FIG. 13B shows a probability plot for PAPR of the carrier aggregation of simulated OFDM signals, including the effect of digital predistortion.

[0250] FIG. 13C shows a block diagram of the simulation with results shown in FIGS. 13A and 13B.

[0251] FIG. 14 shows a block diagram of a system according to one embodiment of the invention.

[0252] FIG. 15 shows a block diagram of a system according to one embodiment of the invention, including digital predistortion to compensate for a nonlinear analog amplifier.

[0253] FIG. 16 shows a block diagram of a system according to the invention, implemented on an FPGA.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0254] OFDM channels are comprised of many sub-channels, each of which is a narrow-band signal (FIG. 1A). An OFDM channel itself has a time-varying envelope, and may exhibit a substantial PAPR, typically 9-10 dB. However, if two separate similar OFDM channels are combined, the resulting signal will typically exhibit PAPR of 12-13 dB, for a gain of 3 dB. This is unacceptably large, since it would require a power amplifier with 4 times the capacity to transmit a combined signal that averages only 2 times larger.

[0255] A preferred embodiment therefore provides a PAPR reduction method which reduces the PAPR of a two OFDM channel combined signal from 12-13 dB back down to the 9-10 dB of the original components. This˜3 dB reduction in PAPR is preferably accomplished without degradation of the signal, and without the need to transmit any special side information that the receiver would need to recover the OFDM symbols. Further, the algorithm is simple enough that it can be implemented in any hardware technology, as long as it is sufficiently fast.

[0256] Conventional methods of PAPR reduction focus on combining the sub-channels and generating a single OFDM channel without excessive PAPR. The present technique can be viewed in certain respects as a combination of Partial Transmit Sequence (PTM) and Selected Mapping (SLM).

[0257] In traditional PTS, an input data block of N symbols is partitioned into disjoint sub-blocks. The sub-carriers in each sub-block are weighted by a phase factor for that sub-block. The phase factors are selected such that the PAPR of the combined signal is minimized.

[0258] In the SLM technique, the transmitter generates a set of sufficiently different candidate data blocks, all representing the same information as the original data block, and selects the most favorable for transmission (lowest PAPR without signal degradation).

[0259] The present hybrid approach combines elements of PTS and SLM for summed carrier modulated signals. The various cyclic time-shifts of the oversampled OFDM waveform are searched, and the time-shift with the lowest PAPR selected. One OFDM signal is used as reference and the other carrier modulated signal(s) are used to generate the time-shifts, in a manner similar to PTS. The search window is determined by the cyclic prefix length and the oversampling rate.

[0260] While the phase space of possible combinations of shifts increases tremendously, it would not be necessary to explore all such combinations. In general, very high values of the PAPR occur relatively rarely, so that most time shifts starting with a high-PAPR state would tend to result in a reduction in PAPR. Shifts in multiple channels could be implemented sequentially or in parallel, or in some combination of the two. Thus, for example, any combination with a PAPR within an acceptable range is acceptable, any unacceptable PAPR states occur 1% of the time, the search space to find an acceptable PAPR would generally be <2% of the possible states. On the other hand, if other acceptability criteria are employed, a larger search space may be necessary or appropriate. For example, assuming that there is a higher cost for transmitting a higher PAPR signal, e.g., a power cost or an interference cost, then a formal optimization may be appropriate. Assuming that no heuristic is available for predicting an optimal state, a full search of the parametric space may then be appropriate to minimize the cost.

[0261] This differs from conventional approaches, wherein different OFDM channels are independent of one another, with separate transmit chains and without mutual synchronization. Further, the conventional approaches operate directly on the baseband signals. In contrast, the present method evaluates PAPR on an up-converted, combined signal that incorporates two or more OFDM channels, and the symbol periods for each of these channels must be synchronized. This will not cause problems at the receivers, where each channel is received and clocked independently.

[0262] Some conventional approaches to PAPR are based on clipping, but these inevitably produce distortion and out-of-band generation. Some other approaches avoid distortion, but require special transformations that must be decoded at the receive end. These either require sending side-information, or involve deviations from the standard OFDM communication protocols. The present preferred approach has neither shortcoming.

[0263] OFDM channels used in cellular communications, may be up to 10 or 20 MHz in bandwidth. However, these channels might be located in a much broader frequency band, such as 2.5-2.7 GHz. So, one might have a combination of two or more OFDM channels, each 10 MHz wide, separated by 100 MHz or more. A 10 MHz digital baseband signal may be sampled at a rate as low as 20 MS/s, but a combined digital signal covering 100 MHz must be sampled at a rate of at least 200 MS/s.

[0264] In a preferred embodiment, the signal combination (including the up-conversion in FIG. 3) is carried out in the digital domain at such an enhanced sampling rate. The PAPR threshold test and CSR control are also implemented at the higher rate. This rate should be fast enough so that multiple iterations can be carried out within a single symbol time (several microseconds).

[0265] In order to verify the expectation that the circular time-shift permits reduction in PAPR for combined OFDM channels, without degrading system performance, a full Monte-Carlo simulation of OFDM transmission and reception was carried out. The block diagram of this simulation is summarized in FIG. 6, which represents the “Carrier Aggregation Evaluation Test Bench”, and shows a transmitter that combines OFDM signals at frequencies F.sub.1 and F.sub.2, subject to the SAA algorithm for PAPR reduction. At the receive end, this is down-converted and the signal at F.sub.2 is recovered using a standard OFDM receiver. Along the way, appropriate Additive White Gaussian Noise (AWGN) is added to the channel. The parameters for the Carrier Aggregation simulations include the following. Each packet contains 800 bytes of information, which is modulated over several OFDM symbol periods, and the modulation is 64-QAM (64-quadrature amplitude modulation). Each SNR point is run until 250 packet errors occur. The cyclic prefix is set to ⅛ of the total symbol time. Carriers at frequencies F.sub.1 and F.sub.2 are spaced sufficiently that their spectra do not overlap. The oversampling rate is a factor of 8. Finally, a raised cosine filter was used, with a very sharp rolloff, with a sampling frequency F.sub.s=160 MHz, and a frequency cutoff F.sub.c=24 MHz. FIG. 12A shows an example of a constellation chart of the 64-QAM received signals for the simulation without noise, where a time shift has been applied that is expected to be compatible with the interpolation equalizer of the receiver. In this example, no pilot symbol was transmitted during this time period. The clustering indicates that each bit is received within its required window, with no evidence of bit errors. More generally, no degradation of the signal was observed for an allowable time shift, as expected. FIG. 12B shows a similar 64-QAM constellation chart for the simulation with added Gaussian noise typical of a practical wireless communication system. Again, the simulation shows proper reception of the signal with no significant increase in bit errors.

[0266] FIG. 13A shows the simulated PAPR distribution for a combination of two OFDM signals, combined according to an embodiment of the invention. The Complementary Cumulative Distribution Function (CCDF) represents the probability that the signal has a PAPR greater than a given value. For practical purposes, a CCDF of 10.sup.−4 can be used to define the effective PAPR of a particular waveform. Each of the two component signals has a PAPR of 11 dB (top curve). The combination of the two signals without modification would lead to an increase in PAPR of almost 3 dB (not shown). In contrast, combination using the Codebook Pre-Weighting algorithm of the present invention leads to a decrease of almost 2 dB to about 9 dB (bottom curve). This benefit would be reduced if this Codebook approach is not applied (middle curve).

[0267] FIG. 13B shows the effect of applying digital predistortion (DPD) in addition to Crest Factor Reduction (CFR), as indicated in the simulation block diagram of FIG. 13C. FIG. 13C shows the combination of three OFDM signals, each corresponding to LTE signals of 20 MHz bandwidth. The individual baseband signals are sampled at 30.72 MHz, followed by upsampling to 122.8 MHz, offsetting the frequencies (using a digital multiplier), and adding together to form an IF signal with a 60 MHz band comprising three 20 MHz bands. This is then subject to Crest Factor Reduction (CFR) according to the Codebook Weighting algorithm of the present invention, followed by upsampling (by a factor of two) and digital predistortion (DPD, to simulate the saturation effect of a nonlinear power amplifier PA). Finally, the predistorted signal is sent to a digital-to-analog converter (DAC) and then amplified in the PA. The curve in FIG. 13B labeled CFR Input shows the combined signal, while CFR Output shows the result of PAPR reduction. The curve labeled “+n SCA Processing” (PlusN Smart Carrier Aggregation) corresponds to the signal as broadcast, including the effects of predistortion.

[0268] These simulations have confirmed not only that the SAA algorithm permits reduction of PAPR in combined OFDM channels by ˜3 dB, but also that this reduction is achieved without signal degradation and without the need to send any special side information on the transformations in the transmit signal. This can also be integrated with digital predistortion, without degradation of the PAPR reduction.

[0269] A block diagram of a system according to one embodiment of the invention is shown in FIG. 14, where at least one of the input signals is identified as a multiple subchannel multiplexed (MSM) signal, essentially a generalization of an OFDM signal. The MSM signal is assumed to include pilot signals independent of the information content, which enable the signal to be properly received in the presence of multi-path, Doppler shift, and noise. Here the MSM signal and another signal are combined in a plurality of alternative aggregated signals, where each such alternative combination could be properly received for both such signals at a receiver, without sending additional side-information. A digital model of the receiver, which may incorporate prior transmitted signals, permits determination of which alternative combinations correspond to MSM pilot signals that can be properly tracked at the receiver. Based on this criterion, and at least one other criterion that may be associated with combined signal amplitude (such as peak-to-average-power ratio or PAPR), one or more of the alternative combinations is selected, which may be subject to further processing or selection, e.g., in an iterative selection process using various criteria, for transmission using an automated processor. This may preferably be carried out using a digital IF signal, which is then converted to an analog signal in a digital-to-analog converter (DAC), and then upconverted to the full radio frequency signal in the standard way in an analog mixer before being amplified in the Power Amplifier and transmitted via an antenna. Other types of RF modulators may also be employed.

[0270] FIG. 15 represents a block diagram similar to that in FIG. 14, but with the addition of digital predistortion modules that compensate for nonlinearities that may be present in nonlinear analog components such as the Power Amplifier (including inherent non-linearities, signal-dependent delays, saturation and heating effects). The predistortion is preferably carried out on the alternative combinations, so that the selected combination(s) properly meet all criteria.

[0271] The predistortion may encompass correction of multiple distortion sources, and represent transformations of the signal in the time (delay) and/or frequency domains, amplitude and waveform adjustments, and may be adaptive, for example, to compensate for aging and environmental conditions. In the case of multiple-input multiple-output (MIMO) radio transmission systems (or other signal transmissions), the distortion model encompasses the entire signal transmission chain. This model may include distinct models for the various multipaths, and therefore the selected alternative predistorted signal may represent an optimum for the aggregate system, and not only the “principal” signal component.

[0272] One preferred implementation of the technique involves using a fast field-programmable gate array (FPGA) with blocks for shift-register memories, lookup tables, digital up-conversion, and threshold testing. This is illustrated in FIG. 16, which also shows the optional addition of digital predistortion. In this embodiment, the input digital baseband signals (in the time domain) are first stored in memory registers within the FPGA, and the MSM signal S2 is transformed in a plurality of digital precoders. In one embodiment, these precoders may comprise circular shift registers (CSRs) with different values of the shift parameter. In other embodiments, the range of parameter variation is not time (i.e., the incremental variation in a CSR), but rather another parameter, such as the time-frequency range of a wavelet transform. These shifted versions are chosen so as to be compatible with pilot signal tracking in the receiver, as determined by the lookup-table discriminator block. This LUT may take into account prior shifts, as shown in FIG. 16. FIG. 16 shows several precoding schemas (e.g., circular shifts) being processed in parallel, although serial processing is also possible. Each baseband signal to be combined, is subjected to digital upconversion in the digital upconverter DUC to a proper intermediate frequency (IF), with an increase in sampling rate as appropriate. Sample S1 and each alternative S2 may then be digitally combined in the Digital IF Combiner unit. This is followed by optional digital predistortion in digital predistorters PD, before each alternative combination is sent to the Threshold Tester. The Threshold Tester may, for example, measure the PAPR of each alternative, and choose the alternative with the lowest PAPR.

[0273] Alternatively, an ultrafast digital technology, such as rapid-single-flux-quantum (RSFQ) superconducting circuits, may be employed. As the number of OFDM channels being combined is increased, one needs either to increase the algorithm speed, or alternatively carry out a portion of the processing in parallel.

[0274] This method may also be applied to a reconfigurable system along the lines of cognitive radio, wherein the channels to be transmitted may be dynamically reassigned depending on user demand and available bandwidth. Both the number of transmitted channels and their frequency allocation may be varied, under full software control. As long as all channels follow the same general symbol protocol and timing, one may apply a similar set of Shift-and-Add algorithms to maintain an acceptable PAPR for efficient transmission.