INVERTER CONTROL DEVICE

Abstract

The present disclosure provides an inverter control device which estimates back electromotive force of an induction motor, and which uses a torque current and a flux current so as to calculate a slip frequency and compensate for same, thereby enabling the motor to operate at a constant speed. To this end, the present invention may comprise: a command voltage generating unit for outputting, to an inverter, a three phase PWM voltage with respect to a command frequency on the basis of a voltage/frequency operation; and a slip frequency determining unit for determining a slip frequency on the basis of a phase current and a phase voltage of a motor driven by the inverter.

Claims

1. An inverter control device comprising: a command voltage generating unit configured to receive a command frequency and output 3-phases PWM voltage to an inverter, based on voltage/frequency operation; and a slip frequency determining unit configured to determine a slip frequency based on phase current and phase voltage of a motor driven by the inverter, wherein the slip frequency determining unit includes: a coordinate converting unit configured to: convert the phase current and the phase voltage of the motor to dq-axis phase currents and phase voltages of a stationary coordinate system; a disturbance measurement unit configured to: estimate a back electromotive force of the motor from the dq-axis currents and voltages; and estimate a phase angle of the rotor magnetic flux from the back electromotive force of the motor; a current estimating unit configured to apply the phase angle of the rotor magnetic flux to the dq-axis phase currents to convert the dq-axis phase currents to a torque-based current and a magnetic flux-based current of the rotation coordinate system; and a frequency estimating unit configured to output an estimated slip frequency based on the torque-based current, the magnetic flux-based current and a rotor time constant.

2. The inverter control device of claim 1, wherein the coordinate converting unit includes: dq-axis phase current converting unit configured to convert 3-phases stator currents into dq-axis phase currents of the stationary coordinate system; and dq-axis phase voltage converting unit configured to convert 3-phases stator voltages into dq-axis phase voltages of the stationary coordinate system.

3. The inverter control device of claim 1, wherein the disturbance measurement unit includes: a back electromotive force estimating unit configured to apply a stator resistance and a leakage inductance to the dq-axis phase currents and voltages and pass the dq-axis phase currents and voltages through a low-pass filter to estimate a back electromotive force of the motor; and a phase angle estimating unit configured to estimate the phase angle of the rotor magnetic flux from the back electromotive force of the motor.

4. The inverter control device of claim 3, wherein the back electromotive force estimating unit is configured to estimate the back electromotive force of the motor based on a following [Equation]: $\begin{array}{cc}{E}_{\mathrm{dqrs\_est}}=\frac{\frac{K_p}{\sigma L_s}s+\frac{K_i}{\sigma L_s}}{\begin{array}{c}{s}^2+\frac{K_p}{o{L}_s}s+\\ \frac{K_i}{\sigma L_s}\end{array}}\left(V_{dqss}-R_s{i}_{dqss}-s\sigma L_s{i}_{dqss}\right)& \left[\mathrm{Equation}\right]\end{array}$ where E.sub.dqrs-est is the back electromotive force, Kp and Ki are respectively gains of a proportional controller and a proportional integral controller, R.sub.s is the stator resistance, s is a Laplace operator, σL.sub.s is the leakage inductance, V.sub.dqss is the dq-axis phase voltage, and i.sub.dqss is the dq-axis phase current.

5. The inverter control device of claim 3, wherein the phase angle estimating unit includes: a magnetic flux converting unit configured to convert the back electromotive force of the motor into a rotation coordinate system back electromotive force phase angle; a proportional integral controller configured to adjust a q-axis component of the back electromotive force phase angle to 0 to output a frequency of the rotor magnetic flux; and an integrator configured to integrate the frequency of the rotor magnetic flux to output the phase angle of the rotor magnetic flux.

6. The inverter control device of claim 5, wherein the phase angle estimating unit further includes a low-pass filter configured to pass the estimated slip frequency therethrough to output a compensated slip frequency.

7. The inverter control device of claim 1, wherein the current estimating unit is configured to apply a value obtained by applying a trigonometric function to the phase angle of the rotor magnetic flux to the dq-axis phase currents to convert the dq-axis phase currents to a torque-based current and a magnetic flux-based current.

8. The inverter control device of claim 3, wherein the frequency estimating unit is configured to output the estimated slip frequency based on a following [Equation]: $\begin{array}{cc}{\omega }_{\mathrm{slip\_est}}=\frac{1}{T_r}.Math.\frac{I_{torque}}{I_{Flux}}& \left[\mathrm{Equation}\right]\end{array}$ wherein w.sub.slip_est is the estimated slip frequency, T.sub.r is the rotor time constant, I.sub.torque is the torque-based current, and I.sub.flux is the magnetic flux-based current.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

[0054] FIG. 1 is a control block diagram showing an inverter control device according to the prior art.

[0055] FIG. 2 is a block diagram showing in detail the command voltage generating unit shown in FIG. 1.

[0056] FIG. 3 is an example diagram to illustrate the frequency-voltage relationship.

[0057] FIG. 4 is a circuit diagram showing the inverter shown in FIG. 1.

[0058] FIG. 5 is a detailed block diagram of the slip frequency determining unit shown in FIG. 1.

[0059] FIG. 6 is a schematic block diagram of an inverter control device according to the present disclosure.

[0060] FIG. 7 is a block diagram showing a slip frequency determining unit shown in FIG. 6.

[0061] FIG. 8A and FIG. 8B are control configuration diagrams showing a coordinate converting unit shown in FIG. 7.

[0062] FIG. 9A and FIG. 9B are control configuration diagrams showing a disturbance measurement unit shown in FIG. 7.

[0063] FIG. 10A and FIG. 10B are control configuration diagrams showing a current estimating unit and a frequency estimating unit shown in FIG. 7.

DETAILED DESCRIPTION

[0064] The above-described objects, features, and advantages will be described later in detail with reference to the accompanying drawings, and accordingly, a person having ordinary knowledge in the technical field to which the present disclosure belongs may easily implement the technical idea of the present disclosure. In describing the present disclosure, when it is determined that a detailed description of a known component related to the present disclosure may unnecessarily obscure gist of the present disclosure, the detailed description is omitted.

[0065] Hereinafter, exemplary embodiments according to the present disclosure will be illustrated in detail with reference to the accompanying drawings. In the drawings, the same reference numerals indicate the same or similar elements.

[0066] Hereinafter, an inverter control device according to one embodiment of the present disclosure will be described.

[0067] FIG. 6 is a schematic block diagram of an inverter control device according to the present disclosure.

[0068] Referring to FIG. 6, an inverter control device 100 may include a motor 110, an inverter 120 and an inverter controller 130.

[0069] In this connection, the motor 110 and the inverter 120 are the same as the motor 10 and the inverter 20 included in the inverter control device shown in FIG. 1. Thus, the descriptions thereof are omitted herein.

[0070] The inverter controller 130 may include a command voltage generating unit 140 and a slip frequency determining unit 150. Unlike the inverter controller 30 included in the inverter control device shown in FIG. 1, the inverter controller 130 may directly estimate a rotor magnetic flux and calculate a phase angle using stator voltage and current of the motor 120.

[0071] The command voltage generating unit 140 may receive a frequency corresponding to a sum of a command frequency w.sub.ref and a compensation slip frequency w.sub.slip_comp as an operation frequency. In this connection, the command voltage generating unit 140 may generate 3-phases PWM voltage V.sub.abc_PWM as a command voltage of the inverter 120 corresponding to the operation frequency, and having a constant ratio of output voltage and frequency, based on voltage/frequency (V/f) operation.

[0072] The command voltage generating unit 140 may output the 3-phases PWM voltage V.sub.abc_PWM to the inverter 120. In this connection, the inverter 120 may operate using the 3-phases PWM voltage V.sub.abc_PWM to provide 3-phases output voltage V.sub.abcn to the motor 110.

[0073] The slip frequency determining unit 150 may determine the slip frequency using phase current and phase voltage of the motor 110. Further, the slip frequency determining unit 150 may estimate the back electromotive force of the motor 110 from the phase current I.sub.abcs and the phase voltage V.sub.abcs of the motor 110, and may estimate a phase angle θ.sub.est of the rotor magnetic flux from the back electromotive force of the motor 110. Further, the slip frequency determining unit 150 may compensate for the slip frequency from a relationship between the current and the slip frequency based on the phase angle θ.sub.est of the rotor magnetic flux.

[0074] FIG. 7 is a block diagram showing the slip frequency determining unit shown in FIG. 6. FIG. 8A and FIG. 8B are control configuration diagrams showing a coordinate converting unit shown in FIG. 7. FIG. 9A and FIG. 9B are control configuration diagrams showing a disturbance measurement unit shown in FIG. 7. FIG. 10A and FIG. 10B are control configuration diagrams showing a current estimating unit and a frequency estimating unit shown in FIG. 7.

[0075] Referring to FIG. 7 to FIG. 10B, the slip frequency determining unit 150 may include a coordinate converting unit 160, a disturbance measurement unit 170, a current estimating unit 180, and a frequency estimating unit 190.

[0076] In this connection, FIG. 8A shows a control configuration diagram of a dq-axis phase current converting unit 162 and FIG. 8B shows a control configuration diagram of a dq-axis phase voltage converting unit 164.

[0077] The coordinate converting unit 160 may include the dq-axis phase current converting unit 162 and the dq-axis phase voltage converting unit 164.

[0078] First, the dq-axis phase current converting unit 162 may convert 3-phases abc-axis stator currents, that is, 3-phases abc-axis currents I.sub.as, I.sub.bs, and I.sub.cs into dq-axis phase currents lass and I.sub.qss of the stationary coordinate system. The dq-axis phase voltage converting unit 164 may convert 3-phases abc-axis stator voltages, that is, 3-phases abc-axis phase voltages V.sub.as, V.sub.bs, and V.sub.cs into dq-axis phase voltages V.sub.dss and V.sub.qss of the stationary coordinate system.

[0079] The disturbance measurement unit 170 may include a back electromotive force estimating unit 172 and a phase angle estimating unit 174.

[0080] The back electromotive force estimating unit 172 may receive the dq-axis phase currents I.sub.dss, and I.sub.qss, and the dq-axis phase voltages V.sub.dss and V.sub.qss as inputs and estimate back electromotive power of the motor 110 using the dq-axis phase currents I.sub.ds, and I.sub.qss, and the dq-axis phase voltages V.sub.dss and V.sub.qss.

[0081] The back electromotive force E.sub.dqrs_est of the motor 110 may be estimated using following Equations.

[00004]$\begin{array}{cc}{V}_{dqss}=R_s{i}_{dqss}+\sigma L_s\frac{d}{dt}{i}_{dqss}\frac{L_m}{L_r}\frac{d}{dt}{\lambda }_{\mathrm{dqrs}}& \left[\mathrm{Equation}2\right]\\ \frac{d}{dt}\left[\begin{array}{c}{i}_{dqss}\\ E_{\mathrm{dqrs\_est}}\end{array}\right]=\left[\begin{array}{cc}-\frac{R_s}{\sigma L_s}& -\frac{1}{\sigma L_s}\\ 0& 0\end{array}\right][\begin{array}{c}{i}_{dqss}\\ E_{\mathrm{dqrs\_est}}\end{array}]+\hspace{1em}\left[\begin{array}{c}-\frac{1}{\sigma L_s}\\ 0\end{array}\right]V_{\mathrm{dqss}}& \left[\mathrm{Equation}3\right]\end{array}$

[0082] In this connection, V.sub.dqss is a stator voltage, i.sub.dqss is a stator current, R.sub.s is a stator resistance, σL.sub.s is a stator leakage inductance, L.sub.r is a rotor inductance and L.sub.m is a mutual inductance, λ.sub.dqre is a dq-axis rotor magnetic flux, and E.sub.dqrs_est is the back electromotive force of the motor 110.

[0083] In this connection, [Equation 2] is a voltage Equation of the induction motor, and may be expressed as a sum of voltage resulting from a stator impedance composed of the stator resistance and the leakage inductance and the back electromotive force.

[0084] That is, [Equation 3] is a state Equation having differential terms of the stator current and the back electromotive force. The back electromotive force estimating unit 172 may be designed from [Equation 3].

[00005]$\begin{array}{cc}{i}_{dqss}=\frac{1}{\sigma L_s}{V}_{\mathrm{dqss}}-\frac{R_s}{\sigma L_s}{I}_{\mathrm{dqss}}-\frac{1}{\sigma L_s}{E}_{\mathrm{dqrs\_est}}& \left[\mathrm{Equation}4\right]\\ {\stackrel{.}{E}}_{\mathrm{dqrs\_est}}=G\left(i_{\mathrm{dqss\_est}}-i_{\mathrm{dqss}}\right)=G\left(\frac{1}{\sigma L_s}{V}_{\mathrm{dqss}}-\frac{R_s}{\sigma L_s}{i}_{\mathrm{dqss}}-\frac{1}{\sigma L_s}{E}_{\mathrm{dqrs\_est}}-i_{\mathrm{dqss}}\right)& \left[\mathrm{Equation}5\right]\end{array}$

[0085] In this connection, [Equation 4] and [Equation 5] have differential term of estimated current and the back electromotive force as in [Equation 3]. In [Equation 5], a controller may be used to compare a differential term of a measured current and a differential term of the estimated current with each other to estimate the back electromotive force. However, the differential terms may not only complicate the system but also cause system instability. Thus, the differential terms may be removed by defining an arbitrary variable.

ξ=E.sub.dqrs_est+Gi.sub.dqss  [Equation 6]

{dot over (ξ)}=Ė.sub.dqrs_est+Ġi.sub.dqss  [Equation 7]

[0086] In this connection, in [Equation 6], an arbitrary variable ξ is defined. [Equation 7] is a differential term of ξ. When applying [Equation 6] and [Equation 7] to [Equation 5], a following [Equation 8] may be acquired.

[00006]$\begin{array}{cc}\stackrel{}{.}=-\frac{G}{\sigma L_s}\xi +\frac{G}{\sigma L_s}{V}_{dqss}+G\left(\frac{G}{\sigma L_s}-\frac{R_s}{\sigma L_s}\right)i_{\mathrm{dqss}}& \left[\mathrm{Equation}8\right]\\ E_{\mathrm{dqrs\_est}}=\xi -G{i}_{\mathrm{dqss}}& \left[\mathrm{Equation}9\right]\end{array}$

[0087] In this connection, [Equation 8] and [Equation 9] may corresponding to the back electromotive force estimating unit 172 in which the differential term of the current is removed and may be expressed as [Equation 10] and [Equation 11].

[00007]$\begin{array}{cc}\left(s+\frac{G}{\sigma L_s}\right)=\frac{G}{\sigma L_s}{V}_{dqss}+G\left(\frac{G}{\sigma L_s}-\frac{R_s}{\sigma L_s}\right)i_{\mathrm{dqss}}& \left[\mathrm{Equation}10\right]\\ \left(s+\frac{G}{\sigma L_s}\right)E_{\mathrm{dqrs\_est}}=\left(s+\frac{G}{\sigma L_s}\right)\xi -\left(s+\frac{G}{\sigma L_s}\right)G{i}_{dqss}& \left[\mathrm{Equation}11\right]\end{array}$

[0088] In this connection, s may be a Laplace operator.

[0089] [Equation 10] and [Equation 11] result from [Equation 8] and [Equation 9], respectively. When [Equation 11] is applied to [Equation 10], a following [Equation 12] may be acquired.

[00008]$\begin{array}{cc}{E}_{\mathrm{dqrs\_est}}=\frac{\frac{G}{\sigma L_s}}{s+\frac{G}{o{L}_s}}\left(V_{dqss}-R_s{i}_{dqss}-s\sigma L_s{i}_{dqss}\right)& \left[\mathrm{Equation}12\right]\end{array}$

[0090] In this connection, [Equation 12] represents the back electromotive force estimating unit 172 according to the present disclosure. A polynomial in parentheses on a right side is related to the back electromotive force in [Equation 2]. In [Equation 12], the back electromotive force may be estimated by applying the calculated back electromotive force to a low-pass filter. In this connection, a gain of the back electromotive force estimating unit 172 may act as a cutoff frequency of the low pass filter. When using a proportional controller and a proportional integral controller, transfer functions may be respectively expressed as following [Equation 13] and [Equation 14].

[00009]$\begin{array}{cc}{E}_{\mathrm{dqrs\_est}}=\frac{\frac{K_p}{\sigma L_s}}{s+\frac{K_p}{o{L}_s}}\left(V_{dqss}-R_s{i}_{dqss}-s\sigma L_s{i}_{dqss}\right)& \left[\mathrm{Equation}13\right]\\ E_{\mathrm{dqrs\_est}}=\frac{\begin{array}{c}\frac{K_p}{\sigma L_s}s+\\ \frac{K_i}{\sigma L_s}\end{array}}{\begin{array}{c}{s}^2+\frac{K_p}{o{L}_s}s+\\ \frac{K_i}{\sigma L_s}\end{array}}\left(V_{dqss}-R_s{i}_{dqss}-s\sigma L_s{i}_{dqss}\right)& \left[\mathrm{Equation}14\right]\end{array}$

[0091] In this connection, Kp and Ki may be gains of the proportional controller and the proportional integral controller, respectively.

[0092] [Equation 13] may correspond to the back electromotive force estimating unit 172 composed of a primary low-pass filter using a proportional controller. [Equation 14] may correspond to the back electromotive force estimating unit 172 composed of a secondary low-pass filter using a proportional integral controller.

[0093] In this connection, FIG. 9A shows a control configuration diagram of the back electromotive force estimating unit 172 and FIG. 9B shows a control configuration diagram of the phase angle estimating unit 174.

[0094] That is, the back electromotive force estimating unit 172 may estimate the back electromotive force of the motor 110 based on the above-described [Equation 14].

[0095] The phase angle estimating unit 174 may include a magnetic flux converting unit, a proportional integral controller, and an integrator. The magnetic flux converting unit may convert the back electromotive force of the motor 110 into the rotation coordinate system back electromotive force phase angle. The proportional integral controller may adjust a q-axis component of the back electromotive force phase angle to 0 to output a frequency of the rotor magnetic flux. The integrator may integrate the frequency of the rotor magnetic flux to output the phase angle of the rotor magnetic flux.

[0096] That is, because the back electromotive force phase angle of the motor 110 is 90° ahead of the phase angle of the rotor magnetic flux, the phase angle estimating unit 174 may delay the back electromotive force phase angle by 90° to estimate the phase angle θ.sub.est of the rotor magnetic flux.

[0097] FIG. 10A shows a control configuration diagram of the current estimating unit 180 and FIG. 10B shows the frequency estimating unit 190.

[0098] The current estimating unit 180 may apply a value obtained by applying the trigonometric function to the phase angle θ.sub.est of the rotor magnetic flux to the dq-axis phase currents I.sub.dss and I.sub.qss to convert the dq-axis phase currents I.sub.dss and I.sub.qss not to the active current but to a torque-based current i.sub.torque and a magnetic flux-based current I.sub.flux.

[0099] The frequency estimating unit 190 may output an estimated slip frequency w.sub.slip_est based on the torque-based current I.sub.torque, the magnetic flux-based current I.sub.flux, and a rotor time constant T.sub.r based on a following Equation 15.

[00010]$\begin{array}{cc}{\omega }_{\mathrm{slip\_est}}=\frac{1}{T_r}.Math.\frac{I_{torque}}{I_{Flux}}& \left[\mathrm{Equation}15\right]\end{array}$

[0100] In this connection, w.sub.slip_est is the estimated slip frequency, T.sub.r is the rotor time constant, I.sub.torque is the torque-based current, and I.sub.flux is the magnetic flux-based current.

[0101] The estimated slip frequency w.sub.slip_est calculated based on [Equation 15] may pass through the low pass filter LPE included in the phase angle estimating unit 174 and thus may be outputted as a compensated slip frequency w.sub.slip_comp.

[0102] The compensated slip frequency w.sub.slip_comp may correspond to the speed error. The inverter controller 130 may determine the operation frequency by adding the slip frequency w.sub.slip_comp to the command frequency, such that the inverter may operate at the constant speed regardless of the load.

[0103] The inverter control device according to the present disclosure estimates the back electromotive force and the phase angle of the rotor magnetic flux without phase distortion using the disturbance measurement unit, and estimates and compensates for the slip frequency using the torque-based current and the magnetic flux-based current calculated based on the estimated phase angle of the rotor magnetic flux, such that the inverter may operate at a constant speed regardless of a load.

[0104] Further, the inverter control device according to the present disclosure may be applicable to both a low speed operation region and a high speed operation region and thus may easily control the inverter.

[0105] The present disclosure as described above may be subject to various substitutions, modifications and changes within the scope of the technical idea of the present disclosure by those with ordinary knowledge in the technical field to which the present disclosure belongs. Thus, the present disclosure is not limited to the accompanying drawings.