Resonant power converter and method for converting a DC input voltage to AC or DC output voltage

Abstract

A resonant power converter for converting a DC input voltage to AC or DC output voltage, includes a transistor, and a first inductor connected to an input port for a DC voltage to be converted, the drain being connected to the input port by way of the first inductor, the converter furthermore comprising a first resonant network, connected between the drain of the transistor and ground, the first resonant network being configured so as to extract the fundamental component of a drain-source voltage of the transistor and to phase-shift it by a phase shift angle such that the fundamental component and the drain-source voltage are in phase opposition and thus generate a sinusoidal drive signal.

Claims

1. A resonant power converter for converting a DC input voltage to AC or DC output voltage, comprising a power switch provided with a control electrode, a first electrode and a second electrode connected to ground of the resonant power converter, and a first inductor connected to an input port for a DC voltage to be converted, the first electrode being connected to the input port by way of the first inductor, wherein the resonant power converter further comprises a first resonant network, connected between the first electrode of the power switch and ground, the first resonant network being configured so as to extract a fundamental component of a voltage between the first electrode and the second electrode of the power switch and to phase-shift the fundamental component by a phase shift angle such that said fundamental component and the voltage between the first electrode and the second electrode are in phase opposition and thus generate a sinusoidal drive signal, the resonant power converter also comprising a capacitive divider bridge connected between the first resonant network and the control electrode of the power switch in order to limit an amplitude of the sinusoidal drive signal for the control electrode of the power switch.

2. The resonant power converter as claimed in claim 1, the first resonant network comprising an oscillating network configured so as to generate and maintain, using the power switch, oscillations at a desired switching frequency, and a filtering module for filtering a DC component of said oscillations, connected between the oscillating network and the divider bridge.

3. The resonant power converter as claimed in claim 2, comprising a first series resonant circuit, connected between the first electrode and ground, and configured so as to resonate at a frequency equal to twice the switching frequency.

4. The resonant power converter as claimed in claim 3, the first series resonant circuit comprising a first capacitor and a second inductor.

5. The resonant power converter as claimed in claim 2, the oscillating network comprising a second capacitor in parallel with an assembly formed of a fourth inductor connected in series with a fifth capacitor and with a sixth capacitor, forming a Clapp oscillator with the power switch, the filtering module being connected to the oscillating network at terminals of the sixth capacitor.

6. The resonant power converter as claimed in claim 5, the filtering module forming a low-pass LC filter, formed of a fifth inductor connected to the sixth capacitor and to the capacitive divider bridge, and a seventh capacitor connected to the capacitive divider bridge and to ground.

7. The resonant power converter as claimed in claim 2, the oscillating network comprising a second capacitor in parallel with an assembly formed of a fourth inductor connected in series with a sixth capacitor, forming a Colpitts oscillator with the power switch, the filtering module being connected to the oscillating network at terminals of the sixth capacitor.

8. The resonant power converter as claimed in claim 2, the switching frequency being set between 3 MHz and 300 MHz.

9. The resonant power converter as claimed in claim 1, the phase shift angle being substantially equal to 180°.

10. The resonant power converter as claimed in claim 1, the first electrode being connected to an output port by way of a second series resonant circuit.

11. The resonant power converter as claimed in claim 10, the second series resonant circuit comprising a third inductor connected in series with a third capacitor.

12. The resonant power converter as claimed in claim 1, the capacitive divider bridge comprising an eighth capacitor, connected to the first resonant network and to the control electrode of the power switch, and a fourth capacitor, connected to the control electrode of the power switch and ground.

13. A power conversion method for converting a DC input voltage to AC or DC output voltage in a resonant power converter comprising a power switch provided with a control electrode, a first electrode and a second electrode connected to ground of the resonant power converter, and a first inductor connected to an input port for a DC voltage to be converted, the first electrode being connected to the input port by way of the first inductor, wherein the method comprises steps: extraction, by a first resonant network connected between the first electrode of the power switch and ground, of a fundamental component of a voltage between the first electrode and the second electrode of the power switch, phase-shifting the fundamental component by a phase shift angle such that said fundamental component and the voltage between the first electrode and the second electrode are in phase opposition, said phase-shifted fundamental component forming a sinusoidal drive signal, reducing an amplitude of the sinusoidal drive signal for the control electrode of the power switch.

14. The power conversion method as claimed in claim 13, furthermore comprising an initial step of generating and maintaining oscillations at a switching frequency of the power switch.

15. The method as claimed in claim 14, further comprising a step of filtering a DC component of said oscillations, between the step of phase-shifting the fundamental component and the step of reducing the amplitude of the signal.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) Other features, details and advantages of the invention will emerge upon reading the description, given with reference to the attached drawings that are given by way of example:

(2) FIG. 1 shows a class Φ2 converter.

(3) FIG. 2 shows a waveform of the drain-source voltage V.sub.DS of a class Φ2 converter.

(4) FIG. 3 shows an electrical circuit of a class Φ2 converter equipped with a gate drive circuit according to a first embodiment of the invention, operating with a Clapp oscillator.

(5) FIG. 4 shows an electrical circuit of a class Φ2 converter equipped with a gate drive circuit according to a second embodiment of the invention, operating with a Colpitts oscillator.

(6) FIG. 5 schematically shows the various steps of a method according to the invention.

DETAILED DESCRIPTION

(7) The invention is described in the case where the power switch is a field-effect transistor (for example MOSFET, JFET). The substrate of the transistor may be made of gallium nitride (GaN), of silicon carbide (SiC), or using any other material. The drain, the source and the gate that are mentioned in the description may more generally be denoted by a first electrode, a second electrode and a control electrode, respectively. The invention may thus also be applied to other types of power switch (for example an IGBT transistor, a bipolar transistor or even a thyristor).

(8) FIG. 3 shows an electrical circuit of a class Φ2 converter equipped with a gate drive circuit according to a first embodiment of the invention. A DC voltage Vin is applied to the input of the converter, between the input port of the voltage to be converted 9 and ground GND. A first inductor L1 is connected between the input port 9 and a node 11 to which the drain of the transistor 2 to be switched at a switching frequency f.sub.0 is connected. A second inductor L2 and a first capacitor C1 form a first series resonant circuit 3, connected between the node 11 and ground GND, and configured so as to resonate at a frequency equal to twice the switching frequency f.sub.0 of the transistor, which corresponds substantially to the second harmonic of the switching frequency f.sub.0, in order to reduce the voltage stress on the transistor.

(9) A second series resonant circuit 4, comprising a third inductor L3 connected in series with a third capacitor C3, is connected between the node 11 and the output port 10 of the converted voltage. The converted voltage is shown schematically in FIG. 3 by a load resistor R1. The second capacitor C2 represents the output capacitance of the transistor Cp, shown in FIG. 1, as well as an optional additional capacitor Copt, not shown. The higher the switching frequency, the smaller the capacitance of the second capacitor C2, the second capacitor C2 may then be formed solely of the stray capacitance Cp, without having to add an optional additional capacitor Copt. The second capacitor C2, the fifth capacitor C5, the fourth inductor L4 and the sixth capacitor C6 form an oscillating network 6. The oscillating network 6 according to the invention thus advantageously uses certain stray components of the transistor, notably its output capacitance Cp. The assembly formed of the oscillating network 6 and the transistor 2 forms a Clapp oscillator, whose role is to create oscillations from the DC input voltage Vin. The oscillations are maintained in the gate drive circuit, at a given frequency f.sub.0. The Clapp oscillator has the advantage of being particularly stable in terms of frequency, notably in the radiofrequency range. By simplifying the depiction of the oscillating network 6, the second capacitor C2 is shown between the first series resonant circuit 3 and the branch of the oscillating network 6 formed of the fifth capacitor C5, of the fourth inductor L4 and of the sixth capacitor C6. However, the second capacitor C2 could also be shown “to the right” of the transistor, to better illustrate the fact that it partly represents the output capacitance of the transistor Cp.

(10) A low-pass LC filtering module 8, formed of a fifth inductor L5 and of a seventh capacitor C7, taps off the voltage across the terminals of the sixth capacitor C6 at input; the output signal from the filtering module 8 is recovered at the terminals of the seventh capacitor C7. The role of this filtering module 8 is to extract the fundamental component of the drain-source voltage signal Vds received by the Clapp oscillator, the waveform of which is illustrated in FIG. 2, in order to remove all of the harmonics therefrom. Moreover, the values of the reactive elements (capacitors and inductors) of the filtering module 8 and of the oscillating network 6 are determined such that the fundamental component of the drain-source voltage signal Vds, at the output of the filtering module 8, and the drain-source voltage Vds are in phase opposition, preferably phase-shifted by a value substantially equal to 180°. A capacitive divider bridge 7, formed of a fourth capacitor C4 and of an eighth capacitor C8, makes it possible both to eliminate the DC component of the voltage across the terminals of the seventh capacitor C7 and to reduce the amplitude of the signal from the gate drive circuit. The value of the fourth capacitor C4 is determined depending on the DC component to be eliminated. The value of the eighth capacitor C8 is determined depending on the amplitude reduction to be applied. A sinusoidal drive signal is thus obtained at the output of the capacitive divider bridge 7.

(11) The sinusoidal drive signal represents the output signal from the gate drive circuit. With reference to FIG. 2, when the voltage Vds is non-zero, the phase shift of 180° and the elimination of the DC component result in a sinusoidal drive signal below the threshold voltage (Vgsth) of the transistor. The transistor is therefore in the off state, and therefore no current flows through it. With continuing reference to FIG. 2, when the voltage Vds is zero or virtually zero (for example below a certain threshold), the sinusoidal drive signal is above the threshold voltage (Vgsth) of the transistor, and the transistor changes to the on state, thus with a non-zero current flowing through it. The operation of the soft switching converter (ZVS) is therefore indeed complied with, limiting switching losses, without the need to use an additional voltage source or other active components. The gate drive circuit is then said to be self-oscillating.

(12) The embodiment illustrated in FIG. 4 differs from the embodiment illustrated in FIG. 3 through the oscillating network. In FIG. 4, the transistor 2 and the oscillating network 6′ form a Colpitts oscillator. The Colpitts oscillator comprises one fewer capacitor compared to the Clapp oscillator. Having one fewer capacitor advantageously makes it possible to reduce dissipations due to stray elements of the capacitor, and thus to increase the efficiency of the converter, also with a lower mass. The numerical values of the fourth inductor L4′, of the sixth capacitor C6′, of the fifth inductor L5′ and of the seventh capacitor C7′ may differ from the numerical values of the corresponding components of the Clapp oscillator, in order to account for the absence of the fifth capacitor C5.

(13) FIG. 5 schematically illustrates the various steps of the power conversion method according to the invention. In step 100, the oscillating network (6, 6′) and the transistor 2 generate and maintain oscillations at a switching frequency f.sub.0 of the transistor 2, as soon as a DC voltage Vin is present. In step 101, the first resonant network 5 extracts the fundamental component of the drain-source voltage V.sub.DS of the transistor 2. In step 102, the fundamental component of the drain-source voltage V.sub.DS of the transistor is phase-shifted by a phase shift angle such that said fundamental component and the drain-source voltage V.sub.DS are in phase opposition. In step 103, the DC component of the phase-shifted fundamental component is filtered by the capacitive divider bridge 7, in order to obtain a sinusoidal signal for driving the gate of the transistor 2. The amplitude of this signal may be limited in step 104, compared to the level required by the gate of the transistor 2.

(14) The following paragraph describes one non-limiting example of a method for dimensioning the components of the gate drive circuit, for an oscillating frequency f.sub.0 equal to 100 MHz, taking into account the numerical values of the components of the Φ2 structure of the converter at this frequency.

(15) For a DC input voltage of 20 V, and delivering an output power of around 2 W to a resistive load of 100Ω, the value of 5 nH may be assigned to the first inductor, the value of 3.3 nH may be assigned to the second inductor, the value of 188 pF may be assigned to the first capacitor, the value of 340 nH may be assigned to the third inductor, and the value of 15 pF may be assigned to the third capacitor. Dimensioning the Clapp oscillator consists in determining the values of the second capacitor C2, of the fifth capacitor C5, of the fourth inductor L4 and of the sixth capacitor C6. In order to reduce the current absorbed in the gate drive circuit, a value of the fourth inductor L4 is set that is much higher than that of the first inductor L1 but lower than that of the third inductor L3. It is therefore possible to set L4=100 nH. The value of the second capacitor C2 may be given by the output capacitance of the transistor 2, substantially equal to 200 pF. It is then possible to set C5=C2=200 pF.

(16) The value of the sixth capacitor C6 is calculated by the formula for the oscillating frequency of the Clapp oscillator:

(17) $f0=\frac{1}{2\pi }\sqrt{\frac{\left(\frac{1}{C2}+\frac{1}{C5}+\frac{1}{C6}\right)}{L4}}$

(18) Knowing the value of C2, C5, L4 as well as the oscillating frequency, which it is desired to set at 100 MHz, a possible value of the sixth capacitor C6 is found. This value may be modified depending on the dimensioning of the components of the filtering module 8.

(19) Dimensioning the low-pass LC filter filtering module 8, the role of which is to extract the fundamental component of the drain-source voltage signal received by the Clapp oscillator and to phase-shift it by 180°, consists in determining the value of the fifth inductor L5 and of the equivalent capacitance of the filter C.sub.filter of the filtering module 8, which takes into account the fourth capacitor C4, the seventh capacitor C7 and the eighth capacitor C8. A first condition to be imposed on the filtering module 8 is that the resonant frequency of the filtering module, determined by the fifth inductor L5 and by the equivalent capacitance of the filter C.sub.filter, must be between the oscillating frequency of the Clapp oscillator (f.sub.0, here 100 MHz) and twice this same frequency (here 200 MHz), so as not to select higher-order harmonics. This results in the equation:

(20) $f0<\frac{1}{2\pi \sqrt{L5.C_{\mathrm{filtre}}}}<2.f0$

(21) A second condition to be imposed on the filtering module 8 is the phase shift of 180° at the output of the filtering module 8. For this purpose, the transfer function of the LC filter is calculated, this being given by:

(22) $H\left(\omega \right)=\frac{1}{1-L5.C_{\mathrm{filtre}}.{\omega }^2}$
Where ω=2π.Math.f.sub.0

(23) In order to achieve a phase shift of 180° at the output of the filtering module 8, the transfer function H is required to be a negative real number, which results in:
L5.Math.C.sub.filter.Math.ω.sup.2>1

(24) The two set conditions make it possible to have possible values for L5 and C.sub.filter.

(25) Dimensioning the capacitive divider bridge 7 consists in determining the values of the fourth capacitor C4, of the seventh capacitor C7 and of the eighth capacitor C8. It is noted that:

(26) $C_{\mathrm{filtre}}=\frac{C_8.C_4}{C_8+C_4}+C_7$

(27) By defining a reduction ratio of 1/9 for the capacitive divider bridge 7, the following is then obtained:
C.sub.4=8.Math.C.sub.8

(28) The value of the fourth capacitor C4 is defined according to the DC component to be eliminated from the signal from the filtering module. For a DC component equal to 6 V, a value of C4=200 pF may be suitable. A value of C8=1600 pF is obtained, thereby making it possible to determine the value of the seventh capacitor C7 from the possible values for L5 and C.sub.filter defined above. It should be noted that the sixth capacitor C6, the fifth inductor L5 and the seventh capacitor C7 form a Chebyshev filter. The value of the sixth capacitor C6 may then be modified so as to correspond to the values of the normalized coefficients from the normalization table of the Chebyshev components.

(29) The method for dimensioning the components of the gate drive circuit is identical for a Colpitts oscillator, illustrated in FIG. 4. The Colpitts oscillator is distinguished from the Clapp oscillator by one fewer capacitor (the fifth capacitor C5), which has an influence on the numerical values of the various components of the gate drive circuit.