Method and device for compensating for interfering influences

Abstract

A method for compensating for noise in a secondary radar system is described. The method includes, using a first transceiver, transmitting, in temporally overlapping manner, a first transmission signal containing a first interfering component and a second transmission signal containing a second interfering component, and compensating for at least one of phase shifts or frequency shifts resulting from the first and second interfering components by evaluation of the first and second transmission signals.

Claims

1. A method for compensating for noise in a secondary radar system, the method comprising: using a first transceiver, transmitting a first transmission signal from the first transceiver (s.sub.11(t)) containing a first interfering component, caused by the noise; using the first transceiver, transmitting a second transmission signal (s.sub.12(t)) containing a second interfering component, caused by the noise, in a temporally overlapping manner with the first transmission signal (s.sub.11(t)), the first transmission signal from the first transceiver (s.sub.11(t)) comprising at least one frequency ramp with a first slope and the second transmission signal from the first transceiver (s.sub.12(t)) comprising at least one frequency ramp with a second slope having a sign opposite the first slope; and compensating for at least one of phase shifts or frequency shifts resulting from the first and second interfering components by evaluation of the first and second transmission signals, the compensation comprising: receiving, using the first transceiver, a first transmission signal from a second transceiver (s.sub.21(t)); generating a first measurement signal (s.sub.m1(t)) using mixing or correlation of the first transmission signal from the first transceiver (s.sub.11(t)) and the first transmission signal from the second transceiver (s.sub.21(t)); receiving, using the first transceiver, a second transmission signal from the second transceiver (s.sub.22(t)); and generating a second measurement signal (s.sub.m2(t)) using mixing or correlation of the second transmission signal from the first transceiver (s.sub.12(t)) and the second transmission signal from the second transceiver (s.sub.22(t)); wherein the first measurement signal (s.sub.m1(t)) and the second measurement signal (s.sub.m2(t)) comprise respective components including complex conjugate representations of each other, the respective components corresponding to the first and second interfering components.

2. The method according to claim 1, wherein a first interfering component, resulting from the noise, of the first measurement signal (s.sub.m1(t)) and a second interfering component, resulting from the noise, of the second measurement signal (s.sub.m2(t)) are complex conjugates of each other.

3. The method according to claim 1, wherein the first transmission signal from the first transceiver (s.sub.11(t)) has at least one first factor which represents a complex conjugate to a second factor of the second transmission signal from the first transceiver (s.sub.12(t)).

4. The method according to claim 1, wherein values of the first and second slopes are substantially equal.

5. The method according to claim 1, wherein a base signal used for generation of the first (s.sub.m1(t)) and second (s.sub.m2(t)) measurement signals, or for the first transmission signal from the first transceiver (s.sub.11(t)) and second transmission signal from the first transceiver (s.sub.12(t)) is generated by a shared generator.

6. The method according to claim 1, wherein the first transmission signal from the first transceiver (s.sub.11(t)) or the first measurement signal (s.sub.m1(t)) is based on an output of a first modulation generator (G11); and wherein the first transmission signal from the second transceiver (s.sub.21(t)) or the second measurement signal (s.sub.m2(t)) is based on an output of a second modulation generator (G12).

7. The method according to claim 1, wherein a fundamental signal for at least one of the first transmission signal from the first transceiver (s.sub.11(t)) or the second transmission signal from the first transceiver ((s.sub.12(t)) is generated and then the respective transmission signal is modulated using a vector modulator; and wherein at least one of the first transmission signal from the first transceiver (s.sub.11(t)) or the second transmission signal from the first transceiver ((s.sub.12(t))_is_generated by applying a modulation signal to a real signal input or complex signal input of the vector modulator, to contemporaneously generate the first transmission signal (s.sub.11(t)) and a mirror representation of the first transmission signal (s.sub.11(t)) defining the second transmission signal from the first transceiver ((s.sub.12(0).

8. The method according to claim 1, wherein a frequency corresponding to propagation time information, is derived from at least one of the first (s.sub.m1(t)) or second (s.sub.m2(t)) measurement signals.

9. The method according to claim 1, wherein the first measurement signal (s.sub.m1(t)) is generated by a first mixer (M11) and the second measurement signal (s.sub.m2(t)) is generated by a second mixer (M12).

10. The method according to claim 1, wherein the first and second measurement signals (s.sub.m1(t), s.sub.m2(t)) comprise mixer outputs representing products of one of frequency modulated continuous wave (FMCW) ramps, which are generated using incoherent local oscillators comprising a first local oscillator in the first transceiver and a second local oscillator in the second transceiver.

11. The method according to claim 1, wherein a clock offset between the first transceiver and the second transceiver is determined by comparing the measurement signals.

12. The method of claim 1, wherein a beat frequency corresponding to propagation time information, is derived from at least one of the first or second measurement signals.

13. A system for compensating for noise in a secondary radar system, the system comprising: a first transceiver configured to: generate and transmit a first transmission signal (s.sub.11(t)) containing a first interfering component, caused by the noise, the first transmission signal (s.sub.11(t)) comprising at least one frequency ramp with a first slope; and generate and transmit, in a temporally overlapping manner, a second transmission signal (s.sub.12(t)) containing a second interfering component, caused by the noise, the second transmission signal (s.sub.12(t)) comprising at least one frequency ramp with a second slope having a sign opposite the first slope; evaluation hardware configured to compensate for at least one of phase shifts or frequency shifts resulting from the first and second interfering components using the transmission signals, the compensating comprising: receiving, using the first transceiver, a first transmission signal from a second transceiver (s.sub.21(t)); generating a first measurement signal (s.sub.m1(t)) using mixing or correlation of the first transmission signal from the first transceiver (s.sub.11(t)) and the first transmission signal from the second transceiver (s.sub.21(t)); receiving, using the first transceiver, a second transmission signal from the second transceiver (s.sub.22(t)); and generating a second measurement signal (s.sub.m2(t)) using mixing or correlation of the second transmission signal from the first transceiver (s.sub.12(t)) and the second transmission signal from the second transceiver (s.sub.22(t)); wherein the first measurement signal (s.sub.m1(t)) and the second measurement signal (s.sub.m2(t)) comprise respective components including complex conjugate representations of each other, the respective components corresponding to the first and second interfering components.

14. The system according to claim 13, wherein a first interfering component of the first measurement signal (s.sub.m1(t)) and a second interfering component of the second measurement signal (s.sub.m2(t)) represent complex conjugates of each other.

15. The system according to claim 13, wherein the first transmission signal from the first transceiver (s.sub.11(t)) has a first factor which represents a complex conjugate to a second factor of the second transmitted signal (s.sub.12(t)).

16. The system according to claim 13, wherein the first transceiver comprises a transmitting antenna (TX) and a receiving antenna (RX) such that the transmitting antenna (TX) transmits the first transmission signal (s.sub.11(t)) and the second transmission signal (s.sub.12(t)) and the receiving antenna (RX) receives the first transmission signal (s.sub.21(t)) from the second transceiver and the second transmission signal from the second transceiver (s.sub.22(t)).

17. The system according to claim 13, comprising one or more mixers (configured to generate at least one of the first measurement signal (s.sub.m1(t)) and to generate the second measurement signal (s.sub.m2(t)).

18. The system according to claim 13, comprising a joint mixer configured to generate the first measurement signal and the second measurement signal by mixing.

19. The system according to claim 13, comprising a shared generator for generating a base signal for the first measurement signal (s.sub.m1(t)) and the second measurement signal (s.sub.m2(t)) measurement signal or for the first transmission signal (s.sub.11(t)) and the second transmission signal (s.sub.12(t)).

20. The system according to claim 13, comprising a vector modulator including an output configured to provide the first transmission signal (s.sub.11(t)) and the second transmission signal (s.sub.12(t)).

21. The system of claim 13, further comprising the second transceiver.

22. The system of claim 13, wherein a beat frequency corresponding to propagation time information, is derived from at least one of the first or second measurement signals.

Description

(1) Exemplary embodiments will be explained in greater detail below on the basis of the figures.

(2) In the figures:

(3) FIG. 1 shows a secondary radar comprising two non-coherent transceiving units;

(4) FIG. 2 shows a schematic structure of a radar system according to the invention;

(5) FIG. 3 shows a diagram of the characteristic of a frequency of used signals as a function of time,

(6) FIG. 4 shows an alternative embodiment of a radar system according to the invention in a schematic representation; and

(7) FIG. 5 shows another alternative embodiment of a radar system according to the invention in a schematic representation.

(8) In the following description, the same reference numerals are used for identical and equivalent parts.

(9) One (optionally independent) aspect of this invention is based on the generation, reception and processing of two signals which are being or have been transmitted over the same transmission channel ÜK. One signal is characterized in particular in that a frequency shift due to (phase) noise is (exactly) opposite to a frequency shift due to (phase) noise of the other signal (which may be for example a conventional FMCW signal; with optionally just one frequency ramp). In addition, said signals are preferably transmitted and received simultaneously (at least in a temporally overlapping manner).

(10) FIG. 2 shows a schematic view of a possible structure in which, in the respective NKSE 1 and NKSE 2, the two signals (s.sub.11(t) and s.sub.12(t) or s.sub.21(t) and s.sub.22(t)) are generated from the same clock source LO1 or LO2 by way of the modulation generators G11 and G12 or G21 and G22. Alternatively, it would also be possible to use a single modulator per NKSE, for example by means of direct digital synthesis (DDS). This has the result that the two transmitted signals are oppositely influenced by noise components (in particular by phase noise or uniformly occurring non-linearities of the FMCW ramps). The mixing process by way of the mixers M11 and M12 (or analogously by way of the mixers M21 and M22 on NKSE 2) generates in principle two signal components, of which a signal component relevant for the detection has a relatively low beat frequency and can be separated from a high-frequency component, for example by way of a low-pass filter configured as hardware and/or software. FIG. 3 shows a characteristic of transmitted and received signals over time. Here, s.sub.11(t) and s.sub.12(t) are the signals transmitted simultaneously by NKSE 1, and s.sub.21(t) and s.sub.22(t) are the signals transmitted (simultaneously) by NKSE 2 and received by NKSE 1. The signals s.sub.12(t) and s.sub.22(t) are characterized in that the frequency ramp has a positive slope. The comparison signals s.sub.11(t) and s.sub.21(t) are characterized by a negative slope of the frequency ramp over time. For the following description, it will be assumed that the distance to NKSE 2 is to be determined on NKSE 1. This requires in principle only a transmission of the signals from NKSE 2 to NKSE 1, but not vice versa. Analogously to this description, NKSE 2 (optionally alone) can also determine the distance to NKSE 1 if NKSE 1 transmits the signals. If the signals are transmitted in both NKSEs, the distance estimation may take place in both transceiving units simultaneously. In addition, it should initially be assumed that the two stations have been synchronized in time.

(11) It should also be assumed that the two stations have been pre-synchronized in time, for example by way of a method from U.S. Pat. No. 7,940,743 or as described in Precise Distance and Velocity Measurement for Real Time Locating in Multipath Environments Using a Frequency-Modulated Continuous-Wave Secondary Radar Approach, S. Roehr, P. Gulden, M. Vossiek, 2008. This pre-synchronization serves primarily to ensure that relevant signal components are retained in the baseband after the low-pass filtering. Precise synchronization can take place by synchronizing clock sources, but also by post-processing and correcting the received signals according to the method described below.

(12) In general, the characteristic in FIG. 3 serves only as one possible embodiment, so that a portion of this signal characteristic, which may also have been shifted in time (for example starting at Ts/2) may also be selected. Alternatively, the starting frequency may also be provided with an offset of B or −B for the two ramps, in order to prevent the two ramps from intersecting. One conceivable mode of implementation would also be the simultaneous use of multiple frequency ramps or multiple comparison signals on NKSE 1 and/or NKSE 2. The core concept of this invention can in principle also be transferred to other radar signal forms, such as for example Stepped Frequency Continuous Wave (SFCW) or Orthogonal Frequency-Division Multiplexing (OFDM), provided that the comparison signal (second measurement signal) is complex conjugate to the actual (first) measurement signal, that is to say that disturbances cancel each other out.

(13) The signals transmitted by NKSE 2 and shown in FIGS. 2 and 3 can be described (initially assuming complete time synchronization) by

(14) $s_{21}^{\prime }\left(t\right)=e^{j2\pi \left(\left(f_{c2}+\frac{B}{2}\right)t-\frac{\mu }{2}{t}^2\right)}{e}^{j{\phi }_{n2}\left(t\right)}\mathrm{and}{s}_{22}^{\prime }\left(t\right)=e^{j2\pi \left(\left(f_{c2}-\frac{B}{2}\right)t+\frac{\mu }{2}{t}^2\right)}{e}^{j{\phi }_{n2}\left(t\right)}$
where B is the bandwidth used by the radar system, f.sub.c2 is the carrier frequency of NKSE 2, φ.sub.n2(t) is the phase noise of the local oscillator LO2, and μ=B/T.sub.s is the sweep rate (that is to say the increase in frequency per unit of time). The received signals on NKSE 1 s.sub.21(t)=As′.sub.21(t−τ) and s.sub.22(t)=As′.sub.22(t−τ) are considered here as an attenuated and time-shifted version of the signal transmitted by NKSE 2.

(15) The two linear frequency-modulated signals

(16) $s_{11}\left(t\right)=e^{j2\pi \left(\left(f_{c1}+\frac{B}{2}\right)t-\frac{\mu }{2}{t}^2\right)}{e}^{j{\phi }_{n1}\left(t\right)}\mathrm{and}{s}_{12}\left(t\right)=e^{j2\pi \left(\left(f_{c1}-\frac{B}{2}\right)t+\frac{\mu }{2}{t}^2\right)}{e}^{j{\phi }_{n1}\left(t\right)}$

(17) on NKSE 1 are dependent on the carrier frequency f.sub.c1, the phase noise φ.sub.n1(t) of the local oscillator LO1 and the variables defined above.

(18) After a process of mixing the signals received by NKSE 1 with the locally generated signals and low-pass filtering (preferably carried out by the hardware of the measuring system in order to reduce thermal noise and interference with other radio applications), the following mixed products are obtained

(19) $s_{m1}\left(t\right)=s_{11}^{*}\left(t\right)s_{21}\left(t\right)={\mathrm{Ae}}^{j2\pi \left(\left(f_{c2}-f_{c1}\right)t-\left(f_{c2}+\frac{B}{2}\right)\tau +\mu \mathrm{rt}-\frac{\mu }{2}{r}^2\right)}{e}^{j\left(-{\phi }_{n1}\left(t\right)+{\phi }_{n2}\left(t-\tau \right)\right)}={\mathrm{Ae}}^{j{\Phi }_{m1}\left(t\right)}\mathrm{and}{s}_{m2}\left(t\right)=s_{22}^{*}\left(t\right)s_{12}\left(t\right)={\mathrm{Ae}}^{j2\pi \left(\left(-f_{c2}+f_{c1}\right)t+\left(f_{c2}-\frac{B}{2}\right)\tau +\mu rt-\frac{\mu }{2}{\tau }^2\right)}{e}^{j\left({\phi }_{n1}\left(t\right)-{\phi }_{n2}\left(t-\tau \right)\right)}={\mathrm{Ae}}^{j{\Phi }_{m2}\left(t\right)}.$

(20) In a system with real-value sampling, the complex representation of the mixed signals may also take place, after digitization, by Hilbert transformation.

(21) A complete calculation of the mixed products can be seen below.

(22) For the argument Φ.sub.m2(t) of s.sub.m1(t):

(23) ${\Phi }_{m1}\left(t\right)=2\pi \left\{-\left[\left(f_{c1}+\frac{B}{2}\right)t-\frac{\mu }{2}{t}^2\right]+\left[\left(f_{c2}+\frac{B}{2}\right)\left(t-\tau \right)-\frac{\mu }{2}{\left(t-\tau \right)}^2\right]\right\}-{\phi }_{n1}\left(t\right)+{\phi }_{n2}\left(t\right)=2\pi \left\{-f_{c1}t-\frac{B}{2}t+\frac{\mu }{2}{t}^2+f_{c2}t+\frac{B}{2}t-f_{c2}\tau -\frac{B}{2}\tau -\frac{\mu }{2}\left(t^2-2t\tau +{\tau }^2\right)\right\}+{\mathrm{\Delta \phi }}_n\left(t\right)=2\pi \left\{\left(f_{c2}-f_{c1}\right)t+\frac{\mu }{2}{t}^2-f_{c2}\tau -\frac{B}{2}\tau -\frac{\mu }{2}{t}^2+\mu t\tau -\frac{\mu }{2}{\tau }^2\right\}+{\mathrm{\Delta \phi }}_n\left(t\right)=2\pi \left\{\Delta \mathrm{ft}+\mu \tau t+\left(-f_{c2}-\frac{B}{2}\right)\tau -\frac{\mu }{2}{\tau }^2\right\}+{\mathrm{\Delta \phi }}_n\left(t\right)$

(24) For the argument Φ.sub.m2(t) of s.sub.m2(t):

(25) ${\Phi }_{m2}\left(t\right)=2\pi \left\{\left[\left(f_{c1}-\frac{B}{2}\right)t+\frac{\mu }{2}{t}^2\right]-\left[\left(f_{c2}-\frac{B}{2}\right)\left(t-\tau \right)+\frac{\mu }{2}{\left(t-\tau \right)}^2\right]\right\}+{\phi }_{n1}\left(t\right)-{\phi }_{n2}\left(t\right)=2\pi \left\{f_{c1}t-\frac{B}{2}t+\frac{\mu }{2}{t}^2-f_{c2}t+\frac{B}{2}t+f_{c2}\tau -\frac{B}{2}\tau -\frac{\mu }{2}\left(t^2-2t\tau +{\tau }^2\right)\right\}-{\mathrm{\Delta \phi }}_n\left(t\right)=2\pi \left\{-\left(f_{c2}-f_{c1}\right)t+\frac{\mu }{2}{t}^2+f_{c2}\tau -\frac{B}{2}\tau -\frac{\mu }{2}{t}^2+\mu t\tau -\frac{\mu }{2}{\tau }^2\right\}-{\mathrm{\Delta \phi }}_n\left(t\right)=2\pi \left\{-\Delta \mathrm{ft}+\mu \tau t+\left(f_{c2}-\frac{B}{2}\right)\tau -\frac{\mu }{2}{\tau }^2\right\}-{\mathrm{\Delta \phi }}_n\left(t\right)$

(26) Adding the two signal forms yields a synthetic mixed signal with the resulting argument Φ.sub.msyn(t):
Φ.sub.msyn(t)=2π{2μτt+Bτ−μτ.sup.2}

(27) The square term can usually be ignored here (particularly in the case of narrowband radar systems with relatively slow FMCW chirps and ranges of a few hundred meters), since μr.sup.2<<Bτ.

(28) As a result, both mixed signals have positive frequency components which are dependent on the propagation time.

(29) It is thus possible (by differentiation) to determine (calculate) the two beat frequencies

(30) $f_{b11}\left(t\right)=\frac{1}{2\pi }\frac{d{\Phi }_{m1}\left(t\right)}{\mathrm{dt}}=f_{c2}-f_{c1}+\mathrm{\mu \tau }-\frac{1}{2\pi }\frac{d}{\mathrm{dt}}\left({\phi }_{n2}\left(t-\tau \right)-{\phi }_{n1}\left(t\right)\right)=\Delta f+\mathrm{\mu \tau }+\delta f\left(t\right)\mathrm{and}{f}_{b12}\left(t\right)=\frac{1}{2\pi }\frac{d{\Phi }_{m2}\left(t\right)}{\mathrm{dt}}=f_{c1}-f_{c2}+\mathrm{\mu \tau }+\frac{1}{2\pi }\frac{d}{\mathrm{dt}}\left({\phi }_{n1}\left(t\right)-{\phi }_{n2}\left(t-\tau \right)\right)=-\Delta f+\mu \tau -\delta f\left(t\right)$

(31) which are subject to a statistical deviation due to the correlated noise component δf(t) and a deterministic frequency shift Δf (caused by different carrier frequencies on NKSE 1 and NKSE 2). On account of the complex conjugate phase characteristic of the mixed signals, these two components cause the signal f.sub.b11(t) to shift towards the higher frequencies and cause the signal f.sub.b12(t) to shift towards the lower frequencies, if these two variables assume a positive value.

(32) Summing then yields a synthetic measurement signal having the measurement frequency
f.sub.b(t)=f.sub.b11(t)+f.sub.b12(t)=2μτ,

(33) which no longer has any dependence on the correlated phase noise δf(t) and on the frequency shift Δf. This result can be solved for τ and the distance between NKSE 1 and NKSE 2 can be estimated via the relationship τ=x/c.sub.0, using the propagation speed c.sub.0 of the electromagnetic wave (usually in air). Due to the linear relationship, it is possible to detect multiple objects, that is to say to receive multiple time-shifted and attenuated copies (superposition, or linear combination of target responses) of the transmitted signal.

(34) The phases of the two mixed signals can be grouped as follows:

(35) ${\Phi }_{m1}\left(t\right)=\stackrel{\stackrel{{\Phi }_1\left(t\right)}{︷}}{\left(2\pi \left(\Delta \mathrm{ft}-f_{c2}\tau \right)+{\mathrm{\Delta \phi }}_n\left(t\right)\right)}+\stackrel{\stackrel{{\Phi }_2\left(t\right)}{︷}}{2\pi \mu \tau t}+\stackrel{\stackrel{{\Phi }_3}{︷}}{2\pi \left(-\frac{B}{2}\tau -\frac{\mu }{2}{\tau }^2\right)}={\Phi }_1\left(t\right)+{\Phi }_2\left(t\right)+{\Phi }_3{\Phi }_{m2}\left(t\right)=-\stackrel{\stackrel{{\Phi }_1\left(t\right)}{︷}}{\left(2\pi \left\{\Delta \mathrm{ft}-f_{c2}\tau \right\}+{\mathrm{\Delta \phi }}_n\left(t\right)\right)}+\stackrel{\stackrel{{\Phi }_2\left(t\right)}{︷}}{2\pi \mu \tau t}+\stackrel{\stackrel{{\Phi }_3}{︷}}{2\pi \left(-\frac{B}{2}\tau -\frac{\mu }{2}{\tau }^2\right)}=-{\Phi }_1\left(t\right)+{\Phi }_2\left(t\right)+{\Phi }_3$
This May be Followed by a Fourier Transformation
S.sub.m1(f)={s.sub.m1(t)}={Ae.sup.j(ϕ.sup.1.sup.(t)+ϕ.sup.2.sup.(t)+ϕ.sup.3.sup.)}=A{e.sup.jϕ.sup.1.sup.(t)}*δ(f−μr)e.sup.jϕ.sup.3,
S.sub.m2(f)={s.sub.m2(t)}={Ae.sup.j(ϕ.sup.1.sup.(t)+ϕ.sup.2.sup.(t)+ϕ.sup.3.sup.)}=A{e.sup.jϕ.sup.1.sup.(t)}*δ(f−μr)e.sup.jϕ.sup.3,

(36) of the two time signals s.sub.m1(t) and s.sub.m2(t), wherein the associations Φ.sub.2(t)=2πμτt and Φ.sub.3=−πBτ apply (according to the assumptions above, the term μr.sup.2/2 is negligibly small). Φ.sub.1(t) contains all the remaining components, the phase shifts of which in s.sub.m1(t) and s.sub.m2(t) behave in a manner complex conjugate to one other. The signal S.sub.m1(f) in the frequency domain has an absolute maximum at the abscissa value f.sub.max,1=Δf+μτ+δf(t), and S.sub.m2(f) at f.sub.max,2=−Δf+μτ−δf(t). If the phase value of these two signals belonging to the maximum is then determined, this yields on the one hand ψ.sub.max,1(t)=Φ.sub.1(t)+Φ.sub.3 and on the other hand ψ.sub.max,2(t)=−Φ.sub.1(t)+Φ.sub.3, where the sum ψ.sub.0=ψ.sub.max,1(t)+ψ.sub.max,2(t)=2Φ.sub.3 of these phase values yields the synthetic phase
ψ.sub.1=ψ.sub.max,1(t)+ψ.sub.max,2(t)=−2πBτ

(37) A phase value can thus be determined which is proportional to the distance value or to the channel propagation time. The phase deviation of the synthetic mixed signal is in this case dependent only on the thermal noise, but not on the phase noise of the local oscillators used. The closed assessment of the synthetic mixed product f.sub.b(t) and also of the determined synthetic signal phase ψ.sub.1 is thus particularly advantageous for an accurate distance estimation.

(38) If a time shift T.sub.0 occurs with respect to the start time of the FMCW sequence, it is known from WO 2010/019975 A1 that the synthetic mixed frequencies f.sub.b(t)=2μ(τ−T.sub.0) in NKSE 1 and f.sub.b2(t)=2μ(τ+T.sub.0) in NKSE 2 are obtained. By mutual assessment (respective addition), the unknown time offset can thus be corrected. With the described signal form, this is advantageously also possible via the detected phase deviation, since ψ.sub.1=2πB(τ−T.sub.0) applies on NKSE 1 and ψ.sub.2=−21B(τ+T.sub.0) applies on NKSE 2. The phase shift of the determined phase values with respect to one another is thus linearly dependent on the propagation time r in the channel and on the time offset T.sub.0. The deviation that occurs can be determined by subtraction and can be corrected. Alternatively, an estimated value for the distance or the channel propagation time can also be determined directly by addition. Furthermore, this synchronization method does not require any simultaneous transmission and reception by both stations (duplex mode), but instead can take place sequentially.

(39) With particular advantage, the unambiguity range of the phase estimation ψ.sub.0 can be influenced by changing the start frequencies of the FMCW chirps, that is to say the parameter B, since this, depending on the quality of the pre-synchronization, changes the phase deviation of the two mixed signals with respect to one another. This type of time synchronization can also be carried out in stages, by changing the start frequencies of upsweep and downsweep. A very precise synchronization can be achieved with start frequencies which are far apart, that is to say with a large “spacing” B of the two FMCW ramps.

(40) FIG. 4 shows a system diagram of an embodiment with a complex mixer and upconversion in the transmission branch. The block diagram shown in FIG. 2 contains two real-value mixers which are used to separate upsweep and downsweep and which can also be replaced by a complex mixer (M.sub.RX1 in FIG. 4). Here, one mixed signal is generated with a positive frequency and one is generated with a negative frequency. It would also be possible for the two modulation generators G1 and G2 of NKSE 1 in FIG. 2 to be replaced by just one (G1 in FIG. 4). The signal thereof can then be mixed by a transmitting mixer M.sub.TX1, wherein the two transmission signals can represent an upper and lower sideband. In addition, the two low-frequency mixing products in the case of a secondary radar can preferably be separated by a suitable choice of frequency offset or time offset between the two stations. If chosen appropriately, the use of a complex mixer would be unnecessary, since the two mixing products appear at different positions in the spectrum.

(41) FIG. 5 shows a system diagram with a transmission mixer and upconversion in the transmission branch. The embodiment shown in FIG. 5 contains a single transmitting and receiving antenna instead of two separate antennas, said single antenna being used jointly for transmitting and receiving. For this, use can be made of a transmission mixer TM which has advantageous transmission properties, particularly in the case of FMCW systems. With this concept, too, complex sampling can be omitted if the two stations have an advantageously selected offset of carrier frequency and start time of the respective FMCW signals.

(42) In principle, the method described above for suppressing interfering influences can also be used to reduce the hardware requirements (such as the quality of the phase-locked loop) for generating a high-frequency carrier signal with little phase noise. The resulting errors can then subsequently be compensated for by the described processing.

(43) A few use examples of the invention will be explained in greater detail below:

(44) It is generally known that the accuracy of a distance estimation in a radar system increases as the signal-to-noise ratio (SNR) increases. An indirect proportionality of a signal power with respect to a square of the distance (in secondary radar systems) represents a significant difference to primary radar systems, since the SNR in the latter decreases with the fourth power of the distance. In addition, in the former system, there is no correlation (and therefore no phase noise suppression) by the mixing process. Therefore, given a sufficiently high signal-to-noise ratio, the phase noise (which is largely independent of the distance) often has a greater influence on the measurement accuracy than thermal noise.

(45) The invention can be used for example in an industrial environment, for example for position determination on cranes, as a landing aid, as a locating beacon, on container vehicles and/or on rail-bound or free-moving vehicles. A precise positioning of objects can thus be achieved, which can be used for example for process automation, to optimize production or storage processes, and/or to avoid collisions. This positioning may take place on the basis of a 1D estimation, but a distance between a multiple distributed radar stations can also be determined. In the 2D or 3D case, a position in a plane (or in space) can be determined preferably by means of a multilateration.

(46) Another advantageous use of the method according to the invention are systems in which the generation of a signal with relatively low phase noise (for example with the aid of an optimized phase-locked loop) is omitted. Particularly in mobile radar units, this signal generation represents a significant effort in the production, which emerges as increased hardware costs and an increased energy demand. By virtue of the invention, the associated higher (phase) noise component can thus be reduced or eliminated.

(47) The embodiments shown in FIGS. 4 and 5 can be used with particular preference for simplified signal generation. In this case, a mirror frequency after the upmixing can usefully be used as a comparison signal. In conventional radar systems, this must be suppressed with considerable effort.

(48) Another use may be security-related applications, such as “Keyless Go” for example. One possible form of attack on such a system is the so-called “relay attack”, which is carried out from a remote (radar station), wherein another subscriber can actively engage in the conversation. In this case, an FMCW signal is used not only for a radar application but also for communication. By shifting the frequency of the local oscillator, the (real-time-capable) third communication unit can synthesize smaller distances, listen in on the communication and/or participate in the latter. This cannot be achieved simply by adjusting to a shifted local oscillator frequency (in particular in the case of the signal characteristic shown in FIG. 3).

(49) In addition, the invention optionally enables the determination of a phase value which is proportional to the distance or to a channel propagation time. In a mode with multiple successive chirps (Chirp Sequence Radar), both the distance and the speed of a target can be estimated for example by means of a 2D FFT. In this case, the unambiguity range is dependent on the bandwidth B and not, as in primary radar systems, on the carrier frequency.

(50) On account of a (significant) enlargement of the unambiguity range, lower requirements can be placed on the time sequence of individual FMCW chirps.

(51) With the aid of phase detection, it is also possible to estimate the difference phase between a moving (non-coherent) transceiving unit and multiple distributed (non-coherent) transceiving units, wherein the distributed (non-coherent) transceiving units optionally have a fixed, known position in space. Using the phase differences, it is possible to determine an azimuth or elevation angle with respect to the moving (non-coherent) transceiving unit, or also a position. It is particularly advantageous to use this information in combination with holographic evaluation methods, since a movement creates a large aperture and a very accurate (or high-resolution) estimation of angle or position is possible.

(52) It should be noted at this point that all the parts and functions described above are claimed as essential to the invention individually and in any combination, particularly the details shown in the drawings. Modifications thereof are familiar to a person skilled in the art.

LIST OF REFERENCE SIGNS

(53) G1 modulation generator G11 modulation generator G12 modulation generator G2 modulation generator G21 modulation generator G22 modulation generator LO1 first local oscillator LO2 second local oscillator M1 mixer M2 mixer M11 mixer M12 mixer M21 mixer M22 mixer M.sub.RX1 (complex) mixer M.sub.RX2 (complex) mixer M.sub.TX1 transmitting mixer M.sub.TX2 transmitting mixer NKSE non-coherent transceiving unit NKSE 1 first non-coherent transceiving unit NKSE 2 second non-coherent transceiving unit RX1 receiving antenna RX2 receiving antenna s.sub.11(t) first first signal s.sub.12(t) second first signal s.sub.21′(t) first second signal s.sub.22′(t) second second signal T.sub.M1 transmission mixer T.sub.M2 transmission mixer TX1 transmitting antenna TX1/RX1 transmitting/receiving antenna TX2 transmitting antenna TX2/RX2 transmitting/receiving antenna ÜK transmission channel