Electronically steerable planar phase array antenna

Abstract

A two-dimensional (2-D) beam steerable phased array antenna is presented comprising a continuously electronically steerable material including a tunable material or a variable dielectric material, preferred a liquid crystal material. A compact antenna architecture including a patch antenna array, tunable phase shifters, a feed network and a bias network is proposed. Similar to the LC display, the proposed antenna is fabricated by using automated manufacturing techniques and therefore the fabrication costs are reduced considerably.

Claims

1. A planar continuously steerable phased array antenna comprising: at least three substrate layers from top to bottom; a solid front dielectric substrate layer, an electronically variable dielectric layer, and a solid back dielectric substrate layer, wherein said electronically variable dielectric layer is in-between said front and back dielectric substrate layers; at least one phase shifter including electrodes mounted on a top of the back dielectric substrate layer; at least two radiating elements; a feeding network for providing the impedance matching between the radiating elements and a signal input port; and at least one biasing line that controls the electrical characteristics of the electronically variable dielectric layer and therefore the phase of the RF signal by applying a bias voltage, wherein the radiating elements are located on the top side of the front dielectric substrate layer, wherein the phase shifters having the electrodes are integrated into the antenna and are electronically tunable by utilizing the electronically variable dielectric layer, and whereby the electrodes of the phase shifters are planar transmission lines that feed the signal to the radiating elements.

2. A phased array antenna according to claim 1, whereas the phase shifter electrodes are meandered regularly or irregularly.

3. A phased array antenna according to claim 1, whereas the phase shifter electrodes are arranged spirally.

4. A phased array antenna according to claim 1, whereas at least two phase shifters build a sub-array.

5. A phased array antenna according claim 1, whereas at least four phase shifters build a sub-array.

6. A phased array antenna according to claim 5, whereas the input feed is in the midst of the sub-array.

7. A phased array antenna according to claim 1, further comprising a plurality of sub-arrays.

8. A phased array antenna according to claim 1, where the phase shifter is a time delay unit.

9. A phased array antenna according to claim 1, wherein the electronically tunable phase shifter includes one or more loaded line phase shifters.

10. A phased array antenna according to claim 1, wherein the front and back dielectric substrates comprise mechanically stable, low loss substrates.

11. A phased array antenna according to claim 1, wherein at least one layer selected from the two substrates and the dielectric layer consists of a uniform material.

12. A phased array antenna according to claim 1, whereas the electronically variable dielectric layer of the phase shifter is a liquid crystal.

13. A phased array antenna according to claim 1, where the phase shifters are integrated to the radiating elements and feeding network.

14. A phased array antenna according to claim 1, where the phase shifters are electromagnetically coupled to the radiating elements.

15. A phased array antenna according to claim 1, wherein the bias voltage is applied to one of the electrodes of the phase shifter through a bias line.

16. A method for operating the phased array antenna according to claim 1, the method comprising: receiving the RF signal at a radiating element of the at least two radiating elements; changing the phase of the RF signal by applying the bias voltage across a ground electrode and the at least one phase shifter through the biasing line, wherein the at least one phase shifter is coupled to the radiating element by an aperture coupling and includes electrodes mounted on a top of the back dielectric substrate layer; and coupling the RF signal to the ground electrode having a coplanar waveguide; and coupling the RF signal to an input port to create a contactless RF interconnection between the at least one phase shifter and the input port.

17. A device comprising one or more phased array antennas according to claim 1.

18. A phased array antenna according to claim 1 comprising: a plurality of individual antenna elements, a feed network, and a biasing network.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) FIG. 1 is a block diagram of an example of a two dimensional, electronically steerable phased array antenna according to the present invention;

(2) FIGS. 2a and 2b are exploded and side views of a unit element of the electronically steerable antenna according to an embodiment of the present invention;

(3) FIG. 3 is a schematic representation of a layout of a spiral shape phase shifter;

(4) FIGS. 4a, 4b and 4c are schematic representations of three layouts of the steerable phased array antenna according to the embodiment of the present invention given in FIG. 2;

(5) FIGS. 5a, 5b and 5c are photos of a realized phased array antenna according to the embodiment of the present invention given in FIG. 4;

(6) FIGS. 6a, 6b and 6c are schematic representations of three layouts of the steerable phased array antenna according to another embodiment of the present invention;

(7) FIGS. 7a and 7b are side views of a unit element and a unit sub-array element of an active phased array antenna according to another embodiment of the present invention;

(8) FIG. 8: Simulated Δϕ.sub.b and FoM of the meander and spiral phase shifters without a cpw to microstrip line transitions.

DETAILED DESCRIPTION OF THE INVENTION

(9) In the following, a detailed description is given according to one possible embodiment of the present invention. The embodiment is not dedicated to present every features of the invention instead it provides a basic understanding of some aspects of the invention. It is a two-dimensional steerable antenna which can be used either in receiving or transmitting mode since it is a passive and reciprocal antenna. However, most of the description is given only for a receiving antenna in order to explain the invention in a clear way. The illustrations and relative dimensions may not necessarily be scaled in order to illustrate the invention more efficiently.

(10) Referring to the drawings, FIG. 1 is a block diagram of an electronically steerable phased array antenna 100 according to the present invention. The phased array antenna includes signal input port 101 for example a RF signal input port, feeding network 102, plurality of power combiners 103-109, plurality of DC block structures 110, plurality of electronically tunable phase shifters 111 and plurality of radiating elements 112.

(11) In another embodiment (not shown) the feeding network is on another substrate.

(12) The feeding network 102 may include plurality of transmission lines with different electrical length and characteristic impedance in order to provide the impedance matching between the radiating elements 112 and input port 101. The power combiners 103-109 may combine the power equally or unequally and deliver it to antenna unit element 200 for a desired radiation pattern. According to the antenna theory the distance between the radiating elements 112 is about 0.5 to 0.8 times of the wavelength in vacuum. A lower distance results in high electromagnetic coupling between the elements and a higher distance leads to a grating lobes in the radiation pattern.

(13) FIGS. 2a and 2b show exploded and side views of a unit element 200 of the electronically steerable antenna according to an embodiment of the present invention. The unit element 200 includes a radiating element 112, a tunable phase shifter 111, a DC blocking structure 110 and a bias line 201 in order to apply a bias voltage to the electronically tunable phase shifter 111. These components are placed on three dielectric layers, namely front dielectric substrate 202, tunable dielectric substrate 205 and back dielectric substrate 206.

(14) A radiating element 112 is mounted on the top side of a low loss, front dielectric substrate 202.

(15) As shown here, the radiating element 112 may be a rectangular patch antenna which can be used for different polarizations. In other embodiments the radiating element 112 is a circular, a square patch or any other kind of patch with a slot. A rectangular or square patch can also be cut from one or more corners. It is made of an electrically high conductive electrode. The bottom side of the front dielectric substrate 202 is covered with electrically conductive electrode which forms a ground electrode 203 for the radiating element 112. The ground electrode 203 includes a slot 204 overlying the antenna element 112. An aperture coupling is formed via the slot 204 in order to couple the RF signal between the radiating element 112 and the phase shifter 111. The ground electrode 203 also includes a coplanar waveguide (CPW) which is a part of the DC blocking structure 110.

(16) The preferred embodiment the signal is coupled between the different transmission lines. In another embodiment the signal is coupled capacitively. This means there are two patches, whereas one is mounted on the bottom side of the front dielectric substrate and the other is placed on the top side of the back dielectric substrate, like a parallel plate capacitor.

(17) A tunable dielectric substrate 205 is encapsulated between the front dielectric substrate 202 and a back dielectric substrate 206. A cavity between these two dielectrics 202, 206 is required when the tunable dielectric substrate 205 is liquid. Such a cavity can be accomplished by using appropriate spacers. The mechanical stability of the front and back dielectrics 202, 206 is significant in order to maintain a uniform cavity height. The cavity height can be in the range of a 1 μm . . . 3 μm to several hundred milli-meters depending on the phase shifter topology. For a microstrip line based phase shifters a higher cavity height corresponds to a higher dielectric thickness and therefore the metallic losses are reduced. However, when a liquid crystal material is utilized, the device response time will be relatively longer due to a thick LC layer. On the other hand, the LC cavity height can reduced to 1 μm . . . 50 μm when a loaded line phase shifter is used. In the embodiment of the invention IMSL phase shifter is used. As a compromise between the metallic loss and phase shifter response time a cavity height of about 100 μm is preferred. However, the height can be reduced or increased according to the aforementioned range. If the height is reduced it lets to an increase of the metallic loss, if it is decreased it lets to a reduction of the metallic loss.

(18) In operation of a unit element 200, the RF signal received by the radiating element 112 is coupled to the microstrip line 111, via the aperture coupling which is formed by a slot 204 on the ground electrode 203. The dielectric properties of the variable dielectric substrate 205, and therefore the phase of the RF signal can be changed by applying a bias voltage across the ground electrode 203 and microstrip line 111 through a bias line 201. The bias line 201 is an electrically low conductive electrode, compared to the electrode of the phase shifter 111. The signal is then electromagnetically coupled to the CPW on the ground electrode 203 which is mounted on the bottom side of the front dielectric substrate 202. After propagating along a short CPW line, the RF signal is coupled to the unit element input port 207. By this way, a contactless RF interconnection as a DC blocking structure 110 is achieved between the phase shifter 111 and the unit element input port 207. The variable dielectric substrate 205 is tuned only underneath the microstrip line 111 because the biasing voltage can not affect the rest of the antenna, i.e. other unit elements, due to the DC blocking 110.

(19) In operation of a unit element 200 for a transmitting mode, the transmitting signal received from the array feed network is first electromagnetically coupled from the unit element input port 207 to the CPW on the ground electrode 203. After propagating along a short CPW line, the signal is coupled to the microstrip phase shifter 111. By this way, a contactless RF interconnection as a DC blocking structure 110 is achieved between the phase shifter 111 and the unit element input port 207. The dielectric properties of the variable dielectric substrate 205, and therefore the phase of the transmitted signal can be changed by applying a bias voltage across the ground electrode 203 and microstrip phase shifter 111 through a bias line 201. The bias line 201 is an electrically low conductive electrode, compared to the electrode of the phase shifter 111. After propagating along the microstrip line 111, the signal is coupled to the radiating element 112 by which it is radiated. The coupling between the phase shifter 111 and the radiating element 112 is accomplished via the aperture coupling which is formed by a slot 204 on the ground electrode 203.

(20) The DC blocking structure 110 utilizes the electromagnetic coupling between the similar or different transmission lines mounted on the different layers. It has to be mentioned that the coupling between CPW and microstrip line according to the embodiment is an example of one of the aspects of the present invention. Such a structure can also be optimized so that it may work as a RF filter. The challenge is to suppress the undesired radiation which can affect the antenna radiation characteristic and this can be solved by using an electromagnetic solver.

(21) Electrically tunable phase shifter 111 is fabricated in, but not limited to, inverted microstrip line topology. A microstrip line 111, preferably in spiral shape, is mounted on the top of the back dielectric substrate 206. Its ground electrode 203 is mounted on the bottom side of the front dielectric substrate 202. The electrical properties of such a transmission line can be changed since its dielectric material is a tunable dielectric substrate 205.

(22) Liquid crystal (LC) material can be used as a tunable dielectric substrate 205 at micro- and milli-meter wave frequencies. LC is an anisotropic material with low dielectric losses at these frequencies. Effective dielectric constant of LC for RF field depends on the orientation of the molecules. This property can be exploited to control the wavelength, and thus the phase of an electromagnetic wave, by changing the orientation of LC. The orientation of the molecules can be varied continuously by using an external electric or magnetic field, using a surface alignment of liquid crystal or a combination of these methods.

(23) In another embodiment (not shown) the antenna might consist of a stack of more layers, including more than one LC layer substrates which are separated with at least one layer of solid substrates.

(24) A tunable phase shifter having a differential phase shift of 360° has to be designed in a limited area which is the area of one unit element. The maximum achievable phase shift is frequency dependent and requirements can be adjusted by setting the length of the phase shifter. Due to the limited area, the phase shifter has to be meandered in order to achieve a desired length. Meantime, the coupling between the transmission lines has to be prevented. According to the present invention, the phase shifter is implemented in spiral shape as shown in FIG. 3. Such a phase shifter has 5% to 15% more differential phase shift compared to a meander transmission line, when identical design rules are used and when it is integrated to a radiating element. Additionally, due to the spiral shape, the coupling of RF signal between the phase shifter and the radiating element is accomplished in the centre of the unit element. When the phase shifter 111 is flipped along the axis 301, the unit element input port 207 shifts to the other side whereas the coupling point 302 is still in the centre. This allows flipping the phase shifters in order to design a compact feeding network. Simultaneously, the distance between the radiating elements is kept constant which is crucial for the antenna radiation characteristic. The shape of the phase shifter is not limited to the spiral shape. Its shape can be optimized in order to design compact, high performance phase shifters which can be integrated the antenna array.

(25) According to another aspects of the invention, loaded line phase shifters can be integrated to the antenna array. Within this approach, a non-tunable transmission line is loaded periodically or non-periodically with varactor loads. The varactors can be loaded either serial or shunt to the transmission line.

(26) FIG. 4 illustrates three layouts of a two dimensional, electronically steerable phased array antenna according to the embodiment of the present invention given in FIG. 2. The antenna includes, but not limited to, 16 (4×4) radiating elements 112 which are mounted on the top of the front dielectric 202.

(27) The bottom side of the front dielectric substrate 202 is covered with ground electrode 203 which includes the CPW line segments 110 and the slots 204 for DC blocking structure and aperture coupling, respectively.

(28) The RF signal input port 101, feeding network 102, plurality of power combiners 103, plurality of electronically tunable phase shifters 111, plurality of bias lines 201 and plurality of biasing patches 402 are placed on the top side of the back dielectric substrate 206. A tunable dielectric which is not shown here is in contact with the ground electrode 203 and the top side of the back dielectric substrate 206. The layers can be aligned accurately by using complementary alignment marks 401. The back dielectric layer 206 is enlarged compared to the front dielectric layer 202 from the sides where contacts for RF input port 101 and biasing patches 402 are required.

(29) FIG. 5 illustrates the top, side and bottom view photos of a two dimensional, electronically steerable antenna prototype according to the embodiment of the present invention given in FIG. 4.

(30) The antenna includes four radiating elements. Overall height of the prototype is 1.5 mm including the front, tunable and back dielectric substrates.

(31) FIG. 6 illustrates a unit sub-array element of a phased array antenna according to another embodiment of the present invention. The unit sub-array element 700 includes, but not limited to, 2×2 radiating elements 112 on the top side of the front dielectric substrate 202. The ground electrode 203, slots 204 and the DC blocking structure 110 are mounted on the bottom side of the front dielectric substrate 202. An electrically tunable phase shifter 111, a power combiner 103 and a bias line 201 are fabricated on the top side of the back dielectric substrate 206. A tunable dielectric which is not shown here is in contact with the ground electrode 203 and the top side of the back dielectric substrate 206.

(32) In operation, the RF signal received by the radiating elements 112 is coupled to the power combiner 103 via the aperture coupling 204. The power combiner 103 delivers the signal to the phase shifter 111 which surrounds the power combiner 103. The electrical characteristics of the tunable dielectric substrate and therefore the phase of the RF signal are controlled by applying a bias voltage.

(33) Such a bias voltage is applied through the bias line 201 across the ground electrode 203 and the phase shifter 111. The RF signal is then coupled the sub-array input port 207 via the DC blocking structure 110.

(34) Required numbers of phase shifter and biasing lines are reduced by a factor of radiating element number in the sub-array architecture since all radiating elements are fed through one electronically tunable phase shifter. Similarly, an active phased array antenna requires less number of amplifiers. Due to that, the antenna becomes cost effective and reliable. Concerning the antenna radiation pattern, a differential phase shift between the radiating elements has to be satisfied in order to tilt the radiated phase front. In case of sub-array architecture, this requirement is accomplished for each sub-array. According to the antenna theory the distance between the sub-arrays is about 0.5 to 0.8 times of the wavelength in vacuum.

(35) This reduces the spacing between the radiating elements and therefore, the antenna aperture efficiency is increased. However, the mutual coupling between the radiating elements increases as well. For such an antenna, an optimization process is necessary between the antenna radiation characteristic and cost effectiveness, reliability and biasing complexity when defining sub-array architecture, i.e. radiating element number.

(36) FIGS. 7a and 7b illustrate the side views of a unit element and a unit sub-array element of an active phased array antenna according to another embodiment of the present invention. A low noise amplifier (LNA) 210 is mounted on the bottom side of the dielectric substrate 206. The RF signal received by the radiating element 112 is coupled to a transmission line 211 which is located on the top side of the back dielectric substrate 206. The signal is then coupled to a LNA 210 which is placed on the bottom side of the back dielectric substrate 206. After amplifying, the RF signal is coupled to the tunable phase shifter 111 which has a tunable dielectric substrate 205. By this way, the noise of the components affecting the antenna noise figure is suppressed and therefore antenna noise level is reduced.

(37) The invention has been described in details by means of embodiments. Any changes and modifications of the embodiments are limited by the scope of the following claims.

(38) The realization of an embodiment is explained here:

(39) Realization of a LC based inverted microstrip line (IMSL) phase shifter is shown in FIG. 2. A seed layer made of chromium/gold layer is evaporated on a low loss dielectric substrate. The chromium (Cr) layer has a thickness of 5 nm and is utilized as an adhesive layer between the substrate and the 60 nm thick gold layer. A photoresist (PR) is applied on the seed layer which is then exposed and developed. The electrodes of the structures are formed by electroplating of 2 μm thick gold. After the plating, the PR is removed and the seed layer is etched and therefore only the plated electrodes exist on the substrate. The substrate is diced precisely, i.e. ±5 μm, into two pieces. Each piece is coated with an alignment layer and rubbed mechanically in order to form grooves on the surface. The substrates are then aligned using alignment marks and bonded using glue. LC is filled between the substrates and therefore, appropriate spacers, i.e. micro pearls, are developed on the substrates after the rubbing. Finally, LC is filled and the structure is sealed by which the material is encapsulated between the two substrates. The mechanical stability of the substrates is significant in order to maintain a uniform cavity height. Hence, a low loss glass or ceramic dielectric substrate is preferred for the fabrication.

(40) An embodiment is described here:

(41) A microstrip patch antenna is mounted on the top side of the front dielectric. The ground electrode of the patch antenna is mounted on the bottom side of the same dielectric. The ground electrode includes a slot overlying the patch (FIG. 5c) which form an aperture coupling between the patch antenna and the phase shifter. The strip electrode of the IMSL phase shifter is mounted on the top side of the back substrate. The LC material is encapsulated between the two substrates. It forms the dielectric of the IMSL and has thickness of 100 μm. In operation of a receiving antenna, the received RF signal is first coupled to the phase shifter. After propagating along the phase shifter, the RF signal is electromagnetically coupled to a coplanar waveguide (cpw) which is located on the ground electrode. The signal propagates along a short cpw line, and then it is coupled to the unit element input port which is placed on the top side of the back dielectric. By this way, a contactless RF interconnection as a dc blocking structure is accomplished between the phase shifter and the unit element input port.

(42) More detailed information about further embodiments are:

(43) The unit element is integrated with a LC based tunable phase shifter. The phase shifter has to satisfy a desired differential phase shift Δϕ.sub.b, i.e. 360°, for an optimum beam steering. The differential phase shift of the IMSL is calculated as

(44) ${\mathrm{\Delta \varphi }}_b=\frac{2\pi \mathrm{fl}}{c_0}\left(\sqrt{{\mathrm{.Math.}}_{r,\mathrm{eff},||}}-\sqrt{{\mathrm{.Math.}}_{r,\mathrm{eff},\perp }}\right)$
Whereas, f is frequency, I is physical length, c.sub.0 is the speed of the light in vacuum, ε.sub.r,eff,⊥ relative effective perpendicular permittivity, ε.sub.r,eff,∥ relative effective parallel permittivity.

(45) The length of a phase shifter operating at 18 GHz with a Δϕ.sub.b of 360° is determined as 5.65λ.sub.0 using a specific type of LC. On the other hand, the size of the unit element is set to be 0.65λ.sub.0×0.65λ.sub.0 in order to prevent grating lobes. Hence, the phase shifter has to be designed in a compact way due to the limited area of the unit element. One possible solution is to meander the phase shifter. In this case, however, the coupling between the lines becomes an issue. It can be minimized within the simulation by optimizing the gap between the lines. The total length of the phase shifter is 75 mm and the phase shifter itself (without the transitions) utilizes an area of 0.5λ.sub.0×0.5λ.sub.0 at 18 GHz. This area is less than the area of the unit element. This is due to the fact that when the unit elements are combined in order to form an array, the RF feed network and the bias network require certain amount of area as well.

(46) The performance and the compactness of the phase shifter can be improved further depending on its geometry. For this manner, the geometry, in which the microstrip line is meandered, is significant. One possible solution is to meander the phase shifter in spiral geometry. Such a phase shifter has several improvements compared to the meander line phase shifter. Both phase shifters are designed on the same size of area using the identical design rules, i.e. identical gap size between two electrodes. In FIG. 8, simulated Δϕ.sub.b and FoM results of the phase shifters are given.

(47) As can be seen from FIG. 8, the ΔΦ.sub.b of the spiral phase shifter is 5% to 15% more compared to that of the meander phase shifter. Meantime, the insertion loss is kept almost constant and therefore the FoM is increased, for instance, from 95°/dB to 105°/dB at 18 GHz. Additionally, due to the spiral geometry, the coupling of the RF signal between the phase shifter and the radiating element is accomplished in the centre of the unit element. When the phase shifter geometry is flipped, the unit element input port shifts to the other side whereas the coupling point is still in the centre. This allows flipping the phase shifters in order to design a compact RF feed network. Simultaneously, the distance between the radiating elements is kept constant which is crucial for the antenna radiation characteristic.

(48) The antenna array requires a bias network in order to tune the phase shifters independently. The voltage applied across the bias pads and the ground electrode are delivered to the RF circuitry through the bias lines. The bias lines have to be implemented using a low electrically conductive material and therefore they have negligible impact on the RF signal. Possible materials are indium tin oxide (ITO), chromium (Cr) or nickel-chromium (Ni—Cr). Although having relatively high conductivity (σ=7.8×106 S/m), the Cr adhesive layer is utilized for implementing the bias lines. It has a thickness of 5 nm which results in a sheet resistance of 25:3=sq. The line width is set to be 10 μm in order to increase the bias line resistance.

(49) The 2D-antenna can also be 3D in structure, e.g. it can be wrapped around an object.

DESCRIPTION OF THE REFERENCE NUMBERS

(50) FIG. 1: Block diagram of an example of a two dimensional, electronically steerable phased array antenna according to the present invention

(51) FIGS. 2a and 2b: Exploded and side views of a unit element of the electronically steerable antenna according to an embodiment of the present invention

(52) FIG. 3: Schematic representation of a layout of a spiral shape phase shifter

(53) FIGS. 4a, 4b and 4c: Schematic representations of three layouts of the steerable phased array antenna according to the embodiment of the present invention given in FIG. 2

(54) FIGS. 5a, 5b and 5c: Photos of a realized phased array antenna according to the embodiment of the present invention given in FIG. 4

(55) FIGS. 6a, 6b and 6c: Schematic representations of three layouts of the steerable phased array antenna according to another embodiment of the present invention

(56) FIGS. 7a and 7b Side views of a unit element and a unit sub-array element of an active phased array antenna according to another embodiment of the present invention

(57) FIG. 8: Simulated ΔΦ.sub.b and FoM of the meander and spiral phase shifters without a cpw to microstrip line transitions. 100 Electronically steerable phased array antenna 101 Signal input port 102 feeding network 103-109 power combiners 110 DC blocking structure 111 phase shifters electrodes 112 radiating element 200 Antenna unit element 201 bias line 202 front dielectric substrate 203 ground electrode 204 slot/aperture coupling 205 tunable dielectric substrate 206 back dielectric substrate 207 unit element input port 210 low noise amplifier (LNA) 211 Transmission line 301 flip axis 302 coupling point 401 alignment marks 402 biasing patch 700 unit sub-array element