UPLINK SIGNAL TO INTERFERENCE PLUS NOISE RATIO ESTIMATION FOR MASSIVE MIMO COMMUNICATION SYSTEMS

Abstract

This invention presents methods for estimating the uplink SINR and channel estimation error level in MU-MIMO wireless communication systems comprising the BS obtaining the channel coefficients between each receiving antenna of a BS and a transmitting antenna of a UE in the uplink; for the BS estimating the SU-MIMO SINR of a UE using the channel coefficients between a UE and the BS; for the BS estimating the channel estimation error level of a UE using the channel coefficients between a UE and the BS.

Claims

1. A method for estimating the uplink SINR and channel estimation error level in MU-MIMO wireless communication systems comprising the BS obtaining the channel coefficients between each receiving antenna of a BS and a transmitting antenna of a UE in the uplink; the BS estimating the SU-MIMO SINR of a UE using the channel coefficients between a UE and the BS; the BS estimating the channel estimation error level of a UE using the channel coefficients between a UE and the BS.

2. The method in claim 2 further comprising a UE transmitting pilot signals in more than one continuous OFDM symbols in the uplink.

3. The method in claim 2 further comprising the BS estimating the channel coefficients between each transmitting and receiving antenna pair at plural of subcarriers of the continuous pilot symbols after receiving the pilot signals and compensating the time offset and frequency offset if needed.

4. The method in claim 1 further comprising the BS estimating the noise level of the estimated channel coefficients by averaging a difference between any two coefficients at the same subcarrier belonging to different pilot symbols over multiple antenna pairs.

5. The method in claim 1 further comprising the BS estimating the power level of the estimated channel coefficients by averaging a difference between two coefficients at two continuous subcarriers belonging to same pilot symbol.

6. The method in claim 1 further comprising the BS calculating the uplink pilot SINR per antenna with the estimated power level of the channel coefficients and power level of the noise.

7. The method in claim 1 further comprising calculating the uplink pilot SU-MIMO SINR with the estimated channel coefficients and the noise power levels.

8. The method in claim 1 further comprising calculating the SU-MIMO SINR for uplink data transmission by adding an offset to the uplink pilot SU-MIMO SINR.

9. The method in claim 1 further comprising calculating the SU-MIMO SINR for downlink data transmission by adding an offset to the uplink pilot SU-MIMO SINR or SU-MIMO SINR for uplink data transmission.

10. The method in claim 1 further comprising calculating the error level of the uplink estimated channel with the estimated channel coefficient power level and the noise power level.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

[0009] The aforementioned implementation of the invention as well as additional implementations would be more clearly understood as a result of the following detailed description of the various aspects of the invention when taken in conjunction with the drawings. Like reference numerals refer to corresponding parts throughout the several views of the drawings.

[0010] FIG. 1 illustrates the processing of CSI and SINR estimation

[0011] FIG. 2 illustrates the structure of the radio time-frequency resource.

[0012] FIG. 3 illustrates the structure of a resource block.

[0013] FIG. 4 illustrates the structure of the combs in the SRS symbols.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0014] For a massive MU-MIMO OFDM communication systems, where the TDD mode is employed for uplink and downlink transmission multiplexing. The radio resource (time and frequency domains) allocation to the uplink and downlink is shown in FIG. 2, where the resource is organized in units of frame consisted of the whole bandwidth in the frequency domain and consecutive time duration denoted by T.sub.F in the time domain. One frame is divided into n subframes further, among which n.sub.1 and n.sub.2=n−n.sub.1 subframes are reserved for the downlink and uplink transmission respectively. Note that the numbers and the indices of downlink and uplink subframes are configurable in a frame. One subframe is consisted of N.sub.sym OFDM symbols in the time domain. FIG. 2 shows an example of the structures of frame 6 and subframe 7. One subframe 7 is consisted of multiple Resource Blocks (RBs) , where one RB is consisted of N.sub.sc consecutive subcarriers in the frequency domain, e.g., N.sub.sc=12 in 3GPP-LTE/LTE-A, and all the SC-OFDM/OFDM symbols 8 in the time domain. One subcarrier in a symbol is called a Resource Element (RE), which is the smallest data transmission unit. FIG. 3 shows an example of RB structure 9, where N.sub.sc=12 and N.sub.sys=14, and a rectangle denotes one RE. The number of RBs and so subcarriers used in one subframe depends on the channel bandwidth as shown in Table 1 below.

TABLE-US-00001 TABLE 1 LTE Channel Number Number of subcarriers Bandwidth of RBs per symbol 1.4 MHz  6 72  3 MHz 15 180  5 MHz 25 300 10 MHz 50 600 15 MHz 75 900 20 MHz 100 1200
The uplink SRS values for each UE are allocated to these subcarriers. The number of SC-OFDM/OFDM symbols reserved for SRS is one or two for each UE in a uplink subframe, which depends on the resource allocation by the BS. In order to multiplex more UEs to transmit SRS on the same symbol, two stratagems can be employed: (1). The whole frequency band is divided into several sections where each one contains a set of continuous subcarriers; (2). The subcarriers in each section can be divided into several groups, where each group is called a comb. The numbers of sections and the combs are configurable according to the specific application scenario. Multiple UEs are multiplexed in one comb through different cyclic shift version of a root sequence, e.g., for a specific comb, the SRS sequences sent by all UEs in one comb, r.sub.u,v.sup.α(n) is defined by a cyclic shift α of a root sequence r.sub.u,v(n) according to

r.sub.u,v.sup.α(n)=e.sup.jαnr.sub.u,v(n), 0≦n≦M.sub.sc.sup.SRS,   (1)

where r.sub.u,v(n) is the root sequence with Constant Amplitude Zero Auto Correlation (CAZAC) and M.sub.sc.sup.SRS is the length of the root sequence, which equals to the number of subcarriers contained in the comb. Note that the indices u and v uniquely determines the root sequence. Although the SRS transmitted by different UEs in one comb are superposed at the receiver, the channel between each UE and the BS can be almost perfectly separated because of the unique value of α used by each UE.

[0015] When two SRS symbols are reserved for each UE in a subframe, the root sequence of a comb in the first symbol should be different from that of the second symbol, which ensures the accuracy of the SINR estimation. This can be realized by a pre-defined root sequence allocation, e.g., the indices u and v are determined by the cell identification and the SRS symbol index in a subframe, or other ways, e.g., the BS allocates u, v, and informs the involved UEs through the downlink control channel.

[0016] When the BS received the SRS, it first estimates the channel coefficients between each receive antenna and each UE with the methods such as in [1]. With these estimated channel coefficients, it estimates the TO and FO of each UE and corrects them by compensates the TO and FO contained in the estimated channel coefficients respectively. After that, the BS estimates the uplink SINR of each UE.

[0017] FIG. 4 is an example of SRS transmission, where two symbols 11,12 are reserved for SRS in uplink subframe. In this example, we assume the number of comb in one section is 2, where K UEs, each UE with a single transmit antenna, are multiplexed on each comb. Let M denote the number of receive antenna at the BS side, then after correcting the TO and FO, the channel coefficient between the k.sup.th UE and the m.sup.th antenna on the i.sup.th subcarrier is denoted by Ĥ.sub.m,k,i.sup.i in a specific comb, where t=1,2, m=1, . . . , M, k=1, . . . , K and i=1, . . . , M.sub.sc.sup.SRS. Note that the following descriptions are based on this example but it would not limit the application of the invention.

[0018] For the k.sup.th UE, the BS first selects the subcarriers set Ω.sub.est,k in a comb, e.g., the subcarriers of a comb except these located on the two boundary frequency band and |Ω.sub.est,k|=M.sub.sc.sup.SRS−N.sub.dis, where |Ω.sub.est,k| where |Ω.sub.est,k| denotes the cardinality of Ω.sub.est,k and N.sub.dis is the number of subcarriers discarded. Then, the noise and interference power of k.sup.th UE can be estimated as

[00001]$\begin{array}{cc}{\hat{P}}_{\mathrm{NI}}=\frac{1}{2.Math..Math.\mathrm{MK}.Math..Math.{\Omega }_{\mathrm{est},k}.Math.}.Math.\underset{m=1}{\overset{M}{.Math.}}.Math..Math.\underset{k=1}{\overset{K}{.Math.}}.Math..Math.\underset{i\in {\Omega }_{\mathrm{est},k}}{.Math.}.Math..Math.{.Math.{\hat{H}}_{m,k,i}^t-{\hat{H}}_{m,k,i+1}^t.Math.}^2.& \left(2\right)\end{array}$

[0019] With (2), the signal power of the k.sup.th UE on the m.sup.th antenna is estimated as

[00002]$\begin{array}{cc}{\hat{P}}_S^{m,k}=\frac{1}{2.Math..Math.{\Omega }_{\mathrm{est},k}.Math.}.Math..Math.\underset{t=1}{\overset{2}{.Math.}}.Math..Math.\underset{i\in {\Omega }_{\mathrm{est},k}}{.Math.}.Math..Math.{.Math.{\hat{H}}_{m,k,i}^t-{\hat{H}}_{m,k,i+1}^t.Math.}^2-{\hat{P}}_{\mathrm{NI}}.& \left(3\right)\end{array}$

Obviously, the SINR of the k.sup.th UE on the m.sup.th antenna can be directly calculated as

[00003]$\begin{array}{cc}{\mathrm{SINR}}_{\mathrm{ant}}^{m,k}=\frac{{\hat{P}}_S^{m,k}}{{\hat{P}}_{\mathrm{NI}}}.& \left(4\right)\end{array}$

[0020] In practical systems, the SINR or channel quality indication (CQI) of each UE has to be estimated for transmission rate prediction. The ideal uplink SU-MIMO SINR in a specific frequency band, e.g., the subcarrier set Ω ∈ Ω.sub.est,k, defined as

[00004]$\begin{array}{cc}{\mathrm{SINR}}_k^{\mathrm{SU},\Omega }=\frac{\underset{t=1}{\overset{2}{.Math.}}.Math..Math.\underset{m=1}{\overset{M}{.Math.}}.Math..Math.\underset{i\in \Omega }{.Math.}.Math..Math.{.Math.H_{m,k,i}^t.Math.}^2}{2.Math..Math.\Omega .Math..Math.{\sigma }_{\mathrm{NI}}^2},& \left(5\right)\end{array}$

where σ.sub.NI.sup.2 is the ideal interference plus noise power. In the practical systems, σ.sub.NI.sup.2 and {circumflex over (P)}.sub.NI has the following approximation relation

σ.sub.NI.sup.2≈K{circumflex over (P)}.sub.NI,   (6)

based on the CSI estimation process. For the channel gain,

[00005]$\underset{t=1}{\overset{2}{.Math.}}.Math..Math.\underset{m=1}{\overset{M}{.Math.}}.Math..Math.\underset{i\in \Omega }{.Math.}.Math..Math.{.Math.H_{m,k,i}^t.Math.}^2,$

it is approximated as

[00006]$\begin{array}{cc}\underset{t=1}{\overset{2}{.Math.}}.Math..Math.\underset{m=1}{\overset{M}{.Math.}}.Math..Math.\underset{i\in \Omega }{.Math.}.Math..Math.{.Math.H_{m,k,i}^t.Math.}^2\approx \underset{t=1}{\overset{2}{.Math.}}.Math..Math.\underset{m=1}{\overset{M}{.Math.}}.Math..Math.\underset{i\in \Omega }{.Math.}.Math..Math.{.Math.{\hat{H}}_{m,k,i}^t.Math.}^2-2.Math..Math.M.Math..Math.\Omega .Math..Math.{\hat{P}}_{\mathrm{NI}}.& \left(7\right)\end{array}$

Hence, the SINR can be estimated as

[00007]$\begin{array}{cc}{k}_{\mathrm{SU},\Omega }^{}=\frac{\underset{t=1}{\overset{2}{.Math.}}.Math..Math.\underset{m=1}{\overset{M}{.Math.}}.Math..Math.\underset{i\in \Omega }{.Math.}.Math..Math.{.Math.{\hat{H}}_{m,k,i}^t.Math.}^2-2.Math..Math.M.Math..Math.\Omega .Math..Math.{\hat{P}}_{\mathrm{NI}}}{2.Math..Math.\Omega .Math..Math.K.Math.{\hat{P}}_{\mathrm{NI}}}& \left(8\right)\end{array}$

on the subcarrier set |Ω|.

[0021] When used to approximate the uplink data SINR on the subcarrier set Ω, some adjustments have to made since the transmit power of data may different from that of the SRS, e.g., .sub.k.sup.UL,Ω=D.sup.UL.sub.k.sup.SU,Ω.

[0022] Since there exists unavoidable interference and noise in the estimated channel coefficients, the CSI error level is used to measure how much the estimated channel vector of a UE deviates from the ideal vector. More specifically, the CSI error level is defined as normalized correlation between the ideal and estimated channel vector

[00008]$\begin{array}{cc}{\alpha }_k=E_i\left[\frac{.Math.{\hat{H}}_{k,i}^{t,H}.Math.H_{k,i}^t.Math.}{{.Math.{\hat{H}}_{k,i}^t.Math.}_2.Math.{.Math.H_{k,i}^t.Math.}_2}\right],& \left(9\right)\end{array}$

where H.sub.k,i.sup.t and Ĥ.sub.k,i.sup.t are the ideal and estimated channel vectors defined by Ĥ.sub.k,i.sup.t=[Ĥ.sub.1,k,i.sup.t . . . Ĥ.sub.M,k,i .sup.t].sup.T and H.sub.k,i.sup.t=[H.sub.1,k,i.sup.t . . . H.sub.M,k,i.sup.t].sup.T respectively. With (2) and (3), α.sub.k is estimated as

[00009]$\begin{array}{cc}{\alpha }_k={\left(\frac{\underset{m=1}{\overset{M}{.Math.}}.Math..Math.{\hat{P}}_S^{m,k}}{M.Math.{\hat{P}}_{\mathrm{NI}}+\underset{m=1}{\overset{M}{.Math.}}.Math..Math.{\hat{P}}_S^{m,k}}\right)}^{1/2}& \left(10\right)\end{array}$

With α.sub.k, the BS can accommodate the effect of non-ideal CSI when select the MCS for each UE.

[0023] Although the foregoing descriptions of the preferred embodiments of the present inventions have shown, described, or illustrated the fundamental novel features or principles of the inventions, it is understood that various omissions, substitutions, and changes in the form of the detail of the methods, elements or apparatuses as illustrated, as well as the uses thereof, may be made by those skilled in the art without departing from the spirit of the present inventions. Hence, the scope of the present inventions should not be limited to the foregoing descriptions. Rather, the principles of the inventions may be applied to a wide range of methods, systems, and apparatuses, to achieve the advantages described herein and to achieve other advantages or to satisfy other objectives as well.