SCALABLE LOW OUTPUT IMPEDANCE BANDGAP REFERENCE WITH CURRENT DRIVE CAPABILITY AND HIGH-ORDER TEMPERATURE CURVATURE COMPENSATION

Abstract

A bandgap reference circuit includes a circuit for high-order temperature curvature compensation; and a circuit for low output impedance and current drive capability, wherein an output voltage of the bandgap reference circuit can be independently adjusted to be either above or below a silicon bandgap voltage without impacting temperature curvature.

Claims

1. A bandgap reference circuit, comprising: a conventional bandgap circuit that comprises two bipolar junction transistors and three resistors; an operational amplifier connected to the conventional bandgap circuit; a transconductance amplifier connected to the conventional bandgap circuit; a circuit for high-order temperature curvature compensation; and a circuit for low output impedance and current drive capability, wherein a current of the transconductance and a gain of the transconductance determine an output voltage of the bandgap reference circuit, which can be independently adjusted to be either above or below a silicon bandgap voltage without impacting temperature curvature.

2. A bandgap reference circuit, comprising: a conventional bandgap circuit that comprises two bipolar junction transistors and three resistors having a first input, a second input, and a first output connected to ground; an operational amplifier having a third input connected to the first input of the conventional bandgap circuit, a fourth input connected to the second input of the conventional bandgap circuit, and a second output; a buffer circuit having a fifth input connected to the second output of the operational amplifier, and a third output; a temperature curvature compensation circuit comprising a third bipolar junction transistor and two resistors having a sixth input connected to the first input of the conventional bandgap circuit, a seventh input connected to the second input of the conventional bandgap circuit, an eighth input, and a fourth output connected to ground; and a transconductance amplifier having a ninth input connected to the third output of the buffer circuit, a tenth input connected to ground, a fifth output connected to the first input of the conventional bandgap circuit, a sixth output connected to the second input of the conventional bandgap circuit, and a seventh output connected to the eighth input of the temperature compensation circuit.

3. The bandgap reference circuit of claim 2, wherein the operational amplifier is implemented as a single or two-stage amplifier, a folded-cascode, or a telescope cascode amplifier.

4. The bandgap reference circuit of claim 2, wherein the operational amplifier is implemented with degeneration resistors in current sources to reduce an input-referred offset voltage.

5. The bandgap reference circuit of claim 2, wherein the operational amplifier has a low output impedance.

6. The bandgap reference circuit of claim 2, wherein the operational amplifier has an input pair comprising an NMOS transistor, a PMOS transistor, an npn transistor, a pnp transistor, a FinFET transistor, or a combination thereof.

7. The bandgap reference circuit of claim 2, wherein the two bipolar junction transistors comprise an npn or a pnp transistor.

8. The bandgap reference circuit of claim 2, wherein the resistors comprise a silicided poly resistor, an unsilicided poly resistor, a diffusion resistor, a well resistor or a combination thereof.

9. The bandgap reference circuit of claim 2, wherein the conventional bandgap circuit is directly connected to the operational amplifier or through a resistor divider, wherein one end of the resistor divider is connected to ground and one end is connected to the conventional bandgap circuit output, wherein an output of the resistor divider is connected to the operational amplifier input.

10. The bandgap reference circuit of claim 2, wherein the transconductance amplifier converts a reference voltage to three separate currents.

11. The bandgap reference circuit of claim 2, wherein the transconductance amplifier is implemented with degeneration resistors in current sources to reduce an input-referred offset voltage.

12. The bandgap reference circuit of claim 2, wherein the buffer can be implemented as a native NMOS transistor, wherein a gate of the NMOS transistor is connected to the output of the operational amplifier.

Description

BRIEF DESCRIPTION OF DRAWINGS

[0011] The appended drawings illustrate several embodiments of the invention and are not to be considered limiting of its scope, for the invention may admit to other equally effective embodiments.

[0012] FIG. 1 shows a schematic block level circuit diagram of a prior art bandgap voltage reference circuit with high-order temperature curvature compensation.

[0013] FIG. 2 shows a schematic block level circuit diagram of a prior art bandgap voltage reference circuit with low output impedance.

[0014] FIG. 3 shows a schematic block level circuit diagram of a bandgap voltage reference circuit with both high-order temperature curvature compensation and low output impedance according to one embodiment of the present invention.

[0015] FIG. 4 shows an example Gummel plot of the bipolar junction transistor and a recommendation as to where to bias it according to one embodiment of the present invention.

[0016] FIG. 5 shows an example circuit diagram of a transconductance amplifier, according to one embodiment of the present invention, that can be used in the bandgap voltage reference circuit of FIG. 3.

[0017] FIG. 6 shows an example circuit diagram of an operational amplifier, according to one embodiment of the present invention, that can be used in the bandgap voltage reference circuit of FIG. 3.

[0018] FIG. 7 shows the simulated output voltage of the bandgap voltage reference of FIG. 3 versus temperature.

[0019] FIG. 8 shows the simulated output voltage of the bandgap voltage reference of FIG. 3 versus load current.

DETAILED DESCRIPTION

[0020] Aspects of the present disclosure are shown in the above-identified drawings and are described below. In the description, like or identical reference numerals are used to identify common or similar elements. The drawings are not necessarily to scale, and certain features may be shown exaggerated in scale or in schematic in the interest of clarity and conciseness.

[0021] Embodiments of the invention relate to a bandgap voltage reference circuit with high-order temperature curvature compensation and low output impedance. In one or more embodiments of the invention, a transconductance amplifier is used to provide three currents with low dependence on temperature to improve the temperature dependency of the output voltage. In one or more embodiments of the invention, an operational amplifier with a low output impedance is used to reduce the overall output impedance of the bandgap voltage reference circuit and allow supporting load or leakage currents via the reference output voltage. Without embodiments of the invention, a bandgap voltage reference circuit cannot provide an output reference voltage that remains constant in the presence of both temperature variations and load or leakage currents taken from the reference output voltage. Those skilled in the art, with the benefit of this disclosure will appreciate that same or similar features are equally applicable to any system whose operation requires high-precision and low output impedance.

[0022] In one or more embodiments, the bandgap voltage reference circuit can be implemented on a microchip, such as a semiconductor integrated circuit or can be implemented out of discrete components. In one or more embodiments, the bandgap reference circuit can optionally use an output capacitor. Throughout this disclosure, the terms bandgap circuit, bandgap reference circuit, bandgap voltage reference circuit, and voltage reference may be used interchangeably depending on the context.

[0023] FIG. 3 shows a schematic block-level diagram of a bandgap voltage reference circuit with high-order temperature curvature compensation and low output impedance in accordance with one embodiment of the invention. The circuit comprises a conventional bandgap circuit made of two bipolar junction transistors Q.sub.1 (301) and Q.sub.2 (302), and the three resistors R.sub.1 (303) and R.sub.2 (i.e. (304) and (305)). The bandgap voltage reference circuit also comprises an operational amplifier (306), which may be optionally followed by a buffer (308), and a transconductance amplifier (307). The transconductance amplifier (307) produces three currents that have very low dependency on temperature variations. Two of the currents are used in the conventional bandgap circuit, while the third is used for a third bipolar junction transistor Q.sub.3 (311). Transistor Q.sub.3 (311) and the two R.sub.NL resistors (312) and (313) are used to cancel the non-linear temperature dependence of the CTAT currents flowing in the R.sub.2 resistors (304) and (305) yielding an almost constant current and voltage reference with respect to temperature variations. This is necessary for high-precision applications, such as when temperature coefficients lower than 50 ppm/ C. are needed over a wide temperature range (e.g., from 25 C. to 125 C.). With the output voltage reference being the output of the operational amplifier, a low output impedance can be guaranteed at the output of the bandgap voltage reference circuit. Accordingly, the output of the circuit can support load or leakage currents or can provide an output current. The transconductance amplifier preferably should have a linear relationship between its input voltage and output current that has very small dependence on temperature.

[0024] The architecture in FIG. 3 can generate a reference output voltage V.sub.REF of any value independent of the silicon bandgap voltage. The reference voltage is equal to

[00001]$V_{\mathrm{REF}}.Math.=\frac{I_o}{G_m},$

where I.sub.0 is the output current of the transconductance amplifier and G.sub.m is its transconductance gain. I.sub.0 is equal to I.sub.VEB+I.sub.PTAT+I.sub.NL, with

[00002]$I_{\mathrm{VEB}}.Math.=\frac{V_{\mathrm{EB}.Math..Math.1}}{R_2},I_{\mathrm{PTAT}}.Math.=\frac{V_T.Math.\ln \left(N\right)}{R_1},\mathrm{and}.Math..Math.I_{\mathrm{NL}}=\frac{V_{\mathrm{EB}.Math..Math.1}-V_{\mathrm{EB}.Math..Math.3}}{R_{\mathrm{NL}}}.$

V.sub.EBi is the emitter-base voltage of the bipolar junction transistor Q.sub.i. V.sub.T is the thermal voltage that is equal to

[00003]$\frac{\mathrm{kT}}{q}$

with k being the Boltzmann's constant, T is the absolute temperature, and q is the electron charge. N is the ratio of the emitter area of the bipolar junction transistor Q.sub.2 to that of Q.sub.1. The reference voltage is thus given as

[00004]$V_{\mathrm{REF}}=\frac{1}{G_m}.Math.\left(\frac{V_T.Math.\ln \left(N\right)}{R_1}+\frac{V_{\mathrm{EB}.Math..Math.1}}{R_2}+\frac{V_{\mathrm{EB}.Math..Math.1}-V_{\mathrm{EB}.Math..Math.3}}{R_{\mathrm{NL}}}\right).$

The emitter-base voltages can be written as

[00005]$V_{\mathrm{EB}.Math..Math.3}=V_{\mathrm{BG}}-\left(V_{\mathrm{BG}}-V_{\mathrm{EB}.Math..Math.0}\right).Math.\frac{T}{T_0}-.Math..Math.V_T.Math.\ln .Math..Math.\left(\frac{T}{T_0}\right),\mathrm{and}$$V_{\mathrm{EB}.Math..Math.1,2}=V_{\mathrm{BG}}-\left(V_{\mathrm{BG}}-V_{\mathrm{EB}.Math..Math.0}\right).Math.\frac{T}{T_0}-\left(-1\right).Math..Math.V_T.Math.\ln .Math..Math.\left(\frac{T}{T_0}\right).$

The difference in the equations for V.sub.EB3 and V.sub.EB1,2 results from the fact that the current in Q.sub.3 is almost constant with temperature, whereas that in Q.sub.1 and Q.sub.2 is proportional to absolute temperature (PTAT). is a technology parameter that depends on the bipolar structure, V.sub.EB0 is the emitter-base voltage at temperature T.sub.0, and V.sub.BG is the silicon bandgap voltage. Choosing

[00006]$R_{\mathrm{NL}}=\frac{R_2}{-1}$

gives

[00007]$V_{\mathrm{REF}}=\frac{1}{G_m}.Math.\left(\frac{V_T.Math.\ln \left(N\right)}{R_1}+\frac{V_{\mathrm{BG}}-\left(V_{\mathrm{BG}}-V_{\mathrm{EB}.Math..Math.0}\right).Math.\frac{T}{T_0}}{R_2}\right)$

where the non-linear term in temperature is cancelled.

[0025] From the above equation, R.sub.1 may be chosen to determine the current flowing through the bipolar junction transistors and thus the overall power consumption. R.sub.1 is usually chosen to make the current fall in the flat region of the bipolar junction transistor Gummel plot as shown in FIG. 4. In FIG. 4, an exemplary usable region is shown to span from around 0.7 A to around 1.8 A. However, depending on the applications, the usable region (i.e., the flat region) may be extend to beyond 2 A, such as 3 A or higher. R.sub.2 may be used to adjust the output voltage curvature with respect to temperature variations and can be made programmable for curvature trimming. G.sub.m can be used to adjust the exact value of the output voltage reference and can also be made programmable for accuracy trimming.

[0026] FIG. 5 shows a possible implementation of the transconductance amplifier (307) that may be used in the bandgap reference circuit of FIG. 3. The input voltage is applied across the resistors R.sub.4 (i.e., (501) and (502)). These resistors should be matched with all the resistors in FIG. 3 in construction and the unit resistor used. The relative sizing of transistors (509) to (510) set the current ratio between I.sub.1 and I.sub.2. Making transistors (503) and (504) equal in size makes I.sub.x equal to I.sub.y. Applying Kirchoff s current law at nodes X and Y can thus yield a relation of the currents I.sub.1 and I.sub.2 in terms of the input voltage and the resistors values. Branches (511) and (512) are used for biasing cascode transistors. Branches (513) and (514) generate the required currents for the bipolar junction transistors Q.sub.1 (301) and Q.sub.2 (302) and the resistors R.sub.1 (303) and R.sub.2 (i.e., (304) and (305)) of FIG. 3, while branch (515) generates the current for the bipolar junction transistor Q.sub.3 (311) of FIG. 3 used for high-order temperature curvature compensation. The relative sizes of these branches can be used to set the final value of the transconductance gain. Degeneration resistors (516-522) are used to improve the matching of the current sources and reduce the input-referred offset voltage of the transconductance amplifier.

[0027] The transconductance gain of the transconductance amplifier shown in FIG. 5 is given by

[00008]$I_o=\frac{{\mathrm{kV}}_{\mathrm{in}}}{\left(N-1\right).Math.R_4},$

wherein k is the ratio of the sizes of the transistors in branches (513), (514), and (515) to transistors (507) and (509), and N is the ratio of the size of transistors (508) and (510) to transistors (507) and (509), respectively. With this implementation, the transconductance gain is inversely proportional to the value of R.sub.4, which can be matched to the other resistors in the bandgap voltage reference circuit in construction and the unit resistor used. R.sub.4 can then be used for accuracy trimming.

[0028] While FIG. 5 illustrates one implementation of a transconductance amplifier that can be used with embodiments of the invention, one skilled in the art would appreciate that this example is for illustration only and that other variations and modifications are possible without departing from the scope of the invention.

[0029] FIG. 6 shows a possible implementation of the operational amplifier (306) and the following buffer (308) used in the bandgap voltage reference circuit of FIG. 3. The operational amplifier (306) comprises a folded-cascode amplifier with a PMOS differential pair input (601) and (602). The buffer (308) may be implemented as a source follower (603). The folded-cascode amplifier provides the necessary gain for the operation of the bandgap voltage reference circuit and guarantees the required accuracy. The source follower (603) provides low impedance at the output node and can also provide any necessary load/leakage current. The degeneration resistors (604), (605), (606), and (607) are used to improve matching of current sources and thus reduce the input-referred offset voltage of the operational amplifier (306) in FIG. 3.

[0030] While FIG. 6 illustrates one implementation of an operational amplifier followed by a buffer that can be used with embodiments of the invention, one skilled in the art would appreciate that this example is for illustration only and that other variations and modifications are possible without departing from the scope of the invention. For example, the operational amplifier can be implemented as a single or two-stage amplifier, a folded-cascode, or a telescope cascode amplifier. In addition, while the above example uses PMOS transistors to implement a folded-cascode amplifier, other implementations are possible, such as NMOS transistors, PMOS transistors, npn transistors, pnp transistors, FinFET transistors or a combination thereof. Another example is the buffer, which can also be implemented as another operation amplifier in a unity feedback configuration, or it can be implemented by a native NMOS device

[0031] FIG. 7 shows the simulated output voltage of the bandgap voltage reference circuit of FIG. 3 as a function of temperature. Simulations are done while an output load current is being drawn out of the circuit. Plot (701) shows the output voltage when the bipolar junction transistor Q.sub.3 (311) and resistors R.sub.NL (i.e., (312) and (313)) are not used. Without high-order temperature curvature compensation, a variation in the output voltage of about 2 mV is observed over a temperature range of 25 C. to 125 C. This may not be suitable for some high-precision applications. Plot (702) shows the same output voltage when the bipolar junction transistor Q.sub.3 (311) and resistors R.sub.NL (i.e. (312) and (313)) are used. This enables the high-order temperature curvature compensation and as a result, a variation in the output voltage of only 200 V is observed over the same temperature range of 25 C. to 125 C. This is better for high-precision applications.

[0032] FIG. 8 shows a plot of the output voltage of the bandgap voltage reference circuit of FIG. 3 as a function of load/leakage current at room temperature with the vertical axis limited to only 1% of the nominal reference voltage. As can be seen, the output voltage V.sub.REF only shows a total variation of 2 mV or 0.2% of the nominal reference voltage over load/leakage current variation of up to 400 A. If more load/leakage current is to be expected, the size of the source follower can be adjusted accordingly.

[0033] The results shown in FIGS. 7 and 8 demonstrate that embodiments of the invention can provide accurate reference voltages that are insensitive to temperature variations and load/leakage current variations. Therefore, embodiments of the invention are suitable for high-precision applications. In addition, an output voltage of a bandgap reference circuit of the invention can be adjusted to be either above or below the silicon bandgap voltage (the golden voltage around 1.2 V) without impacting the temperature curvature.

[0034] While the invention has been described with respect to a limited number of embodiments, those skilled in the art, having the benefit of this disclosure, will appreciate that other embodiments can be devised which do not depart from the scope of the invention as disclosed herein. Accordingly, the scope of the invention should be limited only by the attached claims.