Abstract
An AC-to-DC converter circuit includes DC-to-DC converter that in turn includes a secondary side circuit. The secondary side circuit includes a secondary winding, a pair of bipolar transistor-based self-driven synchronous rectifiers, a pair of current splitting inductors, and an output capacitor. Each of the synchronous rectifiers includes a bipolar transistor and a diode whose anode is coupled to the transistor collector and whose cathode is coupled to the transistor emitter. The current splitting inductors provide the necessary base current to the bipolar transistors at the appropriate times such that the bipolar transistors operate as synchronous rectifiers. As compared to using conventional self-driven synchronous rectifiers based on field effect transistors in the secondary side, using the novel bipolar-transistor based synchronous rectifiers in the secondary side of the forward converter circuit results in lower power consumption and allows the converter to operate from a wider range of VAC input voltages.
Claims
1. A method comprising: providing a pair of self-driven BJT (Bipolar Junction Transistor) synchronous rectifiers and a pair of inductors in a secondary side of a switching converter circuit, wherein each of the self-driven BJT synchronous rectifiers includes a bipolar transistor and a diode disposed in parallel with the bipolar transistor such that the anode of the diode is coupled to a collector of the bipolar transistor and such that a cathode of the diode is coupled to an emitter of the bipolar transistor, wherein a base terminal of a first bipolar transistor of a first synchronous rectifier is connected to a base terminal of a second bipolar transistor of a second synchronous rectifier, wherein the pair of inductors comprises a first inductor having a first end and a second end and a second inductor having a first end and a second end, wherein the second end of the first inductor is coupled to a first collector of the first bipolar and a second collector of the second bipolar transistors, wherein the first end of the second inductor is coupled to the first end of the first inductor, and wherein the second end of the second inductor is coupled to the base terminal of the first bipolar transistor and the base terminal of the second bipolar transistor.
2. A circuit comprising: a transformer winding having a center tap; and means for conducting a charging current through the transformer winding during a first portion of a switching cycle and then for conducting a free-wheeling current during a second portion of the switching cycle such that a forward voltage drop across the means when the means is conducting current during the first and second portions of the switching cycle is substantially less than 0.7 volts on average, and wherein the means includes a pair of input current terminals and no field effect transistor, and an output capacitor having a first terminal connected to the center tap and a second terminal connected to the pair of input current terminals, wherein the transformer winding is a secondary winding of a transformer, and wherein the means comprises a pair of current splitting inductors connected to the pair of input current terminals, respectively, two bipolar transistors, and two diodes, wherein a current splitting inductor of the pair of current splitting inductors comprises a first end, connected to the second terminal of the output capacitor, and a second end connected to a base terminal of a first bipolar transistor of the two bipolar transistors, and a base terminal of a second bipolar transistor of the two bipolar transistors.
3. The circuit of claim 2, wherein the first bipolar transistor and a first diode are connected to form a first synchronous rectifier and the second bipolar transistor and a second diode are connected to form a second synchronous rectifier.
4. The circuit of claim 2, wherein an emitter of the first bipolar transistor is connected to a cathode of a first diode and an emitter of the second bipolar transistor is connected to a cathode of a second diode.
5. The circuit of claim 2, wherein a collector of the first bipolar transistor is connected to anode of a first diode and a collector the second bipolar transistor is connected to an anode of a second diode.
6. The circuit of claim 2, wherein collectors of the first bipolar transistor and second bipolar transistor are connected to a second current input terminal.
7. The circuit of claim 2, further comprising: a primary transformer winding; and means for switching current flow in the primary transformer winding at a frequency of at least ten kilohertz.
8. The circuit of claim 7, wherein the circuit comprises an AC-to-DC forward converter that receives a 110 volt AC supply voltage, and wherein the means for switching has a breakdown withstand voltage rating of not more than 300 volts.
9. The circuit of claim 8, wherein the AC-to-DC forward converter is operable from either the 110 volt AC supply voltage or a 220 volt AC supply voltage.
10. The circuit of claim 2, wherein the first bipolar transistor and a first diode are parts of a first semiconductor die, and wherein the second bipolar transistor and a second diode are parts of a second semiconductor die.
11. The circuit of claim 2, wherein the first bipolar transistor has an emitter-to-collector breakdown withstand voltage that exceeds twenty volts and an emitter-to-base reverse breakdown withstand voltage that exceeds twenty volts, and wherein the second bipolar transistor has an emitter-to-collector breakdown withstand voltage that exceeds twenty volts and an emitter-to-base reverse breakdown withstand voltage that exceeds twenty volts.
12. The circuit of claim 2, wherein a first inductor has an inductance, wherein a second inductor has an inductance, and wherein the inductance of the second inductor is at least five times the inductance of the first inductor.
Description
BRIEF DESCRIPTION OF THE DRAWINGS
(1) The accompanying drawings, where like numerals indicate like components, illustrate embodiments of the invention.
(2) FIG. 1 (Prior Art) is a circuit diagram of an AC-to-DC converter circuit that includes a forward converter.
(3) FIG. 2 (Prior Art) is a simplified diagram that illustrates the relationship of the primary, secondary and tertiary windings of the transformer of the AC-to-DC converter circuit of FIG. 17.
(4) FIG. 3 (Prior Art) is a diagram that illustrates the waveform of the sinusoidal input supply voltage V.sub.S supplied to the AC-to-DC converter circuit of FIG. 1.
(5) FIG. 4 (Prior Art) is a circuit diagram of an AC-to-DC converter circuit that includes a forward converter circuit.
(6) FIG. 5 (Prior Art) is a waveform diagram that illustrates the operation of the AC-to-DC converter circuit of FIG. 4.
(7) FIG. 6 is a diagram of an AC-to-DC converter circuit in accordance with one novel aspect.
(8) FIG. 7 is a table that sets forth relative performance parameters of the prior art AC-to-DC converter circuit of FIG. 2 as compared to the novel AC-to-DC converter circuit of FIG. 6.
(9) FIG. 8 is a waveform diagram that illustrates the operation of the prior art AC-to-DC converter circuit of FIG. 4 in more detail.
(10) FIG. 9 is a simplified cross-sectional diagram of a field effect transistor and a diode of the AC-to-DC converter circuit 23 of FIG. 4.
(11) FIG. 10 is a diagram that illustrates the drain current I.sub.D to drain-to-source voltage V.sub.DS of the field effect transistor structure of FIG. 9.
(12) FIG. 11 shows a conventional AC-to-DC converter circuit that includes a forward converter, and also shows how the conventional circuit can be modified without changing its operation.
(13) FIG. 12 is a circuit diagram that shows the resulting circuit after the diode D2 and the inductor L1 have been moved as indicated by arrows in FIG. 11.
(14) FIG. 13 shows how a pair of parallel-connected inductors can split a current flowing between two nodes into two currents I.sub.1 and I.sub.2.
(15) FIG. 14 is a diagram showing a rough equivalence between a first circuit involving inductor coupled in series with a diode and a second circuit involving two current-splitting inductors, a bipolar transistor, and a diode.
(16) FIG. 15 is a diagram of a circuit 131 that is a rough equivalent of, and can therefore replace, the circuit 130 of FIG. 12.
(17) FIG. 16 is a diagram showing the AC-to-DC converter circuit of FIG. 12 with the circuitry enclosed by the dashed line 130 having been replaced by the circuitry 131 of FIG. 15.
(18) FIG. 17 is a waveform diagram that illustrates operation of the novel AC-to-DC converter circuit of FIG. 6.
(19) FIG. 18 (Prior Art) is a diagram of a conventional AC-to-DC converter circuit that includes a two-switch forward converter.
(20) FIG. 19 is a simplified circuit diagram of a novel AC-to-DC converter circuit that includes a two-switch forward converter in accordance with another novel aspect.
(21) FIG. 20 is a waveform diagram that illustrates an operation of the novel AC-to-DC converter circuit of FIG. 19.
(22) FIG. 21 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit involving a push-pull converter.
(23) FIG. 22 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 300 involving a half-bridge converter.
(24) FIG. 23 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 400 involving a full-bridge converter.
(25) FIG. 24 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 500 involving a phase shift full bridge converter.
(26) FIG. 25 is a diagram illustrating one-sided transformer core magnetization.
(27) FIG. 26 is a diagram illustrating two-sided transformer core magnetization.
(28) FIG. 27 is a circuit diagram of a novel secondary side circuit that can be used in any one of the circuits of FIGS. 21-24 in accordance with another novel aspect.
(29) FIG. 28 is a waveform diagram that shows currents and voltages in the novel secondary side circuit of FIG. 27.
DETAILED DESCRIPTION
(30) Reference will now be made in detail to some embodiments of the invention, examples of which are illustrated in the accompanying drawings.
(31) FIG. 6 is a diagram of an AC-to-DC converter circuit 50 in accordance with one novel aspect. The AC-to-DC circuit 50 includes a full-bridge rectifier 52, a power factor correction circuit 53, an input capacitor 54, a main switch 55, a switch driver circuit 56, a transformer 57, a demagnetization path diode D1 58, a first bipolar transistor Q2 59, a first diode D2 60, a second bipolar transistor Q3 61, a second diode D3 62, a first inductor L1 63, a second inductor L2 64, and an output capacitor 65. Reference numeral 66 identifies an output voltage node of the secondary side of the AC-to-DC circuit 50. Reference numeral 67 identifies a ground node of the secondary side of the AC-to-DC circuit 50. Reference number 85 identifies a base node. The AC-to-DC converter circuit 50 receives a 110 volt RMS AC input supply voltage V.sub.S 68, and supplies a load 69 with a 2.5 volt DC output voltage V.sub.O 70. Transformer 57 includes a primary winding 71 having N.sub.1 turns, a secondary winding 72 having N.sub.2 turns, and a tertiary winding 73 having N.sub.3 turns. The switch driver circuit 56 has a startup power connection (not shown) and a power connection to the output voltage of the secondary side (not shown). Details of the switch driver circuit 56 and output regulation circuitry are not shown. The main switch 55 is controlled by the switch drive circuit 56 to turn on and to turn off at a frequency of approximately ten kilohertz or more.
(32) The AC-to-DC circuit 50 includes a DC-to-DC forward converter circuit. The DC-to-DC forward converter circuit includes all the components of the AC-to-DC converter circuit 50 but for the full-bridge rectifier 52 and the optional power factor correction circuit 53.
(33) In the present example, bipolar transistor Q2 59 and diode D2 60 are provided on the same semiconductor die 74. Bipolar transistor Q3 61 and diode D3 62 are provided on the same semiconductor die 75. The emitter of bipolar transistor 59 is on the substrate side of die 74 and the emitter of bipolar transistor 61 is on the substrate side of die 75. As the AC-to-DC circuit 50 operates, there will be different voltages on the emitters of the two bipolar transistors. The bipolar transistors are therefore provided as two different dice in the illustrated example. Unlike the case of a field effect transistor where a body diode is inherently present, a diode is not inherently present in a bipolar transistor die structure. Each of dice 74 and 75 is a special RBJT (Reverse Bipolar Junction Transistor) integrated circuit die that incorporates both a bipolar transistor as well as a parallel-coupled distributed diode. The RBJT and distributed diode integrated circuit die has emitter-to-collector and emitter-to-base reverse breakdown withstand voltages that exceed twenty volts. For additional information on a suitable RBJT and distributed diode integrated circuit die that is usable in a forward converter with a self-driven BJT synchronous rectifier, see: U.S. patent application Ser. No. 13/299,340, entitled Bipolar Junction Transistor For Current Driven Synchronous Rectifier, filed Nov. 17, 2011, by Kyoung Wook Seok (the entire subject matter of which is incorporated herein by reference).
(34) As compared to a field effect transistor that is a voltage driven device, a bipolar transistor is current driven device in that a base current is driven into or out of the base to make a collector-to-emitter current flow through the transistor. To turn off the transistor, the base current is stopped. To turn on an NPN bipolar transistor of a synchronous rectifier in the circuit of FIG. 6, a relatively large base current is driven into the base in order to cause the transistor to saturate and to conduct a current between its collector and emitter terminals with a relatively low V.sub.CE. The base currents for controlling the bipolar transistors 59 and 61 are supplied via base node 85 by inductor L2 64. The magnitude of the combined base currents (supplied by inductor L2 as current I.sub.L2) is set to be a proportion of the inductor current I.sub.L1 flowing through inductor L1 63, where this proportion is determined by the fixed ratio of the inductance of inductor L1 63 to the inductance of inductor L2 64. The forward converter of the AC-to-DC circuit 50 of FIG. 6 is therefore said to have self-driven BJT (Bipolar Junction Transistor) synchronous rectifiers.
(35) FIG. 7 is a table that sets forth relative performance parameters of the prior art AC-to-DC converter circuit 23 of FIG. 4 as compared to the novel AC-to-DC converter circuit 50 of FIG. 6. The novel AC-to-DC converter circuit 50 is approximately three percent more efficient as compared to the prior art AC-to-DC converter circuit 23 of FIG. 4. This three percent improvement assumes an L1 peak current of 80 A and a valley of 10 A in the prior art AC-to-DC converter circuit 23. The average inductor current is therefore about 80 A plus 10 A divided by two, or 45 A. The output power, assuming an output voltage V.sub.O of 2.5 volts DC and an average output current of 45 A, is 135 W. If the input power is 156 W, then the efficiency of the prior art AC-to-DC converter circuit 23 is about 87 percent. In prior art circuit of FIG. 2, the Q3 transistor stops working as a switch at about 40 A, as explained in further detail below. The body diode of the Q3 transistor conducts the free-wheeling current from 40 A down to about 10 A. So during the time of the diode free-wheeling operation, which is approximately 30 percent of the switching period, the average current is 25 A. The average current of 25 A is estimated as the average of 40 A and 10 A. The power loss is therefore about 7.5 W, assuming a 1.0 volt voltage drop across the body diode of the free-wheeling field effect transistor. Power loss across the free-wheeling field effect transistor is therefore 7.5 W/135 W, or 5.6 percent. By employing the bipolar transistors, the distributed diodes, and the current-splitting inductors of the novel AC-to-DC converter circuit of FIG. 6, the 7.5 W power loss across the body diodes of the conventional circuit can be reduced to about 1.5 W, which is a 6 W reduction. The input power received into the novel AC-to-DC converter circuit is about 150 W. The efficiency of the novel AC-to-DC converter circuit 50 of FIG. 6 is therefore roughly 135 W/150 W, or 90 percent.
(36) Explanation of how and why the novel AC-to-DC forward converter circuit 50 of FIG. 6 is structured and operates is described in relation to the waveform diagram of FIG. 8. FIG. 8 is a waveform diagram that illustrates the operation of the prior art AC-to-DC forward converter circuit 23 of FIG. 4 in more detail. In the waveform labeled I.sub.L1 of FIG. 8, notice that there is current flowing through inductor L1 during time T.sub.idle. In the waveform labeled V.sub.Gs of FIG. 8, notice that the gate-to-source voltage of field effect transistor Q3 25 of FIG. 4 drops to about zero volts roughly at the beginning of time T.sub.idle. The low gate-to-source voltage on field effect transistor Q3 means that the field effect transistor Q3 is substantially turned off. For field effect transistor Q3 to be turned on and to have a reasonably low R.sub.DS(ON), transistor Q3 should have a gate voltage V.sub.Gs substantially greater than zero volts, yet the V.sub.Gs of transistor Q3 during the T.sub.idle period of time is about zero volts. Conduction of the inductor current I.sub.L1 of the I.sub.L1 waveform that flows from node 22 of the circuit of FIG. 4 to node 21 of the circuit of FIG. 4 therefore flows largely through the body diode 27 of the field effect transistor Q3 25.
(37) FIG. 9 is a simplified cross-sectional diagram of the field effect transistor Q3 25 and inherent body diode 27 of the prior art AC-to-DC converter circuit 23 of FIG. 4. The structure in includes a metal source electrode 76, a polysilicon gate 77, a gate insulator layer 78, an N type drift region 79, a P+ type body region 80, an N+ type source region 81, an N++ type substrate layer 82, and a metal drain electrode 83. Regions 79, 80 and 81 are regions of an epitaxial semiconductor layer disposed on the N++ type substrate 82. The inherent body diode 27 is the PN junction between the P+ body region 80 and the N type drift region 79. The diode symbol identified by reference numeral 27 schematically depicts this body diode junction. If the field effect transistor Q3 is turned on, then a conductive channel 84 is formed and current can flow from the source region 81, laterally through the channel 84, into the drift region 79, and then vertically down through the drift region 79, through the N++ type substrate layer 82, and to the drain metal electrode 83. The V.sub.SD voltage drop may, for example, be approximate 0.2 volts depending on the magnitude of the current flow and the magnitude of the gate voltage on the transistor structure. If the field effect transistor Q3 is turned off then current can still flow from the source metal electrode 76 to the drain metal electrode 83, but the voltage drop of the current path will be larger. Current can flow from the source metal electrode 76, through the P+ type body region 80, across the junction of body diode 27, into the N type drift region 79, down through the N++ type substrate layer 82, and to the drain metal electrode 83. The voltage drop across the forward biased inherent body diode 27 is, however, approximately one volt. Because power loss is equal to the product of the current flow and the voltage drop, flowing the current through the larger voltage drop of the body diode 27 is seen to consume more power than flowing the current through the smaller voltage drop across the channel if the transistor is on.
(38) FIG. 10 is a diagram that illustrates the drain current I.sub.D to drain-to-source voltage V.sub.DS of the field effect transistor structure of FIG. 9.
(39) As illustrated in the waveform labeled V.sub.SD in FIG. 8, there is a large source-to-drain voltage V.sub.SD across the transistor Q3 during the T.sub.idle time due to current flow through the inherent body diode 27 of the field effect transistor structure. Current flow across this V.sub.SD voltage drop gives rise to a substantial power loss Eliminating or reducing some or all of this power loss is a desired objective.
(40) FIG. 11 shows a well-known AC-to-DC converter circuit 100 that includes a forward converter. The AC-to-DC converter circuit 100 includes a full-bridge rectifier 101, a power factor correction circuit 102, an input capacitor 103, a main switch Q1 104, a gate drive circuit 105, a demagnetization current path diode D1 106, a transformer 107, a rectifier diode D2 108, a free-wheeling diode D3 109, an inductor 110, and an output capacitor 111. The AC-to-DC converter circuit 100 receives a 110 volt RMS AC input supply voltage V.sub.S 112, and supplies a load 113 with a 2.5 volt DC output voltage V.sub.O 114. Transformer 107 includes a primary winding 115 having N.sub.1 turns, a secondary winding 116 having N.sub.2 turns, and a tertiary winding 117 having N.sub.3 turns.
(41) First, it is recognized by the inventor that the location of diode 108 can be moved as indicated by the arrow 118 labeled EQUIVALENT to the location 119 indicated by the diode symbol shown in dashed lines. Also, it is recognized by the inventor that the location of the inductor L1 110 can be moved as indicated by the arrow 120 labeled EQUIVALENT to the location 121 indicated by the inductor symbol shown in dashed lines. FIG. 12 is a circuit diagram that shows the resulting circuit 100 after the diode D2 and the inductor L1 have been moved as indicated by arrows in FIG. 11.
(42) FIG. 13 shows how a pair of parallel-connected inductors can split a current flowing between two nodes 88 and 89 into two currents I.sub.1 and I.sub.2. The relative magnitudes of the two currents I.sub.1 and I.sub.2 are determined by the relative inductances of the two parallel-connected inductors. The inventor has also recognized that an inductor 122 and a diode 123 connected in series as shown on the left side of FIG. 14 has a rough equivalence with the circuit shown to the right in FIG. 14 involving the two inductors 124 and 125, the bipolar transistor 126, and the diode 127. If a collector-to-emitter forward current is to flow through the bipolar transistor 126, then an adequate base current is be supplied into the base of the bipolar transistor 126 to saturate the transistor. Because the voltage between the collector and the base of a conductive bipolar transistor is small, the voltages across the two inductors 124 and 125 are roughly equal. Because the voltages across the inductors 124 and 125 are roughly equal, the current-splitting relationship illustrated in FIG. 13 can be used to set the inductances of the two inductors 124 and 125 so that the magnitude of the injected base current 128 is a desired fraction (for example, one tenth) of the collector current 129 flowing into the collector of the bipolar transistor. In this example, the inductance of the second inductor 125 is at least five times greater (it may be, for example, ten times greater) than the inductance of the first inductor 124. The base current, when a synchronous rectifier is to be conducting current, needs to be adequately large to fully saturate the bipolar transistor of the synchronous rectifier.
(43) It is desired to use the equivalence set forth in FIG. 14 to replace the components in FIG. 12 that are circled by the dashed line 130. It is recognized that current only flows through one of the diodes 108 and 109 at a time. The inductor 110 can therefore be thought of as being time-shared between the two diodes. Accordingly, the circuit circled by the dashed line 130 in FIG. 12 can be replaced by the circuit 131 shown in FIG. 15. In the circuit 131 of FIG. 15, each diode is replaced by a bipolar transistor and a diode. Diode 108 is replaced by bipolar transistor 132 and diode 133. Diode 109 is replaced by bipolar transistor 134 and diode 135. The parallel inductors 136 and 137 of the substitution are, however, shared between the diode replacement circuits because as mentioned above only one of the two diode replacement circuits will conduct current at a given time.
(44) FIG. 16 is a diagram showing the AC-to-DC converter circuit of FIG. 12 after the circuitry in the dashed line 130 has been replaced by the circuitry 131 of FIG. 15. The resulting AC-to-DC converter circuit 100 of FIG. 16 is the same AC-to-DC converter circuit 50 as shown in FIG. 6, only in FIG. 16 different reference numerals are used to denote the various components due to the explained derivation of the circuit from the prior art circuit of FIG. 11 as explained above in connection with FIGS. 11-16.
(45) FIG. 17 is a waveform diagram that illustrates operation of the novel AC-to-DC converter circuit 50 of FIG. 6. As compared to the prior art operation shown in the waveforms of FIG. 8 where the field effect transistor Q3 in the conventional circuit is off during T.sub.idle, the bipolar transistor 61 in the novel circuit of FIG. 6 is not off during T.sub.idle. As indicated in the waveform labeled I.sub.L1 in FIG. 17, there is a substantial inductor current flowing through inductor L1 during T.sub.idle. This current flows through the free-wheeling synchronous rectifier involving bipolar transistor 61 and diode 62. Due to the current splitting function of inductors 63 and 64, a fixed percentage of the current I.sub.L1 is injected into the base of bipolar transistor Q3 61, thereby keeping bipolar transistor fully turned on and saturated throughout the majority of T.sub.idle.
(46) Toward the end of T.sub.idle, when the current I.sub.L1 stops at time T.sub.4, then the base current into bipolar transistor Q3 61 is so small that the distributed diode 62 conducts. The voltage drop across a forward biased diode 62 is typically a voltage larger than 0.7 volts. Power consumption is therefore larger after time T.sub.4 as a result of diode conduction. The waveform labeled V.sub.CE in FIG. 17 shows the V.sub.CE voltages across bipolar transistor Q2 59 and across bipolar transistor Q3 61. Despite the larger V.sub.CE toward the end of T.sub.idle, the duration of this higher power consumption at the end of T.sub.idle is relatively short. The loss power is therefore also relatively small.
(47) Likewise, at the very beginning of the switching cycle at time T.sub.1 the inductor current is small and has not yet increased to the point that the base current flowing into the bipolar transistor Q2 59 is sufficient to turn the bipolar transistor on. The V.sub.CE drop across the bipolar transistor Q2 is shown in the waveform labeled V.sub.CE in FIG. 17. Despite the larger V.sub.CE voltage drop across bipolar transistor Q2 shortly after time T.sub.1, the duration of this higher power consumption is relatively short. The loss of power is therefore correspondingly small.
(48) After the short period of a higher V.sub.CE following time T.sub.1, the V.sub.CE of bipolar transistor Q2 49 decreases from the higher V.sub.CE voltage that is slightly greater than 0.7 volts down to a lower forward voltage of about 0.2 volts. Then starting at time T.sub.2 and extending until just before the end of the switching cycle at time T.sub.4, the V.sub.CE of the bipolar transistor Q3 61 remains low at about 0.1 volts. Accordingly, as shown in the waveform labeled V.sub.CE, for most of the time that a synchronous rectifier (either Q2/D2 or Q3/D3) is to be on and conductive the voltage drops across the synchronous rectifiers is well below 0.7 volts. This low forward voltage drop across the synchronous rectifiers is favorable as compared to the higher V.sub.SD voltage drops across the synchronous rectifiers in the conventional circuit 23 of FIG. 4 as indicated by the waveform labeled V.sub.SD in FIG. 8.
(49) In addition to the reduced power loss advantage over the prior art of FIG. 2 as explained above, the novel AC-to-DC converter circuit 50 of FIG. 6 has other advantages. In the conventional circuit of FIG. 4 the secondary winding voltage V.sub.N2 between T.sub.2 and T.sub.3 is required to be about ten volts to keep field effect transistor Q3 fully turned on. In the novel AC-to-DC converter circuit 50 of FIG. 6, on the other hand, there is no requirement for the secondary winding voltage V.sub.N2 between T.sub.2 and T.sub.3 to keep a field effect transistor turned on, but rather the minimum acceptable value for V.sub.N2 is determined only by the amount of time available to reset the transformer core. Therefore the secondary winding voltage V.sub.N2 can be smaller in the novel circuit of FIG. 6 as compared to the conventional circuit of FIG. 4. Accordingly, the number of turns N.sub.3 of the tertiary winding can be increased as compared to the conventional circuit. In the waveform labeled V.sub.DS,Q1 of FIG. 17, the voltage V.sub.I during T.sub.idle is the 156 volt peak voltage of the incoming AC signal V.sub.S. The magnitude of the voltage 138 is given by (N.sub.1/N.sub.3)*V.sub.I. The main switch Q1 104 must have a high enough V.sub.DS breakdown withstand voltage to tolerate the maximum V.sub.DS with some margin. The V.sub.DS breakdown withstand voltage required for the main switch Q1 5 in the conventional AC-to-DC converter circuit 23 of FIG. 4 may, for example, be about 400 volts. In accordance with one novel aspect, the number of turns N.sub.3 is larger in the novel AC-to-DC converter circuit 50 of FIG. 6, thereby decreasing the voltage 138. As a consequence, the maximum V.sub.DS that the main switch Q1 104 will experience between times T.sub.2 and T.sub.3 is smaller, and a main switch Q1 104 having a lower V.sub.DS breakdown voltage rating (for example, 300 volts) can be employed. Note that the voltage 138 in the V.sub.DS,Q1 waveform of FIG. 17 is smaller than the corresponding voltage 139 in the waveform labeled V.sub.DS,Q1 of FIG. 8. Using a main switch that has a more relaxed V.sub.DS breakdown withstand voltage rating allows the main switch to have a lower R.sub.DS(ON) and this serves to reduce power loss in the novel forward converter circuit as compared to the conventional forward converter circuit.
(50) In addition to the advantages set forth above, there is also another advantage of the circuit of FIG. 6 over the conventional circuit of FIG. 4. The conventional circuit of FIG. 4 cannot operate from both either an incoming 110 volt AC supply voltage or an incoming 220 volt AC supply voltage. If the voltage of the incoming AC supply voltage were to be changed from 110 volts AC to 220 volts AC, then circuit components would have to be changed or the power supply would be damaged. The reason for the damage is that the gate voltage of field effect transistor Q2 in the conventional circuit of FIG. 4 at time T.sub.1 has to be adequately high (about ten volts) to ensure that the field effect transistor Q2 is fully on, but yet the gate voltage cannot ever be so high as to destroy the field effect transistor Q2. If the gate voltage (between times T.sub.2 and T.sub.3) is ten volts when the incoming supply voltage is 110 volts AC, then the gate voltage would be too high if the incoming supply voltage were increased to 220 volts AC. In contrast to the conventional circuit of FIG. 4, in the novel circuit 50 of FIG. 6 the secondary winding voltage V.sub.N2 between time T.sub.1 and time T.sub.2 does not control the gate voltage of a field effect transistor. In the novel circuit of FIG. 6, if the secondary winding voltage V.sub.N2 is high, the gate voltage on a field effect transistor will not be destroyed because the secondary winding voltage V.sub.N2 does not drive the gate of any field effect transistor. The synchronous rectifiers in the novel circuit of FIG. 6 do not involve field effect transistors, but rather involve bipolar transistors. As a result, the same power supply circuit of FIG. 6 can be operated from either an incoming supply voltage of 110 volts AC or an incoming supply voltage of 220 volts AC, without changing any circuit components in the power supply circuit of FIG. 6.
(51) FIG. 18 (Prior Art) is a simplified circuit diagram of a conventional AC-to-DC converter circuit 140 that includes a two-switch forward converter circuit. Rather than having just one main switch on the primary side of the converter, the two-switch topology has two main switches 141 and 142.
(52) FIG. 19 is a simplified circuit diagram of a novel AC-to-DC converter circuit 150 that includes a two-switch forward converter in accordance with another novel aspect. The primary side of the AC-to-DC converter circuit 150 includes a full-bridge rectifier 151, an optional power factor correction circuit 152, an input capacitor 153, a first main switch Q1 154, a second main switch Q2 155, a first diode D1 156, and a second diode D2 157, and the primary winding 159 of a transformer 158. The transformer 158 includes the primary winding 159 and a secondary winding 160. The secondary side of the novel AC-to-DC converter circuit 150 is the same as in the prior embodiment of FIG. 6. The secondary side circuit includes a third bipolar transistor 161, a third diode 162, a fourth bipolar transistor 163, a fourth diode 164, a first inductor L1 165, a second inductor L2 166, and an output capacitor 167, interconnected as shown in FIG. 19. In the illustrated example, the AC-to-DC converter circuit 150 receives a 110 volt RMS AC input supply voltage V.sub.S 168, and supplies a load 169 with a 2.5 volt DC output voltage V.sub.O 170. The switch driver circuitry that drives the main switches is not illustrated but can be conventional for this type of two-switch forward converter topology. In a typical forward converter involving a single main switch, the drain-to-source voltage V.sub.DS across the main switch much exceeds V.sub.I. See, for example, the waveform labeled V.sub.DS,Q1 of FIG. 5. In that waveform the V.sub.DS voltage across the main switch much exceeds the voltage V.sub.I. If the main switch is off, the main switch will have a voltage much greater than V.sub.I between its drain and source. The main switch therefore has to be able to tolerate a V.sub.DS breakdown voltage that is much higher than V.sub.I. Having to provide a provide main switch having such a high breakdown withstand voltage rating serves to decrease the performance of the main switch.
(53) Advantageously, in the two-switch forward converter of FIG. 19, the V.sub.DS voltage across each main switch does not much exceed V.sub.I. FIG. 20 is a waveform diagram that illustrates operation of the novel two-switch forward converter circuit of FIG. 19. Voltage V.sub.I in this example of approximately 156 volts. Note that neither of the maximum V.sub.DS voltages across the Q1 and Q2 main transistors as indicated in the two top waveforms of FIG. 20 much exceeds voltage V.sub.I. Accordingly, either less expensive main switches can be used as compared to a single switch topology, or a more efficient AC-to-DC converter circuit can be realized by using transistors for Q1 and Q2 that have lower R.sub.DS(ON) values due to more relaxed V.sub.DS breakdown withstand voltage requirements on the main switch transistors.
(54) The use of inductive current splitting to drive bipolar transistors in self-driven synchronous rectifiers is not limited to use in AC-to-DC converter circuits involving forward converters. The use of inductive current splitting to drive bipolar transistors in self-driven synchronous rectifiers is also applicable to AC-to-DC converter circuits involving push-pull converters, half-abridge converters, full-bridge converters, and phase shift full bridge converters.
(55) FIG. 21 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 200 involving a push-pull converter. FIG. 22 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 300 involving a half-bridge converter. FIG. 23 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 400 involving a full-bridge converter. FIG. 24 (Prior Art) is a circuit diagram of a conventional AC-to-DC converter circuit 500 involving a phase shift full bridge converter. In each case, the secondary side of the converter is the same circuit 600. Circuit 600 involves a center-tapped secondary winding 601, a first diode D1 602, a second diode D2 603, an inductor L1 604, and an output capacitor 605. Whereas in a forward converter the magnetization of the transformer core is one-sided as shown in FIG. 25, in the circuits of FIGS. 21-24 the magnetization of the transformer core is two-sided as shown in FIG. 26. In the case of one-sided magnetization, the core must be reset as described above to prevent residual magnetization from building up from switching cycle to switching cycle and eventually saturating the core. Despite core resetting circuitry, for a given power output of the converter the core is typically made somewhat larger in order to accommodate a certain amount of residual magnetization and margin and to prevent inadvertent saturation of the core. In the case of two-sided magnetization, on the other hand, any residual magnetization in the core at the end of a switching cycle will be automatically removed in the subsequent switching cycle because the subsequent switching cycle magnetizes the core in the opposite direction. For a given power output of the converter, the core can be smaller if two-sided core magnetization is employed as compared to if one-sided core magnetization is employed. Minimizing the size of the core in this way is especially desirable in high power applications where the required core is large and expensive.
(56) In accordance with another novel aspect, the secondary side circuit 600 is replaced with the novel secondary side circuit 700 of FIG. 27. Novel secondary side circuit 700 involves an inductive current splitting circuit that self-drives two bipolar transistor-based synchronous rectifiers. The first bipolar transistor-based synchronous rectifier includes bipolar transistor 701 and diode 702. The second bipolar transistor synchronous rectifier includes bipolar transistor 703 and diode 704. Only one of the synchronous rectifiers conducts current at a time. A common inductive current splitting circuit is shared between the two synchronous rectifiers and supplies the necessary base currents into the bipolar transistors 701 and 703 at the appropriate times. The inductive current splitting circuit involves a first inductor L1 705 and a second inductor L2 706. The overall circuit 700 further includes a center-tapped transformer secondary winding 707 and an output capacitor 708, interconnected as shown in FIG. 27. The emitter of the first bipolar transistor and the cathode of the first diode are coupled together and to a first end of the secondary winding. The emitter of the second bipolar transistor and the cathode of the second diode are coupled together and to a second end of the secondary winding. A first terminal of the output capacitor is coupled to the centertap of the secondary winding. The collectors of the two bipolar transistors and the anodes of the two diodes are all coupled together at a first node M.sub.1. The bases of the two bipolar transistors are coupled together at a second node M.sub.2. A first inductor of the current splitting inductor pair is coupled between a second terminal of the output capacitor and the first node M.sub.1. A second inductor of the current splitting inductor pair is coupled between the second terminal of the output capacitor and the second node M.sub.2.
(57) FIG. 28 is a waveform diagram that shows currents and voltages in the circuit 700 as the circuit operates. Operation is substantially the same regardless of which one of the circuits of FIGS. 21-24 the circuit 700 is a part of. As compared to the conventional second side circuit 600 where current flowing through diodes D1 602 and D2 603 experiences a relatively large voltage drop of about one volt, in the novel circuit 700 the voltage drop across the bipolar transistor synchronous rectifiers is substantially smaller. The V.sub.CE,Q2 and V.sub.CE,Q1 waveforms show the voltage drop across one of the bipolar synchronous rectifiers has an average magnitude of about 0.3 volts when a forward current is flowing through the synchronous rectifier. Due to this smaller voltage drop, the novel secondary side circuit 700 of FIG. 27 exhibits less power loss as compared to the conventional secondary side circuit 600 of FIGS. 21-24.
(58) Although the present invention has been described in connection with certain specific embodiments for instructional purposes, the present invention is not limited thereto. Accordingly, various modifications, adaptations, and combinations of various features of the described embodiments can be practiced without departing from the scope of the invention as set forth in the claims.
