System access and synchronization methods for MIMO OFDM communications systems and physical layer packet and preamble design

Abstract

A method and apparatus are provided for performing acquisition, synchronization and cell selection within an MIMO-OFDM communication system. A coarse synchronization is performed to determine a searching window. A fine synchronization is then performed by measuring correlations between subsets of signal samples, whose first signal sample lies within the searching window, and known values. The correlations are performed in the frequency domain of the received signal. In a multiple-output OFDM system, each antenna of the OFDM transmitter has a unique known value. The known value is transmitted as pairs of consecutive pilot symbols, each pair of pilot symbols being transmitted at the same subset of sub-carrier frequencies within the OFDM frame.

Claims

1. A method for transmission by a transmitter in a communication network, comprising: transmitting, from each antenna of a plurality of antennas of the transmitter, a set of subcarriers in an Orthogonal Frequency Domain Multiplexing (OFDM) symbol of a plurality of OFDM symbols in a slot of a frame, wherein the set of subcarriers from each antenna of the plurality of antennas of the transmitter is non-contiguous, wherein the frame contains data for one or more receivers and is one of a plurality of frames in an OFDM frame as part of a packet data frame structure; wherein the non-contiguous set of subcarriers for each antenna of the plurality of antennas of the transmitter comprises a first subset of subcarriers carrying one or more dedicated pilots, wherein each of the one or more dedicated pilots are modulated by a sequence that is related to a cell, wherein the first subset of subcarriers received from the plurality of antennas of the transmitter is non-overlapping; wherein the transmitting is performed in the slot according to an OFDM symbol mode of a plurality of OFDM symbol modes, wherein the OFDM symbol mode determines a number of OFDM symbols per slot, with no change to a duration of the slot for each of the modes; wherein the frame comprises signaling information indicating whether the frame contains data for the one or more receivers; wherein at least the one or more dedicated pilots on the first subset of subcarriers enable cell selection or re-selection for the one or more receivers.

2. The method of claim 1, wherein a different set of subcarriers is transmitted from each antenna of the plurality of antennas of the transmitter.

3. The method of claim 1, wherein the OFDM symbol is at the beginning of the frame.

4. The method of claim 3, wherein the OFDM symbol at the beginning of the frame is a header OFDM symbol.

5. The method of claim 1, wherein the sequence is at least locally unique to the cell.

6. The method of claim 5, wherein the sequence is a PN sequence.

7. The method of claim 1, wherein the first subset of subcarriers carrying the one or more dedicated pilots are used for channel estimation by the one or more receivers.

8. The method of claim 1, wherein the plurality of OFDM symbol modes correspond to a plurality of different cyclic prefix durations.

9. The method of claim 1, wherein the plurality of OFDM symbol modes correspond to a plurality of fast Fourier transform (FFT) sizes.

10. The method of claim 1, wherein the plurality of OFDM symbol modes are received using a same sampling rate at the receiver.

11. The method of claim 1, wherein the slot also includes data symbols, wherein the data symbols do not comprise dedicated pilots.

12. The method of claim 1, wherein the OFDM symbol mode further determines the number of OFDM symbols per slot with no change to the packet data frame structure above the slot.

13. An apparatus, comprising: a plurality of antennas; and an integrated circuit configured to transmit, from each antenna of the plurality of antennas, a set of subcarriers in an Orthogonal Frequency Domain Multiplexing (OFDM) symbol of a plurality of OFDM symbols in a slot of a frame, wherein the set of subcarriers from each antenna of the plurality of antennas is non-contiguous, wherein the frame contains data for one or more receivers and is one of a plurality of frames in an OFDM frame as part of a packet data frame structure; wherein the non-contiguous set of subcarriers for each antenna of the plurality of antennas comprises a first subset of subcarriers carrying one or more dedicated pilots, wherein each of the one or more dedicated pilots are modulated by a sequence that is related to a cell, wherein the first subset of subcarriers received from the plurality of antennas is non-overlapping; wherein the integrated circuit is further configured to transmit, in the slot, the set of subcarriers according to an OFDM symbol mode of a plurality of OFDM symbol modes, wherein the OFDM symbol mode determines a number of OFDM symbols per slot, with no change to a duration of the slot for each of the modes; wherein the frame comprises signaling information indicating whether the frame contains data for the one or more receivers; wherein at least the one or more dedicated pilots on the first subset of subcarriers enable cell selection or re-selection for the one or more receivers.

14. The apparatus of claim 13, wherein a different set of subcarriers is transmitted from each antenna of the plurality of antennas.

15. The apparatus of claim 13, wherein the OFDM symbol is at the beginning of the frame.

16. The apparatus of claim 15, wherein the OFDM symbol at the beginning of the frame is a header OFDM symbol.

17. The apparatus of claim 13, wherein the sequence is at least locally unique to the cell.

18. The apparatus of claim 13, wherein the first subset of subcarriers carrying the one or more dedicated pilots are used for channel estimation by the one or more receivers.

19. The apparatus of claim 13, wherein the plurality of OFDM symbol modes correspond to a plurality of different cyclic prefix durations or to a plurality of fast Fourier transform (FFT) sizes.

20. The apparatus of claim 13, wherein the plurality of OFDM symbol modes are received using a same sampling rate at the receiver.

21. The apparatus of claim 13, wherein the slot also includes data symbols, wherein the data symbols do not comprise dedicated pilots.

22. The apparatus of claim 13, wherein the OFDM symbol mode further determines the number of OFDM symbols per slot with no change to the packet data frame structure above the slot.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) Preferred embodiments of the invention will now be described in greater detail with reference to the accompanying diagrams, in which:

(2) FIG. 1A is the frame structure of IEEE 802.11 standard in the time domain;

(3) FIG. 1B is the frame structure of FIG. 1A in the frequency domain;

(4) FIG. 2A is a packet data frame structure provided by an embodiment of the invention;

(5) FIG. 2B is a packet frame hierarchy provided by an embodiment of the invention;

(6) FIG. 3 is a proposed header structure provided by an embodiment of the invention;

(7) FIG. 4 is a preamble header structure in the time domain provided by an embodiment of the invention;

(8) FIG. 5 is a preamble header structure in the frequency domain provided by an embodiment of the invention;

(9) FIG. 6 is a conceptual schematic view of a MIMO-OFDM transmitter provided by an embodiment of the invention;

(10) FIG. 7A is a block diagram of a MIMO-OFDM coarse synchronization functionality;

(11) FIG. 7B is a block diagram of a MIMO-OFDM fine synchronization functionality;

(12) FIG. 8 is a plot of a signature sequence correlation output for pilot channel showing several candidate synchronization position;

(13) FIG. 9 is a plot of a BTS identification simulation; and

(14) FIG. 10 is a flowchart of a method for cell selection and re-selection for MIMO-OFDM provided by an embodiment of the invention.

DETAILED DESCRIPTION

(15) Referring now to FIG. 2A, an OFDM packet frame structure provided by an embodiment of the invention is shown. Transmit OFDM symbol streams are organized into such frames. Each frame consists of three major components: preamble 300, scattered pilots 302, and traffic data symbols 304. The insertion of the preamble allows UE (user equipment) to perform the following fundamental operations: fast BTS (base station) access, BTS identification and C/I ratio measurement, framing and timing synchronization, frequency and sampling clock offset estimation and initial channel estimation. The design of a frame preamble with minimized overhead is critical to maximum spectral efficiency and radio capacity.

(16) Referring now to FIG. 2B, a frame hierarchy for MIMO-OFDM is organized according to an embodiment of the invention as follows: at the highest level are OFDM superframes 500 (two shown). The duration of the superframe is determined by the network synchronization period (for example 1-second). The superframe is composed of several 10 ms radio frames 502 also referred to as OFDM frames. There would be 100 10 ms OFDM frames 502 in a 1 s superframe 500.

(17) To support adaptive coding modulation (ACM), a fast signaling channel (TPS channel-transmission parameter signaling) is introduced. Each OFDM frame 502 is subdivided into TPS frames 504, in the illustrated example there are five 2 ms TPS frames for each 10 ms radio frame 502. The frame length used for TPS in some embodiments is the same as the duration of the ACM unit. Each TPS frame also contains signaling information which allows each user to determine whether the current TPS frame contains data for them or not. A TPS frame may contain data for multiple users.

(18) The TPS frame 504 can be divided further into several slots 506, each of which consists of several OFDM symbols. In the illustrated example, each TPS frame 504 is subdivided into 3 slots 506. The duration of the slot 506 depends upon the air interface slot size. The smallest transmission unit is one OFDM symbol 508, 510. The duration of one OFDM symbol is determined by the transmission environment characteristics, for example, the maximum channel delay, the system-sampling clock and the maximum Doppler. In the illustrated example, there are four OFDM symbols 508, 510 per slot 506.

(19) To reduce the overhead caused by the insertion of the guard interval between OFDM symbols, different OFDM symbol modes each with a different symbol duration and a different prefix can be designed, for example, 0.5 k mode and 1 k mode. To simplify the system the sampling frequency is kept unchanged when doing the mode switching. These different modes are described in more detail below.

(20) The frame structure of FIG. 2B gives an example of a frame structure hierarchy compatible to the UMTS air-interface. At the OFDM symbol level, there are two different types of OFDM symbols. These include the preamble OFDM symbols 508 and regular data symbols 510.

(21) Referring now to FIG. 4, which is a time domain representation, each OFDM frame starts with a preamble, which consists of several identical header OFDM symbols 603, 605 preceded by a prefix 607 which is a cyclic extension of the header OFDM symbols. A repetition structure is used to assist synchronization. By performing a correlation between adjacent OFDM symbols until two identical symbols are identified, the start of an OFDM frame can be found. By way of example, there may be 1056 samples used per OFDM symbol. For the preamble, during the prefix 607, the last 64 samples of the header OFDM symbols are transmitted. There is no prefix for the second header OFDM symbol. The header is inserted periodically, and for the example of FIG. 2B, this occurs every 10 ms, i.e. at the beginning of every OFDM frame.

(22) Referring again to FIG. 2B, it is noted that for non-header OFDM symbols, i.e. for the regular OFDM symbols 510, every OFDM symbol preferably also has a prefix. In 1K mode, there are 32 prefix samples, and 1024 actual samples representing the FFT size, for a total of 1056 samples per symbol. In K mode, there is a 16 sample prefix, and then 512 samples per symbol (representing the FFT size) for a total of 528 samples/symbol. Advantageously, using the frame structure of FIG. 2B these different modes can be supported without changing the sampling frequency. When in K mode, there are twice as many OFDM symbols 510 per slot 506. The particular mode chosen at a given instant should be such that the prefix size is greater than the maximum channel delay. In 1K mode, more OFDM symbols are sent with fewer sub-carriers. This is more robust to high Doppler, because the symbol duration is shorter. Also, the spacing between the sub-carriers is larger further enhancing tolerance to Doppler. Thus, there is a unified frame structure which accommodates different FFT sizes, but with the same sampling rate as the receiver. Preferably the same preamble is used even for the different modes.

(23) OFDM is a parallel transmission technology. The whole useful bandwidth is divided into many sub-carriers, and each sub-carrier is modulated independently. According to an embodiment of the invention, to separate different antenna with multiple antennas transmission, during the header not all sub-carriers are used on all transmit antennas. Rather, the sub-carriers are divided between antennas. An example of this will now be described with reference to FIG. 3. The sub-carrier frequencies contained within an OFDM symbol are each represented by circles. In this example it is assumed that there are two transmitting antennas in the MIMO system. FIG. 3 shows OFDM symbols with the various sub-carriers spaced along the frequency axis 400, and with the contents of all the sub-carriers at a given instant representing one symbol in time, as indicated along the time axis 402. In this case, the first two OFDM symbols 408, 410 are used for dedicated pilot channel information while the remaining symbols (only two shown, 412, 414) are used for regular OFDM symbols. The dedicated pilot channel information transmitted on the first two OFDM symbols 408, 410 alternates by sub-carrier between being transmitted by the first antenna and the second antenna. This is indicated for the first sub-carrier 404 which is transmitting dedicated pilot channel information for the first transmitter and sub-carrier 406 which is transmitting dedicated pilot channel information for the second sub-carrier, and this pattern then repeats for the remainder of the sub-carriers. The other OFDM symbols 412, 414 contain information transmitted by both antennas. It is to be understood that other spacing could alternatively be used. Furthermore, if there are more than two transmit antennas, the pilot channel information would then alternate by sub-carrier in some predetermined pattern between all of the transmit antennas.

(24) In another embodiment, a common synchronization channel, and dedicated pilot channel are frequency multiplexed onto the header symbols. A respective set of non-overlapping sub-carriers are assigned for each antenna to transmit respective dedicated pilot channel and common synchronization channel.

(25) In another embodiment a common synchronization channel, dedicated pilot channel and a broadcasting channel are frequency multiplexed onto the header symbols. Under this arrangement, the total useful sub-carriers of the header symbols are separated into three groups. These three groups are mapped onto the common synchronization channel, dedicated pilot channel and the broadcasting channel respectively.

(26) An example of the mapping of the different channels in the MIMO-OFDM system with two-transmitter diversity is shown in FIG. 5. In this example, there are shown four OFDM symbols 712, 714, 716, 718 two of which 712, 714 are header symbols. During the header symbols 712, 714 every second sub-carrier is used for the first antenna with the remaining sub-carriers used for the second antenna. This is easily generalized to higher numbers of antennas. For this example, it is assumed that there are two transmit antennas in the MIMO system. Every sixth sub-carrier starting at the first sub-carrier 700 is for the first transmitter dedicated pilot channel sub-carriers. Every sixth sub-carrier starting at the second sub-carrier 702 is for the second transmitter dedicated pilot channel sub-carrier. Every sixth sub-carrier starting at the third sub-carrier 704 is for the first transmitter common synchronization channel sub-carrier. Every sixth sub-carrier starting at the fourth sub-carrier 706 is for the second transmitter common synchronization channel sub-carrier. Every sixth sub-carrier starting at the fifth sub-carrier is for broadcasting channel sub-carriers for the first antenna, and every sixth sub-carrier starting at the sixth sub-carrier 710 is for broadcasting channel sub-carriers for the second antenna.

(27) The common synchronization channel is a universal channel for initial access. It can also be used for synchronization and preliminary channel estimation. The different transmitters share the common synchronization sub-carriers when transmitter diversity is applied. In which case as indicated above the common synchronization channel is divided between different transmitters. A common complex sequence known by all the terminals is used to modulate the sub-carriers reserved for the common synchronization channel. The same common synchronization sequence is transmitted by all base stations within a system. There may be one or more such synchronization sequences in the event that there are multiple transmit antennas such that each transmit antenna can transmit a unique synchronization sequence. Using the synchronization sequence, mobile stations are able to find initial synchronization positions for further BTS identification by looking for a correlation peak between received synchronization sequence and the known transmitted synchronization sequence.

(28) The dedicated pilot channel is used for BTS/cell identification, and supports C/I measurement for the cell selection, cell switching and handoff. A unique complex sequence, for example a PN code, is assigned to each BTS and used to modulate the dedicated pilot sub-carriers. A different unique sequence is transmitted by each antenna in the multiple transmit antenna case. Unlike the case for the common synchronization channel, different base stations transmit using different pilot sequences. The quasi-orthogonality of the PN codes assigned to different BTSs makes it possible to do access point identification and initial interference measurement. The dedicated pilot channel can also be used to assist the synchronization processing.

(29) To fully utilize the sub-carriers in the header OFDM symbols, as indicated above, some sub-carriers are preferably used as a broadcasting channel. In the example of FIG. 5, two of every six sub-carriers are used for this purpose. The broadcasting channel can carry important system information. STTD (space time transmit diversity) schemes cannot be used for the broadcasting channel (or any of the sub-carriers in the header OFDM symbols) because of it will destroy the repetition structure of the header OFDM symbols which is required by synchronization algorithms. However transmitting the broadcasting information by all transmitters on the same sub-carrier may cause destructive interference between transmitters. To solve that problem the broadcasting channel is partitioned between different transmitters, so in the two transmit antenna case, the sub-carriers (mapped for the broadcasting channel) can be assigned alternatively for the transmit antenna to provide diversity. Power boosting may be applied to further enhance the broadcasting channel.

(30) The broadcasting information from different BTS's can be different. In some embodiments broadcasting information is protected so those users close to the cell boundaries can receive it correctly in the presence of strong interference. A short PN code could be used to spread the broadcasting information. The neighboring BTS is assigned to use different code. The insertion of the broadcasting channel reduces the preamble overhead and increases the spectrum efficiency.

(31) The broadcast channel is used to transmit information unique to the particular base station. A single broadcast message may be sent on the combined broadcast channel carriers for the two antennas. By designing the preamble header symbol to consist of pilot channel, synchronization channel and the broadcasting channel, the preamble header overhead is reduced. The common synchronization channel is designed for fast and accurate initial acquisition. The dedicated pilot channel with a BTS specific mapped signature allows an efficient BTS identification. The combined common synchronization channel and the pilot channel are used together for MIMO channel estimation. The use of the combined common synchronization channel and the dedicated pilot channel also allows for high accuracy synchronization. Frequency domain training symbols are robust to timing error and multipath environments. The preamble design allows the flexibility of the user equipment to implement more efficient algorithms.

(32) It is noted that the specific breakdown of sub-carriers between the dedicated pilot channel in one embodiment, between the dedicated pilot channel and common synchronization channel in another embodiment, and between the dedicated pilot channel, common synchronization channel and broadcast channels in another embodiment, are only specific examples. These can be allocated in any suitable manner.

(33) Referring now to FIG. 6, shown is a conceptual schematic of a MIMO-OFDM transmitter 10. A first sample set of four OFDM symbols 201 is shown transmitted from a first transmit antenna 21 and a second sample set of four OFDM symbols 203 is shown transmitted from a second transmit antenna 23. In general an OFDM transmitter will have N.sub.ant transmit antennae, where N.sub.ant is a design parameter. Within the MIMO-OFDM transmitter 10, data originating from a demultiplexer 23 are sent to one of either a first OFDM component 24 connected to transmit antenna 21 or a second OFDM component 26 connected to transmit antenna 23. The components organize the data onto sub-carriers of OFDM symbols and OFDM frames, each sub-carrier being at a different orthogonal frequency. Each OFDM component 24,26 has a respective header inserter 29 which inserts header OFDM symbols. The sample sets of OFDM symbols 201 and 203 represent the first four OFDM symbols of the transmitted OFDM frame from transmit antennae 21 and 23, respectively, where each row of data symbols or pilot symbols is an OFDM symbol. A first OFDM symbol 13 and a second (identical to the first) OFDM symbol 14 represent the two header OFDM symbols unique to the OFDM frame transmitted by first transmit antenna 21. Similarly, a third OFDM symbol 17 and a fourth (identical to the third) OFDM symbol represent the two header OFDM symbols unique to the OFDM frame transmitted by the second transmit antenna 23. Four OFDM symbols 15, 16, 19, 20 are typically non-identical OFDM symbols made up of a plurality of data symbols, with at least one data symbol indicated generally at 11 on each OFDM sub-carrier. An entire OFDM frame would typically have many more data symbols. Also, the OFDM symbols 201 are transmitted concurrently, and with the same timing, as OFDM symbols 203.

(34) In this example, the two identical header OFDM symbols consist of dedicated pilot channel sub-carriers 12 and common synchronization channel sub-carriers 9. There may also be broadcast channel sub-carriers, not shown. The dedicated pilot channel sub-carriers are used for C/I ratio measurement and BTS identification and fine synchronization as detailed below; they can also be used for initial channel estimation. The common synchronization channel sub-carriers 9 are used for coarse synchronization and fine synchronization, initial access, and initial channel estimation.

(35) In the illustrated example, during the two header OFDM symbols, the first of every four consecutive sub-carriers is used to carry dedicated pilot channel symbols transmitted by transmitting antenna 21. Similarly, the second of every four consecutive sub-carriers is used to carry dedicated pilot channel symbols transmitted by transmitting antenna 23.

(36) The dedicated pilot channel symbols transmitted on the pilot channel sub-carriers 12, 25 are defined by base station/sector specific PN sequence. A set of symbols from a complex pseudo-random PN sequence unique to the base station is mapped onto the dedicated pilot channel sub-carrier locations in the header OFDM symbols.

(37) The third of every four consecutive sub-carriers in the two header symbols is used to carry common synchronization channel symbols transmitted by transmitting antenna 21. Similarly the fourth of every four consecutive sub-carriers is used to common synchronization channel symbols transmitted by transmitting antenna 23.

(38) The common synchronization channel symbols transmitted on the common synchronization sub-carriers 9, 27 are defined by unique complex pseudo-random PN sequence for each transmit antenna 21 and 23. A set of symbols from this complex pseudo-random PN sequence is mapped onto the common synchronization channel sub-carriers in the header OFDM symbols. That is, the common synchronization channel symbols of each frame transmitted through each transmitting antenna use a PN code unique to that transmitting antenna but which is the same for corresponding transmitting antennas of other base stations. In the present example PN.sub.SYNC.sup.(1) is associated with transmit antenna 21 and PN.sub.SYNC.sup.(2) is associated with transmit antenna 23. However, similar antennae in different transmitters throughout the communication network will use the same PN code. For example, the common synchronization channel symbols for a first transmit antenna 21 on all transmitters within the network will use one PN code (PN.sub.SYNC.sup.(1)), and the common synchronization channel symbols for a second transmit antenna 22 on all transmitters within the network will use a different PN code (PN.sub.SYNC.sup.(2)).

(39) Referring to FIG. 7A, a block diagram of MIMO-OFDM receiver functionality is shown which is adapted to perform coarse synchronization based on the two repeated OFDM header symbols transmitted by each transmit antenna as detailed above. The OFDM receiver includes a first receiving antenna 734 and a second receiving antenna 735 (although more generally there will be a plurality of N receiving antennae). The first receiving antenna 734 receives a first received signal at RF receiver 736. The first received signal is a combination of the two signals transmitted by the two transmitting antennae 21 and 23 of FIG. 6, although each of the two signals will have been altered by a respective channel between the respective transmitting antenna and the first receiving antenna 734. The second receiving antenna 735 receives a second received signal at RF receiver 739. The second received signal is a combination of the two signals transmitted by the two transmitting antennae 21 and 23, although each of the two signals will have been altered by a respective channel between the respective transmitting antenna and the second receiving antenna 735. The four channels (between each of the two transmitting antennae and each of the two receiving antennae) may vary with time and with frequency, and will in general be different from each other.

(40) Coarse synchronization is performed for the first receive antenna 734 by a coarse synchronizer 737 on discrete time samples of a received signal to determine an approximate range of a location of the starting position of the first header symbol. A similar process is performed by coarse synchronizer 741 for the second antenna 735. Coarse synchronization is facilitated by the use of repeated header symbols at the OFDM transmitter. The coarse synchronizer 737 performs correlation measurements on time domain signal samples in successive OFDM symbols. The time domain signal sample yielding the highest correlation measurement is the coarse synchronization position n.sub.coarse. The coarse synchronization position n.sub.coarse is then used as the position on which to locate an FFT window within the FFT functions used in fine synchronization.

(41) Initially, the coarse synchronizer 737 starts the time domain coarse synchronization processing. A running buffer (not shown) is used to buffer discrete time samples of the received signal over three successive OFDM symbol period, and then calculates the auto-correlation .sub.t(n) between samples collected during two successive OFDM symbol durations as follows:

(42) ${}_t\left(n\right)=\underset{i=0}{\overset{N_{\mathrm{header}}-1}{.Math.}}x\left(n+i\right).Math.x^{*}\left(n+i+N_{\mathrm{header}}\right)$
where x(n) is the time domain samples of the received signal, N.sub.header is the number of samples taken over one OFDM symbol duration.

(43) In some embodiments, a moving correlator is applied in the real time implementation to save calculation power.

(44) In one embodiment, the values of .sub.t(n) are calculated in sequence, for n=1 (until n=N.sub.header), until a correlation value is above a threshold, after which a maximum search is enabled. The computation of the correlation values continues and the maximum search process will continue until the correlation result is below the threshold again. The sample position corresponding to the maximum correlation value is the coarse synchronization position:
n.sub.coarse=arg max(|.sub.t(n)|n{.sub.t(n)>.sub.threshold}

(45) The threshold is typically calculated from the average auto-correlation values within one frame. Alternatively, another way of finding the maximum is to determine a local maximum for each OFDM symbol over an OFDM frame which might be 60 symbols in length for example. Then, the overall maximum is taken to be the maximum of the local maxima. This process is conducted both coarse synchronizers. In the event fine synchronization is to proceed jointly, the overall coarse synchronization position may be taken as some combination of the two synchronization values, and is preferably taken to be the earlier of two coarse synchronization positions thus determined. Alternatively, each fine synchronizer (detailed below) can work from a respective coarse synchronization position.

(46) Referring to FIG. 7B, a block diagram is shown of an MIMO-OFDM fine synchronization functionality is shown. In one embodiment, the fine synchronization functionality is adapted to perform fine synchronization based on the two-repeated OFDM header symbols transmitted by each transmit antenna as detailed above using the common synchronization channel and/or the dedicated pilot channel. More generally, the fine synchronization functionality can perform fine synchronization for OFDM frames within which some known training sequence has been embedded. Also, an input to the fine synchronization process is a coarse synchronization position. This coarse synchronization position may be determined using the above discussed method, or using any other suitable method. The components which are identical to those of FIG. 7A are similarly numbered and in an actual implementation would be shared if the common synchronizers of FIG. 7A are to be used. The functionality of FIG. 7B is replicated for each of the one or more receive antenna.

(47) A fine synchronization process is performed for each of one or more receive antennae, and then an overall synchronization position is taken based on a combination of the fine synchronization positions. By way of overview, once the coarse synchronizers have determined the coarse synchronization position(s) n.sub.coarse, each fine synchronizer performs an FFT on the signal samples on either side of the coarse synchronization position, to generate frequency domain components over the frequency band of OFDM sub-carriers. Each fine synchronizer searches the frequency domain components in order to locate the precise location of the FFT window. The precise location of the FFT window is required in order to perform OFDM demodulation in the frequency domain. The fine synchronizer locates the precise location of the FFT window by performing correlation measurements between the known PN codes (PN.sub.SYNC.sup.(1) & PN.sub.SYNC.sup.(2)) and the frequency components within a searching window defined with respect to the coarse synchronization position n.sub.coarse. The correlation measurements performed by each fine synchronizer are performed in the frequency domain, and one set of correlation measurements is performed for each known PN code (PN.sub.SYNC.sup.(1) & PN.sub.SYNC.sup.(2)), that is, for each transmitting antenna 21 and 23 (or for how many of the one or more transmit antenna there are).

(48) Each fine synchronizer selects N.sub.symbol signal samples starting at an initial signal sample within the searching window, where N.sub.symbol is the number of signal samples in an OFDM symbol. For each transmitting antenna, each fine synchronizer determines a correlation measurement between the frequency domain signal samples and the PN code corresponding to the transmitting antenna.

(49) More specifically, fine synchronization searching is performed near N.sub.coarse. Supposing that the searching window is 2N+1, the searching range is from (n.sub.coarseN) to (n.sub.coarse+N). Let n.sub.start(i)=n.sub.coarse+Ni represent the sample index within the fine searching window, where i=0, . . . , 2N. The fine synchronization starts from i=0. Then N.sub.symbol samples are taken starting from n.sub.start (0), the prefix is removed and FFT is performed. The received OFDM symbol in frequency domain can be written as:
R(l,i)=FFT(x(n(i),l)),n(i)=[n.sub.start(i)+N.sub.prefix,n.sub.start(i)N.sub.symbol1];l=1, . . . N.sub.FFT
where N.sub.prefix is the number of prefix samples and N.sub.FFT is the FFT size.

(50) From R, the complex data R.sup.(j,k).sub.SYNC carried by the common synchronization channel of different transmitters is extracted, since common synchronization channels are divided between different transmitters in MIMO OFDM system. More generally, the complex the data corresponding to a transmitted training sequence is extracted. The correlation between R.sup.(j,k).sub.SYNC and PN*.sup.(j).sub.SYNC is:

(51) ${}_f^{\left(j,k\right)}\left(i\right)=\underset{m=0}{\overset{N_{\mathrm{SYNC}}-1}{.Math.}}{R}_{\mathrm{SYNC}}^{\left(j,k\right)}\left(m,i\right).Math.{\mathrm{PN}}_{\mathrm{SYNC}}^{*\left(j\right)}\left(m\right),i=0,\mathrm{.Math.},2N$
where j=1, 2, . . . , N.sub.Tx indicates transmitter, k=1, 2, . . . , N.sub.Rx indicates receiver, PN.sup.(j).sub.SYNC is the common SYNC PN code for j.sup.th transmitter and N.sub.SYNC is the size of common PN code.

(52) Then the starting point index n.sub.start is shifted by one (n.sub.start (1)=n.sub.start (0)1), and another N.sub.symbol samples are processed as described above. In order to get the new frequency domain data R.sup.(j,k).sub.SYNC(m,i), we need to perform FFT again. An iterative method can be used for this purpose to reduce the computational complexity:

(53) $R\left(l,i\right)=R\left(l,i-1\right).Math.e^{\frac{i2\left(k-1\right)}{\mathrm{NFFT}}}+x(n_{\mathrm{start}}\left(i\right)+N_{\mathrm{prefix})}-x\left(n_{\mathrm{start}}\left(i-1\right)+N_{\mathrm{symbol}}-1\right)$
where NFFT is the FFT size. Extracting R.sup.(j,k).sub.SYNC(m,i), the new correlation is calculated. The above procedure is continued until n.sub.start moves out of the fine searching window.

(54) $n_{\mathrm{fine}}=\arg \max \left(\underset{j=1}{\overset{N_{\mathrm{Tx}}}{.Math.}}\underset{l=1}{\overset{N_{\mathrm{Rx}}}{.Math.}}\left|{}_f^{\left(j,k\right)}\left(i\right)\right|\right)$

(55) For each receive antenna, a respective fine synchronization position can be found by finding n.sub.start(i) corresponding to the maximum of the products of the correlation results from different antennas over i=0, . . . , 2N. In mathematical terms, for the kth receive antenna, a respective fine synchronization position can be selected according to:

(56) $n_{\mathrm{fine}}\left(k\right)=\arg \max \left(\underset{j=1}{\overset{N_{\mathrm{Tx}}}{.Math.}}\left|{}_f^{\left(j,k\right)}\left(i\right)\right|\right)$

(57) To reduce the possibility of false alarm, a criterion may be set. For example, the fine synchronization may be considered to be achieved if the following condition is satisfied,

(58) $\max \left(\underset{j=1}{\overset{N_{\mathrm{Tx}}}{.Math.}}\left|{}_f^{\left(j,j\right)}\left(i\right)\right|\right)>N_{\mathrm{threshold}}.Math.\frac{1}{2N+1}.Math.\underset{i=0}{\overset{2N}{.Math.}}\underset{j=1}{\overset{N_{\mathrm{Tx}}}{.Math.}}\left|{}_f^{\left(j,j\right)}\left(i\right)\right|$
where N.sub.threshold is a factor determined by the pre-set fine searching window size. Preferably, an overall fine synchronization position is then taken to be the earliest of the fine synchronization positions determined for the different receive antennas.

(59) The fine synchronization process for one receive antenna is illustrated diagrammatically in FIG. 7B. At the output of the first receiver 736, blocks D0 738 through D2N 742 represent alignment of the FFT blocks 744, . . . , 748 for the various candidate fine synchronization positions (2N+1 in all). The FFT blocks 744, . . . , 748 compute an FFT on each respective set of samples. Each FFT output is fed to a correlator block for each transmit antenna. If there are two transmit antennae, then there would be two such correlator blocks per FFT output. For example FFT 744 has an output fed to a first correlator block 745 for the first transmit antenna, and fed to a second correlator block 755 for a second transmit antenna. It is noted that if the spacing of the sub-carriers used to transmit the training sequence (the common synchronization sequence or pilot channel sequence in the above examples), a full FFT does not need to be completed in order to recover the training sequence components. The correlator block 745 for the first antenna multiplies with multiplier 747 the recovered training sequence symbol locations of the FFT output by the known training sequence for the first transmit antenna and these multiplications are added in summer 751. This same computation done in correlator 755 for the known training sequence of the second transmit antenna and the training sequence locations for the second transmit antenna. This is done at the first receiver for all of the different possible shifts for each transmit antenna. The correlation results across different transmit antennas for each possible shift are multiplied together in multipliers 753. The shift which results in the maximum of these multiplications is selected to be the fine synchronization position for the particular receiver. The same process is followed for any other receive antennas, and the overall fine synchronization position is preferably taken as the earliest of the fine synchronization positions thus computed.

(60) The timing synchronization can be tracked every frame in case that the synchronization position drifts or losses. For example, in systems employing the previously described preamble, each time a preamble arrives at the receiver the 2-step process of synchronization is repeated, using the same method for coarse synchronization and fine synchronization. In this case, a smaller searching window N may be used based on the assumption that the drift of the synchronization position should be around the vicinity of the current location. After acquisition, the dedicated pilot channel code assigned to modulate dedicated pilot channels for different BTS can be used in the correlator, or the common synchronization sequence can be used, or some other training sequence.

(61) An embodiment of the invention has been described with respect to an MIMO-OFDM transmitter having more than one transmitting antenna. The method of performing synchronization at the OFDM receiver may also be applied to a signal received from an OFDM transmitter having only one transmitting antenna, as long as a known training sequence is inserted in the frame by the OFDM transmitter.

(62) Lastly, in the embodiment of the invention described thus far there has only been one transmitter having multiple antennae and one receiver having multiple antennae. In what follows, the concepts of the invention will be broadened to encompass the multi-cellular environment having many MIMO-OFDM transmitters and many MIMO-OFDM receivers.

(63) Access in a Multi-Cellular Environment

(64) System access in a multi-cellular environment introduces the new problem of cell selection, as there will be many transmitters transmitting the same common pilot symbols. In another embodiment of the invention, the previously introduced transmit header is used by receivers to perform systems access and cell selection.

(65) During initial acquisition, the UE starts by performing coarse synchronization. This may be done using the previously described methods, or some other method. After one frame duration, the coarse synchronization position is determined. Fine synchronization search algorithm is performed afterwards based on the common synchronization channel. Because the data carried by the common synchronization channel are the same for all BTS, several fingers (peaks) can be observed in a multi-cell environment and multi-path fading propagation channels. These fingers usually correspond to different BTS and/or different paths. Referring to FIG. 8, shown is an example of fine synchronization (to the common synchronization channel) raw output computed in a multi-cellular environment as a function of sample index. In the present example there are five significant fingers 400, 402, 404, 406, and 408. The M strongest fingers are chosen and the corresponding positions are located, where M is a system design parameter. These positions are used as candidates for final synchronization and also as the positions upon which BTS identifications are made.

(66) The results of FIG. 8 do not allow BTS identification because BTS transmit the same common synchronization sequences. At each candidate synchronization position, the correlations of the received dedicated pilot channel sub-carriers and all possible complex sequences (dedicated pilot PN sequences) assigned to different BTS are calculated to scan for the presence of all the possible adjacent BTSs. In the multiple transmit antenna case, preferably this correlation is done on the basis of the combined dedicated pilot PN sequences of the multiple antennas over all of the dedicated pilot sub-carriers to generate a single correlation result for each index. FIG. 9 shows an example of the relation between the BTS scanning results and the checking points (candidate synchronization positions). The BTS identification is realized by detecting the PN code corresponding to the maximum correlation value at each candidate synchronization position. C/I can be computed based on all correlation results at each checking position. At the initial acquisition stage, the cell selection is determined by selecting the BTS with the largest C/I ratio. In the present example two BTS are identified, a first BTS BTS1 and a second BTS BTS2. With multiple-antenna receiver diversity, the final decision of the cell selection should be based on the comparison of the highest C/I obtained by different receiver antennae at a receiver.

(67) To obtain the final synchronization position, fine synchronization is performed again, but by using the dedicated pilot channel and the dedicated complex sequence found through the BTS identification. A smaller searching window around the fine synchronization position is used. The final synchronization results from different receivers are compared. The position corresponding to the earliest sample in time is used as the final synchronization position. This step is to reduce the possibility that a weak path (multi-path) is selected because of the short-term fading. To reduce the false alarm probability, a threshold is set. This threshold can be the ratio of the finger strength with respect to the final synchronization position and the average of the correlation within the search window.

(68) In the normal data processing stage, the fine synchronization and the BTS identification steps are repeated every frame when a new preamble is received, but a small set of the candidate PN codes is applied in the BTS scan. After BTS identification, a BTS candidates list can be generated through searching strong interferences. This list is updated periodically, for example every 10 ms, and provides information for BTS switch and even soft handoff. Certain criteria can be set in order to trigger the BTS switch and soft handoff. To average the impact from the fading, the decision for BTS switching and the soft handoff may be based on observation during a certain period. The criteria can be the comparison of the maximum correlation values representing C and the strongest I. It should be noted that after the cell switch and the soft handoff, the synchronization may also be adjusted by the final step in the initial access. The overall cell selection and re-selection method is shown in FIG. 10.

(69) In the first step 600, coarse synchronization is performed for example based on the preamble header in the time domain. This involves finding a coarse boundary between each frame by looking for two identical symbols. Correlating samples over adjacent symbol durations until a peak is found does this. Step 600 relies on a preamble to a frame beginning with two adjacent identical symbols.

(70) Next during step 602, at the coarse synchronization peaks, an FFT is computed, and a switch to the processing of the common synchronization channel in the frequency domain is made. A search window is centered on sync position+/a certain number of samples. The M strongest correlation peaks are selected, as per 604. At this time, it is not known which BTS each peak is associated with. BTS identification has not yet been determined.

(71) Then at step 606, for each correlation peak, the FFT is again computed and the correlations associated with the fine synchronization procedure are computed using the dedicated pilot channelsthese containing a base station specific complex sequences. This is immediately followed by step 608 where the correlation with the BTS identification complex sequences is made in order to allow an identification of the associated base stations. At step 610, a C/I ratio is computed for each BTS thus identified. BTS selection and BTS switching is performed on the basis of these C/I ratios in step 612. AS indicated above, BTS switching is performed on the basis of these C/I ratios averaged over some time interval.

(72) Finally, for access, the FFT is computed and fine synchronization is applied to the dedicated pilot channel of the BTS with the largest C/I ratio as per step 614.

(73) BTS initial synchronization performed on the common synchronization channel. A BTS specific sequence is embedded in the frequency domain and BTS identification processing is performed in the frequency domain allowing the elimination of MIMO-OFDM inter-channel interference. BTS power estimation is performed based on the pilot channel for each MIMO-OFDM BTS. BTS selection is performed based on C/I ratio measurements.

(74) The result is improvement of the synchronization and identification of the serving BTS in a severe multi-path channel and high interference environment by joint BTS synchronization and cell selection. Channel estimation may be performed on a combined common synchronization channel and dedicated pilot channel. Criteria are provided for cell switching and soft handoff by C/I estimation.

(75) In the above example, the access has been performed based on the synchronization channel and pilot channel embedded in the previously discussed preamble. More generally, the access can be performed with such channels embedded in any suitable manner within an OFDM symbol stream.

(76) What has been described is merely illustrative of the application of the principles of the invention. Other arrangements and methods can be implemented by those skilled in the art without departing from the spirit and scope of the present invention.