Abstract
This invention eliminates the need for capacitor coupling or transformer coupling, and the associated undesirable parasitic capacitance and inductance associated with these coupling techniques when designing high frequency (60 GHz) circuits. At this frequency, the distance between two adjacent stages needs to be minimized. A resonant circuit in series with the power or ground leads is used to isolate a biasing signal from a high frequency signal. The introduction of this resonant circuit allows a first stage to be directly coupled to a next stage using a metallic trace. The direct coupling technique passes both the high frequency signal and the biasing voltage to the next stage. The direct coupling approach overcomes the large die area usage when compared to either the AC coupling or transformer coupling approach since neither capacitors nor transformers are required to transfer the high frequency signals between stages.
Claims
1. An apparatus comprising: a first-stage transistor circuit configured to generate a high-frequency signal that is passed to a second-stage transistor circuit, the first-stage transistor circuit comprising: a conducting trace directly connecting the first-stage transistor circuit to the second-stage transistor circuit; a voltage reference circuit comprising a control loop configured to monitor a first biasing voltage and generate a second biasing voltage, wherein the first biasing voltage is a first input to an operational amplifier of the control loop and an output of the operational amplifier is coupled to a gate of a first transistor, wherein the second biasing voltage is formed across the first transistor and fed back as a second input into the operational amplifier, and wherein the operational amplifier and the first transistor form the control loop; a reference current source electrically coupled to a reference transistor connected in saturation, the reference current source programmable to control the voltage reference circuit configured to generate the second biasing voltage; and a resonant circuit configured to pass the second biasing voltage to the second-stage transistor circuit by way of the conducting trace, wherein the voltage reference circuit is adjustable to provide adjustment to the first biasing voltage to control the second-stage circuit's drive strength.
2. The apparatus recited in claim 1, wherein the resonant circuit is electrically coupled in series with at least one of a power lead and a ground lead.
3. The apparatus recited in claim 1, wherein the resonant circuit is configured to block the high-frequency signal from affecting power supplied to the first-stage transistor circuit.
4. The apparatus recited in claim 1, wherein the resonant circuit comprises a resonant parallel LC load.
5. The apparatus recited in claim 1, wherein the resonant circuit is configured to resonate at a predetermined frequency to provide for selection of a frequency band.
6. The apparatus recited in claim 1, wherein the reference transistor generates the first biasing voltage, the control loop is configured to equalize the second biasing voltage to the first biasing voltage, and the resonant circuit electrically couples the second biasing voltage to a gate of a second transistor in the second-stage transistor circuit.
7. The apparatus recited in claim 6, wherein the first transistor comprises a P-channel Metal Oxide Semiconductor (MOS) device or an N-channel MOS device.
8. The apparatus recited in claim 6, wherein the control loop is configured to employ at least one of a negative feedback and a positive feedback.
9. The apparatus recited in claim 6, further comprising a second resonant circuit that electrically couples a power supply to an output of the second transistor.
10. The apparatus recited in claim 9, wherein the resonant circuit and the second resonant circuit are configured to resonate at a frequency to provide for selection of a frequency band.
11. The apparatus recited in claim 1, wherein at least one of power control, current flow, and biasing of the second-stage transistor circuit are configurable by adjusting the second biasing voltage.
12. The apparatus recited in claim 1, wherein the control loop is configured to employ positive feedback through an N-channel transistor connected to the second input of the operational amplifier.
13. The apparatus recited in claim 1, wherein a current through the second transistor in the second-stage transistor circuit is scaled by an amount M=(W.sub.N3/W.sub.N1) relative to a current through the reference transistor, and wherein W.sub.N3 is a gate width of the second transistor and W.sub.N1 is a gate width of the reference transistor.
14. The apparatus recited in claim 13, wherein the reference transistor comprises a variable gate width that is adjustable.
15. The apparatus recited in claim 13, wherein the reference transistor comprises a plurality of devices coupled together with a plurality of switches, wherein the gate width of the reference transistor is adjustable by opening or closing at least one of the plurality of switches.
16. The apparatus recited in claim 1, wherein the conducting trace has a trace length determined by a minimum separation distance of adjacent devices in a layout.
17. The apparatus recited in claim 1, wherein the second-stage transistor circuit comprises an output connected to an antenna.
18. The apparatus recited in claim 1, configured to generate balanced output voltages.
Description
BRIEF DESCRIPTION OF THE DRAWINGS
(1) Please note that the drawings shown in this specification may not be drawn to scale and the relative dimensions of various elements in the diagrams are depicted schematically and not to scale.
(2) FIG. 1a depicts one embodiment of the direct biasing technique using a first operational amplifier stage in accordance with the present invention.
(3) FIG. 1b illustrates a second embodiment of the direct biasing technique using a second operational amplifier stage in accordance with the present invention.
(4) FIG. 1c depicts the output of the stage coupled to a load in accordance with the present invention.
(5) FIG. 1d shows a portion of an equivalent circuit of the resonant circuit in accordance with the present invention.
(6) FIG. 1e presents a portion of a second equivalent circuit of the resonant circuit in accordance with the present invention.
(7) FIG. 1f illustrates an equivalent circuit of the resonant circuit in accordance with the present invention.
(8) FIG. 1g shows a block diagram of FIG. 1f in accordance with the present invention.
(9) FIG. 1h presents the internal circuit of a first operational amplifier stage with compensation.
(10) FIG. 1i illustrates the internal circuit of a second operational amplifier stage without compensation.
(11) FIG. 2a illustrates the one embodiment of the direct biasing technique with the block diagram of FIG. 1e in accordance with the present invention.
(12) FIG. 2b shows a transistor size adjustment in accordance with the present invention.
(13) FIG. 3a depicts the one embodiment of the direct biasing technique with the additional block diagrams in accordance with the present invention.
(14) FIG. 3b depicts a third embodiment of the direct biasing technique for a balanced output driver in accordance with the present invention.
DETAILED DESCRIPTION OF THE INVENTION
(15) This invention has been incorporated into a Power Stage (PA) design for a 60 GHz wireless system. The inventive apparatus is applicable to any high frequency system, for example, where the parasitic inductance of a metallic trace is of the order 1 pH per micrometer. At 60 GHz, the typical inductance within or between stages is on the order of about 50 to 120 pH. The actual dimensions of the capacitor depends on several issues; the type of capacitor, the overall positive reactance in a given stage that requires compensation; and, the physical layout of the capacitor, for example. At these frequencies, as a signal is coupled from the first stage to the next stage, the parasitic capacitance and/or the parasitic inductance of the coupling circuits is critical and needs to be minimized. This invention eliminates the need for capacitor coupling or transformer coupling, and along with them, the associated undesirable parasitic capacitance and inductance. In this invention, the signal from the first stage is directly coupled to the next stage.
(16) FIG. 1a illustrates one embodiment of the inventive circuit 1-1 and will first be described under DC conditions. Assume that the inductor L.sub.1 has a very low resistance causing the voltage at node 1-5 to nearly equal the voltage at node 1-6. Devices N.sub.1 and N.sub.3 are scaled versions of each other and are well matched. In order to simply the description of the invention, assume that the gate length and oxide thickness of the devices will remain constant while the gate width, W, is adjusted to scale the sizes of matched devices. A second device can be scaled to a first device by a value. For example, if the second device has a width of 100 um and the first device has a width of 10 um, then the second device is scaled to the first device by a value of 10. The device nomenclature for the N.sub.2 and P.sub.1 devices imply an N-channel and P-channel MOS (Metal Oxide Semiconductor) device, respectively. The gate width of device N.sub.3 is scaled M times larger (M times the width of N.sub.1) than the device N.sub.1. A bias current of I.sub.1 flows through the diode connected N.sub.1 device (gate connected to its drain) which provides the gate voltage at node 1-2. An operational amplifier stage 1-3 monitors this voltage on its negative input. The output 1-4 of the operational amplifier stage is applied to the gate of device P.sub.1. The drain 1-5 of P.sub.1 is connected to the positive input of operational amplifier stage 1-3. The operational amplifier stage 1-3 and the device P.sub.1 form a control loop causing the voltage at node 1-5 to be forced to the voltage at node 1-2. Furthermore, the DC biasing voltage at node 1-2 (gate voltage of N.sub.1) can range from 700 to 900 mV. The drain current of N.sub.2 (I.sub.2) is set to about 10 mA and since the DC resistance of the inductor L.sub.1 is about 2, the total DC voltage drop across the inductor L.sub.1 is about 20 mV. This DC voltage drop across the inductor L.sub.1 between nodes 1-5 and 1-6 is about 2% of the desired DC biasing voltage. Thus, the voltage at node 1-5 is essentially equal to voltage at node 1-6, the gate voltage of N.sub.3 becomes essentially equal to the voltage on node 1-2 which is also the gate voltage of N.sub.1. Since devices N.sub.1 and N.sub.3 are well matched devices and the gate voltages of N.sub.1 and N.sub.3 are equal, the current I.sub.3 through N.sub.3 is then M times that of current through device N.sub.1, or I.sub.3=M*I.sub.1, where M=(W.sub.N3/W.sub.N1), W.sub.N3 is the width of device N.sub.3 and W.sub.N1 is the width of device N.sub.1. The current I.sub.3 is supplied by VDD 1-8 and passed through the inductor L.sub.2 to the drain of N.sub.3, node 1-7. By this scheme, the current I.sub.3 through the output device N.sub.3 is controlled by adjusting the device size of N.sub.1, programming the I.sub.1 current source or performing both adjustments simultaneously. The current through the N.sub.2 device is determined by its gate bias voltage of the input signal V.sub.in. The output signal V.sub.out is available on node 1-7.
(17) A very important advantage of this scheme is that the first stage (N.sub.2) is directly coupled to the next stage (N.sub.3) by the metallic trace between node 1-6 and the gate of N.sub.3. The trace length between the node 1-6 and the gate of device N.sub.3 is now determined by the design rules in a given technology regarding the minimum placement of adjacent devices in a layout. Thus, the parasitics of any resistances, inductances or capacitances on this node coupling these two stages has been reduced to the minimum possible for the given technology, thereby reducing any inter-stage coupling losses. Both the inductance and the capacitance are composed of a parasitic and non-parasitic component.
(18) Recall that both AC coupling and transformer coupling introduce either a series capacitor or transformer in the trace between node 1-6 and the gate of N.sub.3 or their equivalent. Both of these components consume large die area. For example, a typical transformer has side dimensions in the range of 80 by 80 um while the coupling capacitor would use about 4 times less area with side dimensions in the range of 40 by 40 um. Thus, the use of a directly coupled trace between node 1-6 and the gate of N.sub.3 has advantages when compared to connecting node 1-6 to the gate of N.sub.3 using either a transformer or a capacitor. Some of these advantages include: 1) a simple trace (metallic conductor) is used to transfer the signal from the drain of N.sub.2 to the gate of N.sub.3; 2) the same metallic conductor carries the DC biasing voltage to accurately control the behavior of N.sub.3; 3) area consumption of high frequency coupling trace decreases to the minimum; 4) die size reduces; 5) cost of die decreases; 6) the parasitic inductance and parasitic capacitance of the short metallic conductor is reduced in complexity; 7) less power is dissipated (less parasitic capacitance/inductance is driven); 8) computation time to model metallic conductor decreases; and 9) simulation time of final layout decreases.
(19) FIG. 1b depicts a second embodiment of the inventive circuit. The circuit is equivalent to the circuit in FIG. 1a except for the adjustable current source I.sub.4, the second operational amplifier stage 1-17 and the device N.sub.8. The adjustable current source I.sub.4 is coupled to the N-channel N.sub.1 with the gate connected to the drain 1-2 and applied to a positive input of the second operational amplifier stage 1-17. The control loop is formed by the second operational amplifier stage 1-17, the second operational amplifier stage output 1-18, the N-channel N.sub.8 and the feedback path 1-19 to the negative terminal of the second operational amplifier stage. The adjustable current source I.sub.4 is used to adjust the gain of the N-channel device N.sub.3.
(20) In FIG. 1a, note that the first operational amplifier stage 1-3 uses negative feedback to form the control loop to control the stability of the first operational amplifier stage 1-3. The control loop is formed by the first operational amplifier stage 1-3, the first operational amplifier stage output 1-4, the P-channel P.sub.1 and the feedback path to the positive terminal of the first operational amplifier stage. FIG. 1h illustrates a compensation network composed of R.sub.2 and C.sub.1 required in the first operational amplifier stage 1-3 to maintain stability. In FIG. 1a, the gain at the output of the amplifier V.sub.out is very difficult to control at millimeter-wave frequencies. In the control loop, the P-channel device P.sub.1 gives an additional phase shift, which must be compensated by adding a zero inside the operational amplifier stage 1-3. The compensation network limits the response time of the operational amplifier stage 1-3. A large compensation capacitor (4 pF) and a 4K resistor are required to overcome any stability concerns causing large time constants. The operational amplifier stage 1-3 in FIG. 1a and the device P.sub.1 form a control loop causing the voltage at node 1-5 to be forced to the voltage at node 1-2 by negative feedback. Due to the large compensation capacitor, the response time of the operational amplifier stage is limited by the RC delay and can as long as 500 nsec.
(21) In FIG. 1b, however, the second operational amplifier stage 1-17 uses positive feedback to form the control loop since the P-channel device has been replaced by the N-channel device N.sub.8. Use of the N-channel device does not required compensation, thus the response time is reduced and the gain control can be accomplished very quickly. Power can be saved if the gain control is fast since the final stage of the transmitter N.sub.3 adjusts the gain on the order of 100 nsec. The second operational amplifier stage 1-17 is depicted in FIG. 1i. A compensation network is not required in the second operational amplifier stage 1-17. In FIG. 1b, the operational amplifier stage 1-17 and the device N.sub.8 form a control loop causing the voltage at node 1-5 to equal the voltage at node 1-2 by positive feedback.
(22) The N-channel N.sub.8 in FIG. 1b eliminates a 180 phase shift in the feedback path 1-19 to the negative input of the operational amplifier stage since the node 1-5 now follows the output 1-18 of the operation stage 1-17 so there is no phase inversion. The stability issue of the control loop in FIG. 1b is eased to the point where the compensation capacitor is not required. Since this operational amplifier stage does not require feedback, the response time can be reduced to 100 nsec.
(23) The internal circuit for the first operational amplifier stage 1-3 is illustrated in FIG. 1h. A current I.sub.5 flows through the N-channel device N.sub.9 providing a bias for current mirror devices N.sub.10 and N.sub.11. The current mirror devices N.sub.10 and N.sub.11 provide a current sink for a differential first stage composed of devices; P.sub.2, P.sub.3, N.sub.13 and N.sub.12 and the output driver device P.sub.4, respectively. The negative input V.sub.n is applied to the gate of N.sub.13 while the positive input V.sub.p is applied to the gate of N.sub.12. The intermediate output 1-20 is then applied to the gate of device P.sub.4. The compensation network couples the output of the first operational amplifier stage V.sub.opamp to node 1-20.
(24) The internal circuit for the second operational amplifier stage 1-17 is illustrated in FIG. 1i. A current I.sub.6 flows through the N-channel device N.sub.19 and current mirror device N.sub.18 provides a current sink for the differential first stage composed of devices; P.sub.6, P.sub.7, N.sub.16 and N.sub.17. The negative input V.sub.n is applied to the gate of N.sub.16 while the positive input V.sub.p is applied to the gate of N.sub.17. The intermediate outputs 1-21 and 1-22 are then applied to the gate of device P.sub.5 and P.sub.8, respectively. Both devices P.sub.6 and P.sub.7 have the drain coupled to their gate. The device N.sub.14 generates the bias current control 1-23 for device N.sub.15. The intermediate output 1-22 and the bias current control 1-23 are applied to the output devices P.sub.8 and N.sub.15, respectively, to generate the output of the second operational amplifier stage V.sub.opamp. This circuit does use feedback internally; thus, the second operational amplifier stage 1-17 should respond quickly to gain changes in the inventive circuit of FIG. 1b.
(25) In FIG. 1c, an example 1-11 of the output node V.sub.out of the next stage (final stage) is shown connected to the load resistor R.sub.ant which can represent the impedance of at least one antenna. The high frequency signal is generated at node 1-7 and is routed on a metallic trace to a point on the die where the signal exits the die, V.sub.out. In addition, the interconnect (dotted line) to the load R.sub.ant can comprise metallic trances, a bonding pad, bonding wire, a solder bump, package traces, interconnects, wires, transmissions lines, vias, etc. Each of these components introduce parasitic resistance, inductance and capacitance into the trace. The antenna, in some cases, may be fabricated on the same die thereby allowing greater control of the parasitics of the metallic trace. The inductor L.sub.2, as will be discussed later, is a load that is set to resonate at 60 GHz.
(26) Very simplified models are presented in FIG. 1d-f. These models, although basic, help present some of the features of the inventive idea. FIG. 1d shows the equivalent impedance of the inductor L.sub.1 at DC. Since Z.sub.L1=jL.sub.dc=0, (where =2f and f is the frequency set equal to zero) the impedance of the inductor L.sub.dc is zero while the impedance of the resistor R.sub.dc would utilize the full cross-sectional area of the metallic trace to carry current (infinite skin depth), so that resistance is minimized at this point. FIG. 1e shows the equivalent impedance the inductor L.sub.1 at some frequency f. When f>0, Z.sub.L1=jL.sub.ac and has a value greater than 0, but because of the skin effect, the impedance R.sub.ac increases as frequency increases. The skin depth is proportional to f.sup.1/2, and as f increases, the current is forced to flow closer to the outer cross-sectional area of the metallic trace presenting more resistance to the flow of the AC current (for instance, the skin depth in copper, Cu, is 0.27 um at f=60 GHz). Thus, when the DC signal and the AC signal are transferred though the metallic trace, the AC signal would experience the skin effect resistance and large reactance while the DC signal will experience the full cross-sectional area of the metallic trace and no reactance.
(27) The capacitance connected to each end of the inductor L.sub.1 within the dotted areas 1-9 and 1-10 of FIG. 1a has been added to the crude model as depicted in FIG. 1f. In FIG. 1f, the capacitance C.sub.p corresponding to the dotted region 1-10 comprises the capacitance of the input of the operation stage 1-3, the drain and miller capacitance of P.sub.1, the near end capacitance of the inductor L.sub.1, and any metallic trace capacitance. The capacitance C.sub.sig corresponding to the dotted region 1-9 comprises the capacitance of the drain and miller capacitance of device N.sub.2, the gate capacitance and overlap capacitance of device N.sub.3, the near end capacitance of the inductor L.sub.1, and any metallic trace capacitance between node 1-6 and the gate of N.sub.3. The nodes 1-6 and 1-7 transfer the signal provided at V.sub.in to the load R.sub.ant. An intermediate signal is generated at 1-6 (the drain of N.sub.2) by V.sub.in. The intermediate signal is directly coupled from 1-6 to the gate of N.sub.3. The output signal V.sub.out is generated at 1-7 (the drain of N.sub.3) and is used to drive the load.
(28) The circuit 1-14 in FIG. 1f models the load of N.sub.2 and is a parallel LC circuit with resistive loss R. The capacitance across the inductor-resistor (L-R) circuit comprises the series connection of C.sub.p and C.sub.sig, or C.sub.eff=(C.sub.p*C.sub.sig)/(C.sub.p+C.sub.sig). The capacitance C.sub.sig can be made dominant by significantly increasing the value of C.sub.p by the introduction of additional capacitance at node 1-5. The larger value capacitance also helps to stabilize the voltage at node 1-5 which is applied to the input of the operational amplifier stage 1-3. The components of the parallel LC circuit are designed to resonate at the frequency of interest (for example, 60 GHz). The resistor resistive loss R broadens the response.
(29) The inductor L.sub.2 would also have a similar model as that given in FIG. 1f. In the case of L.sub.2, C.sub.p would be located at node 1-8. Since this is a lead for the power supply, the capacitance of the power supply traces and would be much larger than the capacitance C.sub.sig at node 1-7 (V.sub.out). Similar techniques can be used for the complimentary circuit where all N-channels are replaced by P-channels and all P-channels are replaced by N-channels, power supplies VDD and VSS are flipped, and the circuit now presents the complement form.
(30) One design illustrating the embodiment would adjust the inductor L.sub.1 and L.sub.2 such that the capacitance C.sub.sig on each node 1-6 or 1-7 resonates at the desired frequency of interest. Once the LC tank circuits resonate, their impedance is at maximum thereby isolating the high frequency signal at nodes 1-6 and 1-7 from the nodes on the other side of the two inductors 1-5 and 1-8. Furthermore, the impedance of the parallel LC tank circuit would also be the load for the devices N.sub.2 and N.sub.3.
(31) Another design illustrating the embodiment would adjust the value of the capacitance C.sub.sig on each node 1-6 or 1-7 while minimizing changes to the inductors L.sub.1 or L.sub.2, respectively, such that both LC circuits resonate at the desired frequency of interest. Once both LC tank circuits resonate, their impedance is at maximum thereby isolating the high frequency signal at nodes 1-6 and 1-7 from the nodes 1-5 and 1-8 on the other side of the two inductors L.sub.1 and L.sub.2, respectively. Furthermore, the impedance of the parallel LC tank circuit would also be the load for the devices N.sub.2 and N.sub.3. A block diagram 1-15 of the LC circuit in FIG. 1f is illustrated in FIG. 1g.
(32) In addition, another embodiment is to alter both the capacitances C.sub.sig and the inductor L1 and L2 simultaneously. In a resonating LC tank circuit, as the capacitance of C decreases, the inductance of L must increase to maintain the circuit in electrical resonance according to: f=(2(LC).sup.1/2).sup.1.
(33) Finally, another design illustrating the embodiment is to minimize both the inductance and the capacitance in both stages, thereby reducing the overall parasitic capacitance and resistance which then leads to a reduction in power dissipation. Then, each LC circuit is set to resonate at the desired frequency by increasing the inductance value of L.sub.1 and L.sub.2. This is done by increasing the physical length of the inductors L.sub.1 and L.sub.2 by linearly increasing their length or by using a circular or spiral pattern layout.
(34) FIG. 2a uses the block diagram 1-15 to simplify the circuit illustrated in FIG. 1a. The current I.sub.3 can be adjusted by varying the size of the device N.sub.1 where I.sub.3=(W.sub.N3/W.sub.N1)*I.sub.1. Another method is by altering the value of the current I.sub.1 in the programmable current source. Altering the current I.sub.1 is straight forward and will not be covered since it is known in the art.
(35) The size of the device N.sub.1 in FIG. 2a can be varied to adjust the current I.sub.3, as illustrated in FIG. 2b. In FIG. 2b the N-channel transistors identified by their channel widths. The summation of the widths of the devices W.sub.a, W.sub.b and W.sub.c in FIG. 2b equals W.sub.N1, since the width of the initial device I.sub.3 is W.sub.N1. One embodiment of a circuit to adjust the size of N.sub.1 is illustrated in FIG. 2b. Thus, if all devices are enabled (all switches connected to node 1-2) by their corresponding switches S.sub.1, S.sub.2 and S.sub.3, the total width would be W.sub.n1. As the switches S.sub.1, S.sub.2 or S.sub.3 become disabled (switched to ground), the summation of the widths of the composite devices N.sub.1a, N.sub.1b and N.sub.1c, respectively, decreases. Thus, the current I.sub.3=(W.sub.N3/W.sub.b)*I.sub.1 in the case illustrated in FIG. 2b.
(36) FIG. 3a depicts another simplification 3-1 to the circuit illustrated in FIG. 2a. The Voltage Ref Block 3-2 has been added to represent the operational amplifier stage 1-3, devices N.sub.1 and P.sub.1 and connectivity as illustrated in FIG. 2a. The biasing voltage circuit comprising of the voltage Ref 3-2 and of the current source I.sub.3 generates and applies a biasing voltage at node 1-5. In addition, this voltage is used to power the device N.sub.2. FIG. 3b shows an embodiment where the inventive concept has been extended to a balanced output circuit. The biasing voltage circuit applies the biasing voltage to the LC Load1 2-2 which then applies the biasing voltage to the nodes 3-4 and 3-6. Because direct coupling is utilized, the biasing voltage is directly coupled to the gates of devices N.sub.5 and N.sub.7, respectively. The LC Load2 2-3 presents a high impedance high frequency load to the devices N.sub.5 and N.sub.7 at the operating frequency. V.sub.in is applied to the gate of N.sub.6 while V.sub.in (180 degrees out of phase) is applied to the gate of N.sub.4. The balanced output V.sub.out and V.sub.out are available at nodes 3-7 and 3-5, respectively.
(37) Reducing the trace lengths between the stages minimizes the overall inductance in the network. In addition, other benefits include: smaller die area, lower cost, lower power (drive less parasitic capacitance/inductance), and reduced design time. When the network of the parasitic capacitances or inductance increases, the simulation of the circuit slows down and requires a long time to evaluate. By minimizing the distance between stages, the area of the die containing the stages is decreased. As the area of the die decreases, the network of the parasitic capacitances or inductance decreases offering quicker simulation results.
(38) Finally, it is understood that the above description are only illustrative of the principle of the current invention. Various alterations, improvements, and modifications will occur and are intended to be suggested hereby, and are within the sprit and scope of the invention. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that the disclosure will be thorough and complete, and will fully convey the scope of the invention to those skilled in the arts. It is understood that the various embodiments of the invention, although different, are not mutually exclusive. In accordance with these principles, those skilled in the art may devise numerous modifications without departing from the spirit and scope of the invention. This inventive technique is applicable to direct biasing the high frequency design of a multi-stage circuit. The stage can have active electronics, reactive loads and resistance or any combination therein. It is a challenging layout task to minimize all parasitic inductance and capacitance between, as well within, stages in order to operate the circuit at the smallest possible area in an integrated circuit. As the area is reduced, the propagation time is also reduced. This allows the RF designer to extend the concept to even higher frequency circuits for a given technology. The LC circuit can be set to resonate at frequencies above or below 60 GHz to select other frequency bands within the allotted spectrum. At electrical resonance, the parallel LC circuit can also be called a resonant parallel LC circuit. A trace can be a metallic interconnect that couples one node to another node. Many portable wireless systems as well as non-portable systems can benefit from the inventive techniques presented here. In addition, the network and the portable system can exchange information wirelessly by using communication techniques such as TDMA (Time Division Multiple Access), FDMA (Frequency Division Multiple Access), CDMA (Code Division Multiple Access), OFDM (Orthogonal Frequency Division Multiplexing), UWB (Ultra Wide Band), WiFi, WiGig, Bluetooth, etc. The network can comprise the phone network, IP (Internet protocol) network, LAN (Local Area Network), ad hoc networks, local routers and even other portable systems.
