Multiple stream transmission method comprising multicarrier modulation selection according to the associated communication type

Abstract

A method for transmitting multiple communications of different types, in particular sporadic (MTC) or cellular (broadband) communication including symbols to be transmitted, corresponding to communication services implementing a modulation having M subcarriers of the FBMC-OQAM or OFDM-OQAM type. The method uses linear frequency filtering of a sequence of length N including symbols having L coefficients that are parametrizable according to the communication, L, N, and M being natural numbers, so as to generate N+L1 symbols, reducing out-of-band spurious emissions. The method also uses, no matter what the communication is, a single frequency/time transform (IFFT) having a size M, where N<M. For sporadic communications (MTC) in a lower frequency band, the constraints on the out-of-band side-lobes are more stringent on the filter and FBMC modulation is thus better adapted.

Claims

1. A multiple stream transmission method performed by a telecommunication device, comprising: transmitting symbols corresponding to different types of communications including a sporadic type and a cellular type by a multiple stream telecommunication device, wherein transmitting comprises: implementing a multicarrier modulation whatever the communication type using a same frequency time transform associated with a band of size M, N<M, and a linear frequency filtering using a filter, of a sequence of length N comprising symbols, with L coefficients that are parameterizable according to the communication type, L, N and M being natural numbers, for generating N+L 1 symbols, and wherein multiple sequences of symbols are processed in parallel with a parameterization of the filtering for each different communication type associated with the sequences and the filtered sequences are mapped to an input of the frequency time transform each on a sub-band, and wherein the number of symbols can be different between the sequences, and wherein the coefficients of the filter are equal to {1, w} with w[0,1], L=2.

2. The method as claimed in claim 1, in which the filtered symbols are mapped to the input of the frequency time transform and in which, during a framing, a first and a second formatted cyclic prefix are added to a multicarrier symbol obtained at an output of the transform.

3. The method as claimed in claim 2, in which the N symbols are precoded before being filtered, the precoding preequalizing a phase shift effect of the filtering.

4. The method as claimed in claim 1, in which w=0, in which the symbols to be transmitted are complex and in which the sequence of length N is composed of N complex symbols to be transmitted.

5. The method as claimed in claim 4, in which the filtered symbols are mapped to the input of the frequency time transform and in which the method further includes mapping pilots to the input of the frequency time transform.

6. The method as claimed in claim 1, in which w0, in which the symbols to be transmitted are complex and the sequence of length N is composed of successions of the real part and the imaginary part of the symbols to be transmitted.

7. The method as claimed in claim 6, in which the method further includes an insertion of complex pilots into the sequence of N symbols before the filtering such that for the same pilot of form P.sub.r+jP.sub.i its real part P.sub.r is inserted in the form P.sub.r/w and its imaginary part is inserted thereafter in the form jP.sub.i.

8. The method as claimed in claim 1, including transmitting a signaling message encoding the parameterization.

9. A multiple stream telecommunication device for symbols to be transmitted corresponding to different types of communications including sporadic type or cellular type implementing a frequency time transformation, wherein the device comprises: a processor, which is configured to process multiple sequences of symbols in parallel with a number of symbols which can be different between the sequences, and which includes as many linear frequency filters with L parameterizable coefficients according to the communication type as sequences in parallel for generating N+L1 symbols from a sequence of length N including symbols and includes a common frequency time transformer of band M whatever the communication type, L, N and M being natural numbers, N<M, and the filtered sequences are mapped to the input of the transformer each on a sub-band, and wherein the coefficients of the filters are equal to {1, w} with w[0,1], L=2.

10. A multiple stream telecommunication reception method performed by a telecommunication receiver, comprising: receiving symbols corresponding to different types of communications, including sporadic type or cellular type, implementing a multicarrier modulation with M subcarriers, using a same time frequency transform of size M and, under control of a received signaling message encoding a parameterization used in transmission, and using a linear frequency filtering, which is an inverse of a linear filtering used in transmission, in which the coefficients of the filter in transmission are equal to {1, w} with w[0,1], L=2.

11. A multiple stream telecommunication receiver for symbols corresponding to different types of communications, including a sporadic type or a cellular type implementing a multicarrier modulation with M subcarriers, wherein the receiver comprises a same time frequency transform of size M and, under control of a received signaling message encoding a parameterization used in transmission, a linear frequency filter, which is an inverse of a linear filter used in transmission and wherein the coefficients of the linear frequency filter in transmission are equal to {1, w} with w[0,1], L=2.

Description

LIST OF FIGURES

(1) Other features and advantages of the invention will become apparent during the following description made with reference to the appended figures, given by way of a non-restrictive example.

(2) FIG. 1 is a diagram of the transmission and reception chain of a conventional OFDM modulation.

(3) FIG. 2 is a diagram of the transmission chain of an FBMC modulation.

(4) FIGS. 3a and 3b are diagrams of the main processing implemented respectively at transmission and reception according to the multiple stream transmission method provided.

(5) FIG. 4 is a diagram of the precoding at transmission according to one embodiment of the invention corresponding to a modulation with QAM symbols.

(6) FIG. 5 is a diagram of the precoding at transmission according to one embodiment of the invention corresponding to a modulation with OQAM symbols.

(7) FIG. 6 is a diagram of the function HS of the precoding in FIG. 5.

(8) FIGS. 7a and 7b are diagrams of examples of the structure of the transmission and reception filters respectively for a sequence of N=5 symbols, coefficients of the filters equal to {1, w} and an OFDM modulation according to one embodiment of the invention.

(9) FIG. 8 is a diagram of the OFDM symbol constructed with QAM symbols, formatted thanks to the filter and to which the method adds a prefix at the beginning of the symbol and a prefix at the end of the symbol (postfix).

(10) FIG. 9 is a diagram of an exemplary implementation of the symbol decoding at the receiver in the absence of a filter at reception.

(11) FIG. 10 is a diagram of an exemplary implementation of the symbol decoding at the receiver in the presence of a filter at reception.

(12) FIG. 11 is a diagram of the structure of the transmission filter for a sequence of N=5 symbols, coefficients of the filters equal to {1, w} and an FBMC modulation according to one embodiment of the invention.

(13) FIG. 12 is a diagram of the structure of the transmission filter for a sequence of N=5 symbols, coefficients of the filters equal to {1, w} with an insertion of pilots into the sequence to be filtered and an OFDM modulation according to one embodiment of the invention.

(14) FIG. 13 is a diagram of the arrangement carried out on the received symbols when there has been an insertion of pilots at transmission in the case of an OFDM modulation according to one embodiment of the invention.

(15) FIG. 14 is a diagram of the structure of the transmission filter for a sequence of N=5 symbols, coefficients of the filters equal to {1, w} with an insertion of pilots into the sequence to be filtered and an OFDM modulation with OQAM symbols according to one embodiment of the invention.

(16) FIG. 15 is a diagram of a multiple stream transmission device with the transmission band of size M divided into sub-bands each dedicated to a communication of a given type.

(17) FIG. 16 is a diagram of a multiple stream transmission device with the transmission band M divided into sub-bands dedicated respectively to an MBB service and an MTC service.

(18) FIG. 17 is a diagram illustrating the adaptation of the sequences before filtering in the form of a temporal stretching for symbols of an MBB service.

(19) FIG. 18 is a diagram illustrating the adaptation of the mapped symbols in the protection sub-band with OQAM symbols and a half rate.

(20) FIG. 19 is a diagram illustrating the adaptation of the mapped symbols in the protection sub-band with QAM symbols and a full rate.

(21) FIG. 20 is a diagram illustrating the rate for FBMC modulation.

(22) FIG. 21 is a diagram illustrating the addition of cyclic prefixes onto a symbol at the IFFT output which combines the MTC or V2X and MBB services.

(23) FIG. 22 is a diagram illustrating the addition of cyclic prefixes onto a symbol at the IFFT output which is only composed of the MTC or V2X service.

(24) FIG. 23 is a diagram of the summation with overlapping between successive symbols after the addition of cyclic prefixes.

DESCRIPTION OF EMBODIMENTS OF THE INVENTION

(25) The transmission technique according to the invention is illustrated in FIGS. 3a and 3b. The multiple stream transmission method 1, 2 corresponding to different communication types uses a linear frequency filtering with L parameterizable coefficients according to the communication type, L and M being natural numbers. Such a linear filtering is a linear convolution: y(p)=.sub.n=0.sup.L-1x(pn)(n) of the input samples x(p) with the coefficients of a filtering function (n). Moreover, whatever the communication type the method uses the same frequency time transform of size M commonly termed IFFT.

(26) The modulated symbols, e.g. QAM complex symbols, resulting from the modulation MOD are grouped by block of N or sequence of symbols of length N, N<M. Optionally, the symbols are precoded by a precoder PreCOD. They are then filtered by a linear filter FIL with L coefficients in the frequency domain. The linear filtering consists of a linear convolution. Thus, if the input sequence has a length N then the filtered sequence has a length N+L1, N+L1 M. L1 is known as the convolution tail; it reduces the spectral efficiency. In noting a.sub.j with j[0, N1], the precoded symbols then the filtered symbols y.sub.k with k[0, N+L1] are expressed in the form:
y.sub.k=.sub.l=0.sup.L-1.sub.la.sub.k-l with .sub.l the coefficients of the filter, l[0,L1].

(27) The filtered symbols are then mapped by a mapper MAP to the input of the IFFT frequency time transform of size M.

(28) The filtering introduces interference between the subcarriers. A precoding may then be introduced to allow the receiver to cancel out the interference more easily. Since the filtering is linear, the function of the precoding is to preequalize the phase of the filter.

(29) In the case of a modulation constructed with QAM symbols, the precoding amounts to weighting the symbols with a coefficient G and then transforming them with an IFFT frequency time transform as illustrated in FIG. 4. The coefficient G has the value:

(30) $G=\frac{H_{\mathrm{ef}}^{*}}{.Math.H_{\mathrm{ef}}.Math.}$
with H.sub.ef the frequency coefficients of the equivalent filter. The equivalent filter is different from the filter at transmission if there is a filtering at reception. For example, if the filter at reception is sinusoidal then the frequency coefficients are {1, w}. When there is no filtering at reception then H.sub.ef=DFT({1, w}, N) with DFT a discrete Fourier transform of size N equal to the length of the sequence of symbols a.sub.j at the input of the precoder. When there is a filtering at reception of the type in FIG. 7b, then H.sub.ef=DFT({w,1+w,1}, N). The outputs of the IFFT then feed into the transmission filter.

(31) In the case of a modulation constructed with OQAM symbols, the precoding is illustrated in FIG. 5. FIG. 6 gives the details of the box HS in FIG. 5. The modulated QAM input symbols are considered to be distributed into two paths and weighted with a coefficient G. The weighted symbols are then arranged in a manner similar to that illustrated in FIG. 6 when the input sequence includes N/4=4 modulated symbols, a, b, c and d by paths. The output sequence of each path is transformed by an IFFT of size N/2. The output samples have real values. The outputs of one of the two IFFTs are multiplied by j to obtain imaginary values. The outputs of the other IFFT and the imaginary values are directed to the transmission filter input.

(32) The parameterization of the linear filter meets the constraints imposed on the transmitted signal, in particular the required shape of the spectrum, relating to the corresponding communication service. In particular, a broadcasting service, e.g. the MBMS service specified by 3GPP LTE, requires a more robust filter against the effects of spreading than a conventional voice communication service.

(33) FIGS. 7a and 7b are diagrams for illustrating the structure and the parameterization of an example of a sinusoidal shaped filter respectively at transmission and at reception for generating a formatted OFDM (shaped OFDM) modulation with QAM symbols.

(34) The example illustrated relates to N=5 symbols i.e. a, b, c, d and e. The filtering at transmission applies to N=5 symbols at the input and supplies N+1=6 samples at the output. The coefficients of the filter are {1, w} with w a weight of weighting, w[0,1]. The structure of the filter is such that the 1.sup.st output corresponds to the 1.sup.st symbol, a, weighted by the 1.sup.st coefficient, 1: a, the n.sup.th output is the result of the difference between the n.sup.th weighted symbol by the 1.sup.st coefficient, 1, and the (n1).sup.th weighted symbol with the 2.sup.nd coefficient, w, for 1<nN: bwa, cwb, dwc, ewd. The (N+1).sup.th output corresponds to the n.sup.th weighted symbol with the 2.sup.nd coefficient, w, to which the negative sign is assigned: we. This structure implements the linear convolution between the input symbols a, b, c, d and e and the coefficients {1, w} of the filter.

(35) The value w=0 makes it possible not to have a filtering function on the QAM input symbols: each of the N=5 symbols is found intact at the output. Such a parameterization amounts to mapping the QAM symbols directly at the input of the frequency time transform which makes it possible therefore to generate a conventional OFDM signal with a formatting (filtering) function identical to a gate function. Moreover, in this case there is no longer any need of precoding since the system is orthogonal. The parameterization of the linear filter with w=0 is, for example, suitable for an MBB service.

(36) The value w=1 makes it possible to filter with a temporal half-period sinusoidal function. The OFDM signal obtained is therefore filtered with a sinusoidal shape which improves the spectrum by reducing its frequency spread. The parameterization of the linear filter with w=1 makes it possible to construct a formatted modulation with a sinusoidal temporal profile.

(37) The temporal shape of the filter becomes a truncated and stretched sinusoidal shape when w]0,1[.

(38) Thus the parameterization of the value of the coefficient w makes it possible to format the temporal shape of the transmission filter.

(39) According to another example, the filter has the shape of a raised cosine. The cosine shape is defined over a temporal window of duration T, k[0, T1] equal to the size of the IFFT: T=M. The filter has a temporal expression:

(40) $F\left(k\right)=\{\begin{array}{cc}0.5\left(1+\cos \left(\frac{\left(1+k\right)}{M/2}\right)\right)& k\left[0,\left(M/2\right)-1\right]\\ 1& k\left[\left(M/2\right)-1,M-\left(M/2\right)\right]\\ 0.5\left(1+\cos \left(\frac{\left(k\right)}{M/2}\right)\right)& k\left[M-\left(\frac{M}{2}\right)+1,M-1\right]\end{array}$

(41) with the roll-off factor. The coefficients of the frequency filter are obtained after a time frequency transformation of the temporal coefficients. Given that these coefficients have a real value, only the real values at the FFT output are considered. Moreover, only the most significant coefficients are considered. According to a first selection criterion, the method only considers the 2L1 with L=round(2/), the most significant real values for constituting the coefficients.

(42) Optionally, the method adds one or more prefixes to the output symbol of the IFFT. FIG. 8 is a diagram of the OFDM symbol constructed with QAM symbols, formatted thanks to the filter and to which the method adds two prefixes, one at the beginning of the symbol and another at the end of the symbol (postfix). The prefixes are formatted. The addition of two prefixes is particularly useful when the filter has a sinusoidal shape and the coefficient w is close to zero. Indeed, they make it possible to limit the spreading of the spectrum.

(43) At reception as illustrated in FIG. 3b, the processing is the reverse of that at transmission. The prefixes are extracted from the received signal if they have been inserted at transmission. The received symbols are then transformed into frequency symbols by means of an FFT time frequency transform. The demapper MAP.sup.1 at the output of the FFT reorders the symbols by extracting them from the subcarriers according to a reverse order of that at transmission. The ordered symbols are optionally filtered by a filter FIL.sup.1. Conventional processing then goes on to perform an estimation of the channel, an equalization of the channel and optionally a MIMO decoding. Finally, the symbols are decoded by means of a decoder DeCOD if a precoding has occurred at transmission.

(44) The structure of the reception filter FIL illustrated in FIG. 7b applies to N+1=6 input samples and supplies N+2=7 samples at the output. The structure of the filter is such that:

(45) the 1.sup.st output corresponds to the 1.sup.st sample received, i.e. the 1.sup.st sample from the filtering at transmission which corresponds to the 1.sup.st symbol a weighted by the 1.sup.st coefficient, 1: 1a=a,

(46) the n.sup.th output is the result of the difference between the (n1).sup.th received sample and the (n).sup.th received sample, 1<nN+1,

(47) the (N+2).sup.th output corresponds to the N.sup.th received sample, i.e. the (N+1).sup.th sample from the filtering at transmission corresponding to the N.sup.th symbol weighted with the 2.sup.nd coefficient, w, to which the negative sign is assigned: we.

(48) In the absence of filter FIL.sup.1 at reception, the symbol decoding DeCOD illustrated in FIG. 9 reduces the sequence of N+1 received symbols into a sequence of N symbols and feeds the FFT with these N symbols. The box in dashed lines represents the action of the filter at transmission on the symbols. The reduction consists in adding together the first and the last symbols of the received sequence. This reduction makes it possible to construct a cyclic convolution. The result of the addition of these two symbols is then taken as the first symbol at the FFT input. At the FFT output, the symbol decoding equalizes the symbols at the FFT output by weighting the symbols with the coefficient G2=|H.sub.ef|.

(49) In the presence of filter FIL.sup.1 at reception, the symbol decoding DeCOD illustrated in FIG. 10 reduces the sequence of N+2 received symbols into a sequence of N symbols and feeds the FFT with these N symbols. The box in dashed lines represents the action of the filter at transmission on the symbols. The reduction consists in subtracting the first symbol received from the penultimate symbol received and adding together the second and the last symbols of the received sequence. This reduction makes it possible to construct a cyclic convolution. The result of the subtraction is taken as the first symbol at the FFT input. And the result of the addition is taken as the second symbol at the FFT input. At the FFT output, the symbol decoding equalizes the symbols at the FFT output by weighting the symbols with the coefficient G2=|H.sub.ef|.

(50) When the modulation to be generated is an FBMC modulation, the transmission chain is identical to that previously described, in particular the filter at transmission has exactly the same structure as that illustrated in FIG. 7a. On the other hand, the symbols at the filter input are taken into account differently. This difference illustrated in FIG. 11 amounts to considering the real part and the imaginary part of each QAM symbol separately, e.g. a=a.sub.r+ja.sub.i, and to considering them in sequence. Thus by keeping the same length (N) of sequence as for FIG. 7a, the method considers that the sequence is formed of {a.sub.r, ja.sub.i, b.sub.r, jb.sub.i, c.sub.r} and not formed of {a, b, c, d, e}.

(51) Pilots may be inserted into the formatted OFDM symbols generated. This insertion takes place between the precoding when it exists and the linear filtering.

(52) In the case of a modulation with QAM symbols, the description of the insertion of the pilots is illustrated in FIG. 12 which corresponds to the case of a sinusoidal shaped filter with coefficients {1, w} such that w0. The pilots are generally complex and may be written P=P.sub.r+jP.sub.i, with P.sub.r the real part and P.sub.i the imaginary part. The real part of the pilot is inserted into the input sequence of the filter in the form P.sub.r/w and the imaginary part jP.sub.i is inserted afterward in the sequence. Given the structure of the filter, the pilot P=P.sub.r+jP.sub.i is identical on one of the outputs and according to the illustration on the 4.sup.th output.

(53) Knowing the sequence arrangement used at transmission, the receiver is capable of performing a channel estimation on the basis of the transmitted pilot P=P.sub.r+jP.sub.i. Knowing the transmitted pilot, the receiver is able, with reference to FIG. 13, to add the quantity P.sub.r/w to the third sample in the received sequence for retrieving wb and to add the quantity jwP.sub.i to the fifth sample of the received sequence for retrieving c bearing in mind that the fourth sample, the pilot P, does not appear in the figure. By adding the two results, the receiver finds the sample cwb that was missing from the received sequence in accordance with what is illustrated in the dashed block in FIG. 9.

(54) In the case of a modulation with OQAM symbols, the insertion of the pilots is illustrated in FIG. 14 which corresponds to the case of a sinusoidal shaped filter with coefficients {1, w} such that w0.

(55) According to one implementation the transmission band of size B is divided into sub-bands, the band B is given by the size of the IFFT: B=M. Each sub-band may be dedicated to a service and different services may be dedicated to different sub-bands. The sub-bands may be of a different size. A corresponding transmission device is illustrated in FIG. 16. According to the illustration, the formatted OFDM symbols at the IFFT output include symbols associated with at least three different services. Two services, e.g. MBB services, are associated with an OFDM modulation with QAM symbols and one service, e.g. V2X or MTC, is associated with an FBMC modulation with OQAM symbols. The parameterization is specific to each service. The symbols associated with a service are processed according to the processing illustrated and detailed in FIG. 3a. The only difference with respect to FIG. 3a is that the filtered symbols associated with the service are mapped onto a sub-band of the band M and not onto the total band M. The invention thus makes it possible to simultaneously transmit symbols emanating from different services and requiring conventional OFDM and FBMC modulations which is not allowed by the existing modulators. The structure provided is indeed very flexible and easily configurable.

(56) Thus, with a transmission device constructed around a single IFFT frequency time transform and parameterizable filters of the same structure, the invention makes it possible to simultaneously transmit MBB, MTC and V2X communications. Such an embodiment is more fully described below in the case of a simultaneous transmission of an MTC service and an MBB service illustrated in FIG. 16. The MTC service is rendered with an OFDM/OQAM modulation and the MBB service is rendered with an OFDM modulation. The system therefore simultaneously processes different types of symbols, QAM and OQAM for simultaneously transmitting the MBB and MTC services. A sub-band is assigned to each of the services, MTC band (OQAM signaling) and MBB band (QAM signaling). The two sub-bands are separated by a protection sub-band MBB protection band. This sub-band forms part of the sub-band of one of the two services so as not to lose in spectral efficiency. However, the symbols mapped onto this sub-band have a processing suitable for limiting the interference between the MTC and MBB sub-bands given the non-orthogonality between the symbols mapped onto the MTC and MBB sub-bands.

(57) Given that the IFFT of an OFDM/OQAM modulation has a frequency of implementation twice as high as that of the IFFT of a conventional OFDM modulation, the sequence of symbols to be filtered for the OFDM modulation has to be adapted.

(58) For the QAM symbols mapped onto the MBB sub-band, one embodiment of this adaptation in the form of a temporal stretching is described in relation to FIG. 17.

(59) The IFFT which is common to the conventional OFDM and OFDM/OQAM modulations is implemented with the highest frequency, i.e. 2/T0. The symbols to be filtered for the conventional OFDM modulation arrive in sequence/block with a frequency of 1/T0. The sequences must therefore be stretched for being transformed with the IFFT at the frequency 2/T0. The simplest stretching consists in inserting a sequence of null symbols between each sequence of symbols to be filtered. The parameterization of the filter is such that the coefficients are {1, 0} for obtaining a conventional OFDM modulation.

(60) According to the described embodiment, the protection band is taken in the sub-band allocated to the MBB service.

(61) The QAM symbols mapped in the protection sub-band may be adapted according to two possibilities. According to the first possibility illustrated in FIG. 18, the method uses OQAM symbols with a half rate. The QAM input symbols are therefore put in sequences to be processed as OQAM symbols but with a half rate, i.e. T0. Two successive sequences are separated by a sequence of zeros. The coefficients of the filter are {1, w}, with w=1. According to the second possibility illustrated in FIG. 19, the method uses QAM symbols with a full rate and a filter of coefficients {1, w}, with 0<w<1. A precoding is then necessary.

(62) The first possibility does not require any precoding which makes it possible for the receiver to detect the symbols directly after the IFFT and for the detector to benefit from the frequency diversity of the channel. But the half rate causes spectral efficiency to be lost on the protection sub-band. The second possibility provides a better spectral efficiency but requires a precoding. This then calls for a symbol decoding on the receiver before detection which does not allow the detector to benefit from the frequency diversity of the channel. The two possibilities each allow the receiver to efficiently filter the received symbols for reducing the interference between the MBB and MTC sub-bands.

(63) For the OQAM symbols mapped onto the MTC sub-band, the QAM input symbols are put in sequences for being processed as OQAM symbols. They therefore arrive before filtering in sequence at the rate of T0/2 since the real part and the imaginary part of each symbol are considered in series in the sequences to be filtered as illustrated in FIG. 20. There is no rate adaptation. The parameterization of the filter is such that the coefficients are {1, 1} for obtaining an OFDM/OQAM modulation.

(64) The output of the IFFT therefore supplies every T0 a signal that combines the MTC and MBB services and every T0+T0/2 a signal composed only of the MTC service. The width of the protection band is a parameter of the system and depends on the power offset between the MBB and MTC services. For example, if this offset is zero then only a few subcarriers are necessary for the protection sub-band.

(65) Every T0, the method adds a cyclic prefix CP formatted at the beginning and at the end of the IFFT output symbol, as illustrated in FIG. 21. The cyclic prefix CP added at the beginning of the IFFT output symbol is a copy of the last L samples of the M samples of the symbol. The cyclic prefix CP added at the end of the IFFT output symbol is a copy of the first L samples of the M samples of the symbol. The cyclic prefixes are formatted with a window of a particular shape. The symbol thus constituted is denoted by US-OFDM; it appears at a rate of T0.

(66) Every T0+T0/2, the method adds a prefix formatted according to a concatenation different from that previously described. The concatenation is illustrated in FIG. 22. The IFFT output symbol benefits from a well localized spectrum taking into account the filtering {1, 1} of the OFDM/OQAM modulation. There is therefore no need to add cyclic prefixes at the edges of the symbol. For reasons of orthogonality, the method inserts two cyclic prefixes in the middle of the symbol of length M. Thus, the method inserts a copy of the last L samples of the first part of M/2 samples of the symbol at the beginning of the second part of M/2 samples. The method further inserts a copy of the first L samples of the second part of M/2 samples of the symbol at the end of the first part of M/2 samples. The method formats these copies so that the middle samples equal zero.

(67) The method then proceeds to a summation with overlapping between consecutive symbols as illustrated in FIG. 23. The overlapping consists of

(68) $\frac{M}{2}+L$
samples. Thus, a symbol at T0, i.e. combining MTC and MBB services, is added together with the next symbol at T0+T0/2, i.e. which consists only of the MTC service. This addition takes place only between the last part of length

(69) $\frac{M}{2}+L$
of the symbol at T0 and the first part of the symbol at T0+T0/2 of length

(70) $\frac{M}{2}+L.$
Similarly, a symbol at T0+T0/2 is added together with the next symbol at 2T0 over an overlapping length of

(71) $\frac{M}{2}+L$
samples. And so on for the different symbols. [3GPP] Motivation for new WI on Low Complexity and Enhanced Coverage LTE UE for MTC, 3GPP TSG RAN Meeting #64 RP-140845, Sophia Antipolis, France, 10-13 Jun. 2014 [Bellanger] FBMC physical layer: a primer, M. Bellanger, PHYDYAS, June 2010