Abstract
The present invention relates to a method and a device for stabilization of amplitude of a mechanical vibration of a mechanical resonator in a microelectromechanical sensor device. The method comprises exciting the mechanical resonator with an oscillating excitation force by an input transducer. The input transducer is driven with an input AC voltage having essentially constant amplitude at a frequency that deviates from the resonant frequency of the mechanical resonator by a first frequency difference. The first frequency difference is configured to stabilize the amplitude of the mechanical vibration.
Claims
1. A method for stabilizing amplitude of a mechanical vibration of a mechanical resonator in a microelectromechanical sensor device, the method comprising: exciting the mechanical resonator into the mechanical vibration with an oscillating excitation force by an input transducer; and driving the input transducer with an input AC voltage having an essentially constant amplitude, wherein the driving with the input AC voltage occurs at a frequency that deviates from the resonant frequency of the mechanical resonator by a first frequency difference, and wherein the first frequency difference is configured to stabilize the amplitude of the mechanical vibration.
2. The method of claim 1, wherein the method further comprises: causing the stabilizing by maintaining the amplitude of the mechanical vibration essentially at a pre-defined constant value by adjusting the first frequency difference.
3. The method of claim 2, wherein the method further comprises: providing an output AC voltage induced by the mechanical vibration; controlling the adjusting the first frequency difference on basis of a detected amplitude difference between a detected amplitude of the output AC voltage and a preset target amplitude value.
4. The method of claim 3, wherein the method further comprises: detecting a first phase difference between the input AC voltage and the output AC voltage; controlling the first frequency difference on basis of the first phase difference.
5. The method of claim 3, wherein the method further comprises: causing a phase shift to the output AC voltage, wherein the phase shift produces a phase shifted output AC voltage.
6. The method of claim 5, wherein the method further comprises: detecting a second phase difference between the phase shifted output AC voltage and the input AC voltage; and controlling of the first frequency difference on the basis the second phase difference.
7. The method of claim 5, wherein the method comprises: generating the input AC voltage by amplifying the phase shifted output AC voltage.
8. The method of claim 1, wherein the method comprises: generating the input AC voltage by a voltage controlled oscillator.
9. The method of claim 1, wherein the microelectromechanical sensor device comprises any one of a gyroscope and an induction type magnetometer.
10. The method of claim 1, wherein the first frequency difference is between +/0.0001 and +/0.001 times the resonant frequency.
11. A microelectromechanical sensor device comprising: a mechanical resonator, a drive loop circuitry for stabilizing amplitude of a mechanical vibration of the mechanical resonator, and an input transducer configured to excite the mechanical resonator into the mechanical vibration with an oscillating excitation force, wherein the drive loop circuitry is configured to drive the input transducer with an input AC voltage having an essentially constant amplitude, and wherein the driving with the input AC voltage is configured to occur at a frequency that deviates from a resonant frequency of the mechanical resonator by a first frequency difference, and wherein the first frequency difference is configured to stabilize the amplitude of the mechanical vibration.
12. The microelectromechanical sensor device of claim 11, wherein the drive loop circuitry further comprises: a feed-back circuitry configured to cause the stabilizing by maintaining the amplitude of the mechanical vibration of the mechanical resonator at a desired constant value by adjusting the first frequency difference.
13. The microelectromechanical sensor device of claim 12, further comprising: an output transducer configured to provide an output AC voltage induced by the mechanical vibration; and wherein the feed-back circuitry comprises: an amplitude control circuitry configured to control the adjusting the first frequency difference on basis of a detected amplitude of the output AC voltage and a preset target amplitude value.
14. The microelectromechanical sensor device of claim 13, wherein the feed-back circuitry further comprises: a phase detecting circuitry configured to detect a first phase difference between the input AC voltage and the output AC voltage; and a controller circuitry configured to control the first frequency difference on basis of the first phase difference.
15. The microelectromechanical sensor device of claim 13, wherein: the amplitude control circuitry is configured to cause a phase shift to the output AC voltage, wherein the phase shift produces a phase shifted output AC voltage.
16. The microelectromechanical sensor device of claim 15, wherein the feed-back circuitry further comprises: a phase detecting circuitry configured to detect a second phase difference between the phase shifted input AC voltage and the output AC voltage; and a PLL controller circuitry configured to control the adjustment of the first frequency difference on basis of the second phase difference.
17. The microelectromechanical sensor device of claim 15, wherein the drive loop circuitry comprises an amplifier configured to generate the input AC voltage by amplifying the phase shifted output AC voltage.
18. The microelectromechanical sensor device of claim 11, wherein the drive loop circuitry comprises a voltage controlled oscillator configured to generate said input AC voltage.
19. The microelectromechanical sensor device of claim 11, wherein the microelectromechanical sensor device is any one of a gyroscope and an induction type magnetometer.
20. The microelectromechanical sensor device of claim 11, wherein the first frequency difference is between +/0.0001 and +/0.001 times the resonant frequency.
Description
BRIEF DESCRIPTION OF THE DRAWINGS
(1) In the following the invention will be described in greater detail, in connection with preferred embodiments, with reference to the attached drawings, in which
(2) FIG. 1 is an illustration of an envelope curve of amplitude rise of an oscillating resonator.
(3) FIG. 2 illustrates a MEMS device with AGC feed-back loop.
(4) FIG. 3 is an illustration of an envelope curve of amplitude rise of an oscillating resonator with AGC.
(5) FIG. 4 illustrates capacitive coupling in a MEMS gyroscope.
(6) FIG. 5 illustrates a MEMS device using DC biasing.
(7) FIG. 6 is an illustration of frequency responses of resonators with different Q-values.
(8) FIG. 7 illustrates phase difference as a function of relative frequency deviation.
(9) FIG. 8a illustrates in-phase component of input AC-voltage as function of resonance gain.
(10) FIG. 8b illustrates out-of-phase component of input AC-voltage as function of resonance gain.
(11) FIG. 9 is a first illustration of a transient response envelope curve.
(12) FIG. 10 is a second illustration of a transient response envelope curve.
(13) FIG. 11 is a magnification of part of envelope curve of FIG. 10.
(14) FIG. 12 is a third illustration of a transient response envelope curve.
(15) FIG. 13 is a magnification of part of envelope curve of FIG. 12.
(16) FIG. 14 is an illustration of resonance gain as function of a phase difference.
(17) FIG. 15 illustrates a first embodiment of a MEMS sensor device with a drive loop circuitry.
(18) FIG. 16 illustrates a second embodiment of a MEMS sensor device with a drive loop circuitry.
(19) FIG. 17 illustrates a third embodiment of a MEMS sensor device with a drive loop circuitry.
(20) FIG. 18 illustrates a fourth embodiment of a MEMS sensor device with a drive loop circuitry.
DETAILED DESCRIPTION
(21) The following embodiments are exemplary. Although the specification may refer to an, one, or some embodiment(s), this does not necessarily mean that each such reference is to the same embodiment(s), or that the feature only applies to a single embodiment. Single features of different embodiments may be combined to provide further embodiments.
(22) As used in this application, the term circuitry refers to all of the following: (a) hardware-only circuit implementations, such as implementations in only analog and/or digital circuitry, wherein the circuitry may comprise discrete and/or integrated components, and (b) combinations of circuits and software (and/or firmware), such as (as applicable): (i) a combination of processor(s) or (ii) portions of processor(s)/software including digital signal processor(s), software, and memory(ies) that work together to cause an apparatus to perform various functions, and (c) circuits, such as a microprocessor(s) or a portion of a microprocessor(s), that require software or firmware for operation, even if the software or firmware is not physically present.
(23) This definition of circuitry applies to all uses of this term in this application, including use of discrete and integrated electronic circuitry and any combination thereof. As a further example, as used in this application, the term circuitry would also cover an implementation of merely a processor for multiple processors, or a portion of a processor and its (or their) accompanying software and/or firmware. The term circuitry would apply to one or more application specific integrated circuit ASIC(s) or applications processor integrated circuit(s) for a microelectronic sensor device or system. Terms controller, detector and shifter refer to circuitry configured for a specific task.
(24) The first condition, independence of the stable resonance amplitude from Q-value of the resonator, may be achieved by introducing to the driving input AC voltage a frequency difference f from the resonant frequency f.sub.0 of the mechanical resonator. The FIG. 6 shows an exemplary force to amplitude frequency response of a mechanical resonator at frequencies close to the resonant frequency f.sub.0. Three different responses of devices with three different Q-values are illustrated. At the resonant frequency f.sub.0 of the mechanical resonator, amplitude varies significantly depending on the Q-value. Thus, exciting mechanical resonators with such different Q-values into oscillations with equal amplitudes would require using significantly different amplitude input AC voltages. However, it can be noticed that at frequencies deviating from the resonant frequency by a frequency difference f, the amplitude responses of mechanical resonators become practically independent of the Q-values. In other words, when the input transducers of mechanical resonators are driven by an essentially constant amplitude input AC voltage having a frequency deviating by f from the resonant frequency f.sub.0, the steady state of the resonance of mechanical resonators with different Q-values depend essentially on the frequency difference f, but are essentially independent on the Q-values of the mechanical resonators. The frequency difference f may be achieved by decreasing or increasing the frequency of the input AC voltage from the resonant frequency f.sub.0.
(25) In the following analysis, the input AC voltage is assumed to have a frequency difference to the mechanical resonant frequency of the resonator. The amplitude of the input AC voltage is variable in order to adjust the mechanical oscillation amplitude to a constant desired level. As illustrated in FIG. 6, in order to stabilize the amplitude of the mechanical resonator to an exemplary value of 2000 arbitrary amplitude units, the frequency difference f may be selected either above or below the resonant frequency f.sub.0. In an exemplary embodiment, the frequency difference f may be set to +/0.000245 times the resonant frequency f.sub.0. If the resonant frequency f.sub.0 equals for example 20 kHz, the frequency difference f of approximately 4.9 Hz causes the resonator to be excited into a resonance oscillation with an amplitude of approximately 2000 arbitrary units. Variation of the amplitude between a resonator with Q-value of 14000 and a resonator with Q-value of 7000 is approximately 80 arbitrary units, which corresponds to just 4% of the resonator amplitude. The absolute input voltage needed at this frequency difference is 5 times larger than when driving at the resonant frequency. Resonance gain at the frequency with this frequency difference f is 2000, which is one fifth of the resonance gain of 10 000 at the resonant frequency f.sub.0. Thus, the maximum absolute amplitude variation of the required variable input AC voltage for driving oscillation with constant amplitude at a frequency that is offset from the resonant frequency f.sub.0 by this exemplary frequency difference f is in level of 5 times 4%=20% of the corresponding amplitude variation of the required variable input AC voltage needed for driving oscillation with constant amplitude with resonance frequency drive. In another exemplary embodiment, the frequency difference f may be set to +/0.000498 times the resonant frequency f.sub.0. If the resonant frequency f.sub.0 equals the same exemplary 20 kHz, the frequency difference f is approximately 10 Hz, and the variation of the mechanical oscillation amplitude between a resonator with Q-value of 14000 and a resonator with Q-value of 7000 is just 8 arbitrary units, which corresponds to just 0.8% of the resonator oscillation amplitude. The absolute input voltage needed at this frequency difference is 10 times larger than when driving at the resonant frequency f.sub.0. Resonance gain at the frequency with this exemplary frequency difference f is 1000, while the resonance gain at the resonant frequency is 10 000. The maximum absolute amplitude variation of the required variable input AC voltage for driving a mechanical oscillation with constant amplitude at a frequency that is offset from the resonant frequency f.sub.0 by this exemplary frequency difference f is in level of 10 times 0.8%=8% of the corresponding amplitude variation needed at resonance frequency drive. Thus, it can be noticed that the larger the frequency difference f, the smaller the effect of the Q-value on the required input AC-voltage amplitude or, if the input AC voltage amplitude is kept constant, on the mechanical oscillation amplitude. Preferably, the frequency difference f is between +/0.0001 and +/0.001 times the resonant frequency f.sub.0.
(26) The second condition is that the phase of the capacitively coupled voltage should preferably be in quadrature phase relative to the Coriolis-force induced voltage at the output transducer of the secondary resonator. This condition is also met with the frequency difference f stabilization method. FIG. 7 shows the phase of the input AC voltage with respect to the mechanical oscillation of the resonator. With large or moderate frequency difference f values the input AC voltage has either close to 0 or close to (180) phase relation to the mechanical resonator oscillation phase. This means that the input AC voltage is close to quadrature relation to the Coriolis-force induced output voltage since the Coriolis-force is in quadrature phase with respect to the mechanical oscillation.
(27) Some in-phase component also exists, as illustrated in FIGS. 8a and 8b. FIG. 8a illustrates in-phase component of the input AC-voltage as function of resonance gain at stabilized amplitude. In-phase component varies slightly depending on the Q-value, as indicated by the three exemplary plots for different Q-values. For example, FIG. 8a shows that in an exemplary device with Q-value of 10000, at resonance gain of 100, the in-phase component is just 0.01, in other words 1% of the input AC voltage, and share of the in-phase component rises steadily to level of 10% when the resonance gain rises to level of 1000. FIG. 8b illustrates share of the quadrature component of the input AC-voltage. Although on very high resonance gain values dependency can be seen between different Q-values, share of the quadrature phase component of the input AC voltage remains essentially stable essentially at level of 100% (1) at any resonance gain value below 1000, independent of the Q-value. Thus, the unwanted in-phase component is orders of magnitude smaller than the wanted quadrature phase component. The in-phase component depends on the frequency difference f, and also on the Q-value.
(28) Combining the effects of the two desirable conditions at resonance gain 1000 and Q-value 10 000 we find that the total change of the in-phase component of the capacitively coupled signal due to Q-value variation is only 0.8% of the variation found in the conventional resonance drive case (8% multiplied by 10%). At resonance gain 2000 the coupled in-phase signal is 4% and at gain 500 it is 0.2% of the conventional coupled signal. This analysis proofs that by driving the input transducer at a frequency difference to the resonant frequency, the input AC voltage amplitude will, compared to the resonant frequency drive, have a greatly reduced dependency on the Q-value variation of the resonator and thus the input AC voltage amplitude will be essentially constant.
(29) Let us next have a look at a transient response of the mechanical oscillation amplitude, when the input AC voltage has been switched on at time=0, and simulate the amplitude rise towards a steady state where the oscillation amplitude is essentially stable. For a practical MEMS sensor application, it is important to confirm that no detrimental side effects will arise during switch-on or due to other possible transients during the steady state operation of the MEMS device operation.
(30) FIG. 9 illustrates a transient response for a resonator, when the input transducer is driven with an essentially constant amplitude input AC voltage at relative frequency difference f=0.0025 and Q-value 10000 over a number of oscillation cycles of the mechanical resonator. FIG. 10 illustrates a transient response for a resonator, when the input transducer is driven with an input AC voltage at relative frequency difference f=0.0075 and Q-value 10000. FIGS. 9 and 10 show that rise-time of the amplitude with frequency difference f driving is as fast as with resonance excitation shown in FIGS. 1 and 3, and in the steady state the amplitude approaches to that predicted by FIG. 6. FIGS. 10 and 11 show however, that some fluctuation of the amplitude takes place around the steady state value over time. FIG. 11 shows a magnification of the tail of the amplitude envelope curve of FIG. 10, and it shows that this fluctuation will continue in small scale for thousands of cycles. If the Q-value is decreased to 3000, this fluctuation can be decreased as shown in FIG. 12, and further in FIG. 13 that shows a magnification of the tail of the amplitude envelope curve of FIG. 12. However, this decrease of fluctuation by decreasing the Q-value is done at cost of increasing capacitive coupling variation.
(31) In order to improve stabilization of the vibration amplitude in transient situations, a feed-back system may be introduced, that reduces the fluctuations illustrated in FIGS. 9 to 13 occurring if no stabilizing feed-back system is in place. A system according to FIG. 2 may be used in combination with driving the input transducer with an AC voltage having a frequency difference to the resonant frequency of the mechanical resonator. In this case the amplitude of the input AC voltage is not totally independent of the Q-value variation, but the dependency is still greatly reduced according to the previous analysis.
(32) An improved stabilization can be achieved if the input transducer is driven with an AC-voltage having essentially constant amplitude independent on the Q-value variation of the resonator. In this improved system the stabilization feed-back circuitry shall actively change the frequency difference f to counteract the variation of the vibration amplitude. A direct feed-back to, for example, a voltage controlled oscillator is not possible, since for every desired mechanical oscillation amplitude level, there are two possible frequency difference f values. A positive or negative feed-back system would be confused, since the feed-back would need to be positive on one side and negative on the other side of the resonant frequency f.sub.0. Thus, it is necessary to be able to distinguish on which side of the resonant frequency f.sub.0 the resonator is vibrating and preferably, use only the frequency difference f values on one side of the resonant frequency f.sub.0 for controlling the resonator amplitude.
(33) One exemplary way to limit the used frequency difference f values to only one side of the resonance frequency f.sub.0 in the resonance curve of FIG. 6 and thus to one of the two theoretically possible frequency difference f values without causing confusion in the feed-back system is to use the phase difference between the input AC voltage and the mechanical resonator oscillation as an additional control variable. The phase dependence of an input AC voltage and an output AC voltage due to polarity and amount of frequency difference f illustrated in FIG. 7 can be utilized for phase to frequency conversion. The phase shift of the input AC voltage with respect to the detected output AC voltage may be forced to a value range between 0 and /2 or between /2 and , for operation below or above the resonant frequency f.sub.0, respectively.
(34) It can be noticed that phase shift to resonance gain relation is quite linear in the range of phase differences between 0 and /4, or resonance gains below 70% of the maximum gain, as illustrated in FIG. 14. This relation between the phase difference and resonance gain is strongly dependent on the Q-value of the resonator, but this does not cause major problems since the exact value of the phase shift does not necessarily need to be known when negative feed-back is used.
(35) FIG. 15 shows a first exemplary embodiment block diagram of a microelectromechanical sensor device with a MEMS part and a drive loop circuitry implementing the above disclosed inventive principles. The MEMS part comprises the mechanical resonator and functionality of input and output transducers. The block diagram illustrates functionalities of the MEMS part, but not the specific structure of the MEMS part. The input and/or output transducers may be partially physically separate from the resonator, for example in a capacitive transducer, or they may be part of the resonator structure, as for example in a piezoelectrical transducer. The MEMS part functionality may be disposed on the same chip with the drive loop circuitry, or it may be a separate chip electrically coupled with the drive loop circuitry. The essentially constant amplitude input AC-voltage from a voltage controlled oscillator VCO 505 is fed to the input transducer 110, which excites the mechanical resonator 100. Oscillation of the mechanical resonator 100 is detected with an output transducer 120, which provides at its output an output AC voltage corresponding to the resonator movement. The frequency and phase of the VCO 505 is adjusted by a controller 500, so that the input AC voltage has a set frequency difference f from the natural resonant frequency f.sub.0 of the mechanical resonator. A phase detector 515 compares phases of the input AC voltage and the output AC voltage, and provides a phase difference information at its output. The controller 500 receives at one of its inputs the phase difference information between the input AC-voltage and the output AC voltage. In a second input of the controller 500, information on amplitude of the output voltage is received from an amplitude detector 510. The amplitude information may be provided in form of a DC-voltage. A skilled person knows that various types of signals may be provided by the amplitude detector 510 and the phase detector 515 for indicating the amplitude information and phase difference information respectively. For example, signals may be analog or digital, and signals may comprise voltage or current. A preset value of amplitude, which corresponds to a desired value of the amplitude of the mechanical resonator 100 motion in steady state oscillation is provided to the controller 500 through a control input. The controller 500 controls the frequency and phase of the VCO 505 in order to produce an amplitude stabilized oscillation of the mechanical resonator 100 with the preset amplitude value based on the phase difference information provided by the phase detector 515 and the detected amplitude information provided by the amplitude detector 510.
(36) FIG. 16 shows a second exemplary embodiment block diagram of a microelectromechanical sensor device with a MEMS part and a drive loop circuitry implementing the above disclosed inventive principles and which embodiment represents a more detailed exemplary implementation of the more general embodiment of FIG. 15. Control polarity of the VCO 505 is assumed positive: an increase in the control voltage will increase the frequency difference f. With this polarity selection, and for operation at frequency difference f below the resonant frequency f.sub.0, the output of the XOR circuit 611 may be fed to the inverting input of a first amplifier A1 601. The essentially constant amplitude AC-voltage received from the voltage controlled oscillator VCO 505 is fed to the input transducer 110 for exciting the mechanical resonator 100. The frequency and phase of the VCO 505 is adjusted by the first amplifier A1 601, which feeds a controlling voltage towards the VCO 505. The VCO 505 generates the essentially constant amplitude input AC voltage to the input transducer 110. Thus, the VCO 505 is an exemplary embodiment of a drive circuit configured to generate the input AC voltage. Phase detector functionality illustrated with reference 515 in FIG. 15, is implemented with an exclusive or circuit XOR 611, which is preceded by sinusoidal-to-square wave conversion circuitry 610, which converts both received AC voltages. The phase detector circuitry (610, 611) will have a zero voltage output for phase difference equal to 0, a maximum voltage output for phase difference equal to and it follows a linear law for phase differences in-between. Half of the maximum voltage is obtained for a phase difference equal to /2. Information on amplitude of the output AC voltage is received from an amplitude detector circuitry 510 similarly to the first embodiment. The amplitude information may be provided in form of a DC-voltage. A preset value of amplitude, which corresponds to a desired value of the amplitude of the mechanical resonator 100 motion in steady state oscillation is provided to the non-inverting input of a second amplifier A2 602 through its non-inverting input, and the detected amplitude value is received at the inverting input of the second amplifier A2 602. Output of the second amplifier A2 602 is limited to voltages that correspond to a phase difference between 0 and /2, or preferably to even smaller range of voltages corresponding to a phase difference between 0 and /4, since in this phase difference range the desired stabilization of the capacitive coupling may be achieved with high precision. The output of the second amplifier A2 602 is fed to the non-inverting input of the first amplifier A1 601. Amplifications and frequency responses of the first (A1 601) and second (A1 602) amplifiers are selected such that the control system, i.e. the negative feed-back loop, is stable and has fast enough response for being able to cancel the amplitude variations of the mechanical vibration. The first amplifier A1 601 and the second amplifier A2 602 may be considered as an exemplary discrete circuitry implementation of the controller circuitry 500 of FIG. 15.
(37) For operation with a frequency difference f above the resonant frequency f.sub.0, the polarity of amplifier A2 602 may be changed so that the output of the amplitude detector is fed to the non-inverting input and the preset amplitude level is fed into the inverting input. Further, the output of the second amplifier A2 602 shall be limited to values corresponding to phase shift range between /2 and , or preferably to a narrower range between 3/4 and for good amplitude and capacitive coupling stabilization results.
(38) FIG. 17 shows a third exemplary embodiment block diagram of a microelectromechanical sensor device with a MEMS part and a drive loop circuitry implementing the above disclosed inventive principles. The input AC-voltage from a voltage controlled oscillator VCO 505 is fed to the input transducer 110 of the mechanical resonator 100. Oscillation of the mechanical resonator 100 is detected with an output transducer 120, which provides at its output the output AC voltage corresponding to the resonator movement. The frequency of the VCO 505 is adjusted by a PLL controller 700. The PLL controller 700 receives at its input a phase difference information between the input AC-voltage and a phase shifted output AC voltage from the output transducer. The phase of the phase shifted output AC voltage is adjusted with a phase shifter circuitry 710. The phase shift is made adjustable over range from 0 to /2, or preferably over range from 0 to /4, if an operation frequency is selected that is below the resonant frequency f.sub.0, in other words if the frequency difference f has a negative value. The phase shift is made adjustable over range from /2 to , or preferably over range from 3/4 to , if an operation frequency is selected that is above the resonant frequency f.sub.0, in other words if the frequency difference f has a positive value. Amplitude of the resonator 100 motion is controlled by an amplitude controller 701, that adjusts the amount of phase shift in the phase shifter 710 based on detected difference between the detected amplitude of the output AC voltage and an intended preset value of amplitude. The preset value of amplitude, which corresponds to a desired value of the amplitude of the mechanical resonator 100 motion in steady state oscillation is provided to the amplitude controller 701 through a control input.
(39) FIG. 18 shows a fourth exemplary embodiment block diagram of a microelectromechanical sensor device with a MEMS part and a drive loop circuitry implementing the above disclosed inventive principles. A self-sustaining closed loop oscillator is provided with an amplifier A 800 and an adjustable phase shift by a phase shifter 515 in the feed-back part of the drive loop circuitry. The amplifier A 800 generates the input AC voltage to the input transducer 110, which amplitude of the input AC voltage will under steady state conditions approach the maximum value limited by the supply voltage of the amplifier or by any other known means of limiting an AC voltage to a maximum value. Thus, the amplifier A 800 is an exemplary embodiment of a drive circuit configured to generate the essentially constant amplitude input AC voltage. Amplitude is controlled by an amplitude controller 701 that adjusts the phase shift. Preset desired value of the amplitude is provided to the amplitude controller 701 through a control input. The amplitude controller 701 compares the detected amplitude value provided by an amplitude detector 501 to the desired preset value of amplitude, and controls the phase shifter 515 accordingly. This type of oscillator will start from the noise of the output amplifier 800, or from an artificially generated wide-band noise, which noise the mechanical resonator 100 will filter so that the output AC-voltage will contain a sinusoidal component. This component will be at the resonant frequency f.sub.0 first when the oscillation amplitude starts to increase, but the control loop will adjust the phase shift generated by the phase shifter 515 and thus the frequency so that in steady state the oscillation shifts to a frequency different from the resonant frequency. Steady state frequency of the mechanical resonator 100 will in this embodiment equal a frequency at which sum of the phase difference between the input AC voltage and the output AC voltage and the phase shift by the phase shifter 515 equals n*2, wherein n is an integer.
(40) In case the MEMS device of FIGS. 15 to 18 was a MEMS gyroscope, the resonator represents the primary resonator, the input transducer 110 may comprise the primary excitation transducer configured to provide an excitation force to the primary resonator of the gyroscope and the output transducer 120 may comprise the primary output transducer configured to provide electrical information corresponding to mechanical oscillation of the primary mechanical resonator of the gyroscope. In case the MEMS device of FIGS. 15 to 18 was an induction type magnetometer, the resonator represents the coil, the input transducer 110 may comprise the excitation transducer configured to provide an excitation force to resonate the coil of the magnetometer and the output transducer 120 may be configured to provide electrical information corresponding to mechanical oscillation of the coil of the magnetometer. In FIGS. 15 to 18, only those blocks of the device are shown that are needed for describing the invention. In addition to the presented blocks, the MEMS device may comprise further blocks and functions. For example, a gyroscope may comprise a secondary resonator and a secondary output transducer as shown for example in FIG. 4.
(41) It is apparent to a person skilled in the art that as technology advanced, the basic idea of the invention can be implemented in various ways. The invention and its embodiments are therefore not restricted to the above examples, but they may vary within the scope of the claims.
