Vehicle on-board charger for bi-directional charging of low/high voltage batteries

Abstract

Compact light-weight on-board three-port power electronic system built in various configurations of triple-active-bridge-derived topologies, including modular implementations, with control strategies capable of bi-directional power transfer among the three ports of the power electronic system, including simultaneous charging of a high voltage (HV) battery and a low voltage (LV) battery from a single phase power grid or a three-phase power grid with minimized reactive power and active circulating current, with ensured soft-switching for MOSFET devices, and with enhanced synchronous rectification and reduced power losses.

Claims

1. An on-board charger system for plug-in electric vehicles (PEVs), the on-board charger system comprising: an on-board charger (OBC) having an input port operatively coupled to an alternative current (AC) power grid, a first output port operatively coupled to a first battery, and a second output port operatively coupled to a second battery, wherein said OBC is configured for substantially simultaneous bi-directional power transfer between at least two of said input port, said first output port, and said second output port, respectively, wherein said OBC includes: a transformer sub-system integrated therewith and operatively coupled, by a primary side thereof to said input port, and by a secondary side thereof, to said first output port and said second output port, and a DC/DC converter integrated in said OBC and operatively coupled to said transformer sub-system, wherein said DC/DC converter includes: a first converter sub-system operatively coupled between said input port and said primary side of said transformer sub-system, a second converter sub-system operatively coupled between said secondary side of said transformer sub-system and said first output port, and a third converter sub-system operatively coupled between said secondary side of said transformer sub-system and said second output port; and a power transfer control sub-system integrated with said OBC and operatively coupled to said input port and said first output port and said second output port for bi-directional regulated power transfer therebetween while maintaining predetermined voltage levels, said power transfer control sub-system being configured for a combined phase shift and duty ratio control at said input port and said first output port and said second output port in a resonant-based mode or a pulse-width-modulation based mode of operation.

2. The on-board charger system of claim 1, wherein said first converter sub-system, said second converter sub-system, and said third converter sub-system of said DC/DC converter, and said transformer sub-system form a triple active bridge (TAB) converter sub-system.

3. The on-board charger system of claim 1, wherein said OBC has a configuration selected from a group including a resonant-based configuration and a pulse-width modulation (PWM)-based configuration, wherein each of said first converter sub-system and said second converter sub-system includes at least two MOSFET devices interconnected to form a half-bridge DC/DC converter circuit, or at least four MOSFET devices interconnected to form a full bridge DC-DC converter circuit, and wherein said third converter sub-system includes MOSFET devices interconnected to form a half-bridge DC/DC converter circuit, a full-bridge DC/DC converter circuit, or an active current doubler rectifier circuit.

4. The on-board charger system of claim 3, wherein in said resonance-based configuration, when each of said first converter sub-system and said second converter sub-system includes a half-bridge configuration, said OBC includes a resonant capacitor C.sub.1 connected between said first converter sub-system and said primary side of said transformer sub-system, and a resonant capacitor C.sub.2 connected between said secondary side of said transformer sub-system and said second converter sub-system.

5. The on-board charger of claim 3, wherein, in said resonance-based configuration, said OBC further includes resonance capacitors C.sub.1/2 and C.sub.2/2 included in a half-bridge configuration of each of said first converter sub-system and said second converter sub-system, said resonant capacitors C.sub.1/2 and C.sub.2/2 being coupled to said at least two MOSFET devices, and wherein in said PWM-based configuration, said DC/DC converter further includes a first shim inductor L.sub.1 and a first DC-blocking capacitor C.sub.1, each coupled between said first converter sub-system and said primary side of said transformer sub-system, a second shim inductor L.sub.2, and a second DC-blocking capacitor C.sub.2, each coupled between said secondary side of said transformer sub-system and said second converter sub-system, and a third inductor L.sub.3 interconnected between said secondary side of said transformer sub-system and said third converter sub-system.

6. The on-board charger system of claim 3, further comprising a resonant inductor L.sub.1 interconnected between said first converter sub-system and said primary end of said transformer sub-system, and a resonant inductor L.sub.2 interconnected between said secondary side of said transformer sub-system and said second converter sub-system, said L.sub.1 and L.sub.2 enhancing power density of said DC/DC converter.

7. The on-board charger of claim 3, further including a switching sub-system coupled between said secondary side of said transformer sub-system and said third converter sub-system, said switching sub-system controlling a power flow during a grid-to-vehicle power transfer or a vehicle-to-grid power transfer.

8. The on-board charger of claim 7, further including a power interruption device coupled between said input port and said first converter sub-system, between said first converter sub-system and said primary side of said transformer sub-system, and at said primary side of said transformer sub-system.

9. The on-board charger system of claim 3, wherein said power transfer control sub-system operates in a mode of operation selected from a group consisting of: a simultaneous charging mode of operation, a reactive power flow optimization mode of operation, and a synchronous rectification mode of operation; wherein, in said simultaneous charging mode of operation, said first converter sub-system, said second converter sub-system, and said third converter sub-system are configured as full-bridge DC/DC converter circuits, wherein a power flow towards the first and second output ports and the output voltage levels V.sub.1, V.sub.2, V.sub.3 at said input port, and said first and second output ports, respectively, are controlled by independent control variables including duty rations δ.sub.1, δ.sub.2, δ.sub.3 of said full-bridge DC/DC converter circuits of said first, second and third converter sub-systems and phase angle differences φ.sub.1 and φ.sub.2, between fundamental voltage waveforms of said first and second and said first and third converter sub-systems, respectively, wherein said first converter sub-system includes Q.sub.1, Q.sub.2, Q.sub.3, Q.sub.4 MOSFET devices, said second converter sub-system includes Q.sub.5, Q.sub.6, Q.sub.7, Q.sub.8 MOSFET devices, and said third converter sub-system includes Q.sub.9, Q.sub.10, Q.sub.11, Q.sub.12 MOSFET devices, and wherein said power transfer control sub-system is a three-loop control structure independently controlling said control variables φ.sub.1, φ.sub.2, δ.sub.1, δ.sub.2, δ.sub.3 to generate gate pulses applied to said MOSFET devices in accordance with: Q.sub.1: delay=0; Q.sub.3: delay=π2 δ.sub.1; Q.sub.5: delay=φ.sub.2+δ.sub.2−δ.sub.1; Q.sub.7: delay=π+φ.sub.2−δ.sub.2−δ.sub.1; Q9: delay=φ.sub.3+δ.sub.3−δ.sub.1; Q11: delay=π+φ.sub.3−δ.sub.3−δ.sub.1, and wherein MOSFET devices Q.sub.2, Q.sub.4, Q.sub.6, and Q.sub.8 are driven with gate pulses complimentary to gate pulses applied to said MOSFET devices Q.sub.1, Q.sub.3, Q.sub.5, and Q.sub.7, respectively.

10. The on-board charger system of claim 9, wherein, in said reactive power flow optimization mode of operation, said power transfer control sub-system generates said variables φ.sub.2, φ.sub.3, φ.sub.3, sets δ.sub.3=0, and determines δ.sub.1 for ensuring a soft switching condition for said MOSFET devices.

11. The on-board charger of claim 10, wherein, in said synchronous rectifier mode of operation, said power transfer control sub-system includes: a PWM generator supplying a PWMA control signal to said first converter sub-system, a generalized harmonic computational sub-system computing a phase angle difference value θ.sub.SRbetween the gate pulses for said first converter sub-system and said second converter sub-system required for synchronizing with a zero current crossing of a resonant current in said second converter sub-system, a delay computational sub-system coupled to said generalized harmonic computational sub-system receiving the PWMA control signal and said phase angle difference value θ.sub.SR therefrom, and computing a gate signal PWMB based on said PWMA and θ.sub.SR, said delay computational sub-system supplying said gate signal PWMB to said second converter sub-system to attain a synchronous rectification in said first converter sub-system and said second converter sub-system, and wherein a synchronous rectification is incorporated in said MOSFET devices Q.sub.5, Q.sub.6, Q.sub.7, and Q.sub.8 of said second converter sub-system during a grid-to-vehicle mode of operation, and in said MOSFET devices Q.sub.1, Q.sub.2, Q.sub.3, and Q.sub.4 of said first converter sub-system during a vehicle-to-grid mode of operation.

12. A three-phase on-board charger (OBC) system for plug-in electric vehicles (PEVs), the three-phase OBC system comprising: a modular on-board charger (OBC) having an input port coupled to a three-phase alternative current (AC) power grid and a first output port and a second output port coupled to an on-board high voltage (HV) battery and an on-board low voltage (LV) battery, respectively, said modular OBC including a plurality of interconnected electronic modules and has a configuration selected from a group consisting of: a one-stage configuration and a two-stage configuration; wherein in said two-stage configuration, said modular OBC includes: a first electronic module operatively coupled between said input port and a DC link, said first electronic module including a bi-directional three-phase Power Factor Correction (PFC) rectifier sub-system, a second electronic module operatively coupled to a capacitor C.sub.DC of said DC link, said second electronic module including a plurality of MOSFET devices interconnected in a bridge sub-system, a third electronic module operatively coupled to said bridge sub-system, said third electronic module including an integrated transformer sub-system, a fourth electronic module operatively coupled between said integrated transformer sub-system and said first output port, wherein said fourth electronic module includes a rectifier sub-system for said on-board HV battery, said on-board HV battery being coupled to said fourth electronic module, a fifth electronic module operatively coupled between said integrated transformer sub-system and said second output port, wherein said fifth electronic module includes a rectifier sub-system for said on-board LV battery, said on-board LV battery being coupled to said fifth electronic module, and a control sub-system operatively coupled to said plurality of integrated electronic modules to implement a combined phase shift and duty ration based power flow control for charging said on-board HV battery and said on-board LV battery from the three-phase power AC grid, the on-board HV battery to the on-board LV battery charging, charging of the PEVs from the AC power grid, and discharging of the PEVs to the three-phase AC power grid; and wherein in said one-stage configuration, said modular OBC includes: a sixth electronic module operatively coupled to said input port, said sixth electronic module including a single-stage AC/DC converter sub-system, and said third electronic module operatively coupled to said sixth electronic module, said fourth electronic module and said fifth electronic module operatively coupled to said integrated transformer sub-system in said third electronic module, with said on-board HV battery and said on-board LV battery coupled to said first output port and said second output port, respectively, and said control sub-system operatively coupled to said plurality of integrated electronic modules.

13. The three-phase OBC system of claim 12, wherein, in said two-stage configuration, said bi-directional three-phase PFC rectifier sub-system of said first electronic module is selected from a group consisting of: three phase boost PFC rectifier, three-phase buck PFC rectifier, three-phase Vienna-type PFC rectifier, three single-phase buck PFC rectifier, and a modular multi-level converter, said bridge sub-system in said second electronic module is selected from a group consisting of: two full bridges connected in parallel, two half bridges connected in parallel, two individual half bridges, two individual full bridges, a three-phase bridge, three full bridges, and three half bridges; said transformer sub-system in said third electronic module is selected from a group consisting of: at least two separate transformers, at least two transformer sets with primary windings delta interconnected with capacitors and secondary windings interconnected in series with capacitors, and a single three-phase transformer; said rectifier sub-system of each of said fourth and fifth electronic modules, respectively, is selected, depending on the configuration of the transformer sub-system from a group consisting of: at least two half bridges, at least two full bridges, at least two half bridges connected in parallel, at least two full bridges connected in parallel; and wherein said DC Link is selected from a group consisting of: split DC Link, and non-split DC Link.

14. The three-phase OBC system of claim 12, wherein in said one-stage configuration, said AC/DC converter of said sixth electronic module is selected from a group consisting of: three single-phase AC-DC converters connected in parallel, a Matrix-based three-phase AC-DC Triple Active Bridge (TAB) converter, and a SWISS-based TAB converter with split AC capacitors; wherein said transformer sub-system in said third electronic module is selected from a group consisting of: at least two separate transformers, three transformers with primary windings delta-interconnected with capacitors and secondary windings serially connected with capacitors, and a single three-phase transformer; and wherein said rectifier sub-system in said fourth and fifth electronic modules, respectively, is selected from a group consisting of: at least two half bridges, at least two full bridges, at least two half bridges connected in parallel, and at least two full bridges connected in parallel.

15. The there-phase OBC system of claim 12, wherein said control sub-system executes a pulse frequency modulation (PFM) or pulse width modulation (PWM).

16. The three-phase OBC system of claim 12, having a splitted power flow, wherein said two-stage configuration includes: a bi-directional three-phase PFC rectifier in said first electronic module, two triple active full bridge converters connected in parallel in said second electronic module integrated with two transformers and six shim inductors of said third electronic module.

17. The three-phase OBC system of claim 12, having a splitted power flow, further including a three-phase boost PFC rectifier in said first electronic module, and two single-phase DC/DC converters in each of said fourth electronic module and said fifth electronic module, respectively, each single-phase DC/DC converter being integrated with a respective transformer sub-system in each split power flow.

18. The three-phase OBC system of claim 17, wherein said capacitor C.sub.DC of said DC link is split in two sub-capacitors.

19. The three-phase OBC system of claim 12, further including: a three-phase boost PFC rectifier in said first electronic module, and an integrated three-phase DC-DC converter in each of said fourth electronic module and said fifth electronic module.

20. The three-phase OBC system of claim 12, further including three single-phase H-bridge PFC rectifiers in said first electronic module integrated with triple active bridge (TAB) converters coupled in parallel.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) FIG. 1 is a schematic diagram of a prior art integrated configuration of the on-board reconfigurable battery charger for electric vehicles;

(2) FIG. 2 is a schematic diagram of the prior art charger with the auxiliary power module connected to the battery at the DC side;

(3) FIG. 3 is a schematic diagram of a prior art charger for PEV hybrid energy management system which integrates the auxiliary power module at the DC side;

(4) FIG. 4 is a schematic diagram of a prior art three-winding transformer-based integrated OBC with a half-bridge CLLC resonant converter and LLC resonant converter;

(5) FIG. 5 depicts a schematic of a prior art 3-winding transformer-based integrated onboard charger topology with a dual active bridge converter and a passive current doubler;

(6) FIG. 6 is a prior art integrated configuration of the on-board charger where a propulsion is used as a coupled DC inductor for the charger;

(7) FIG. 7 depicts a prior art configuration in which the split-winding AC motor structure is utilized for integration;

(8) FIG. 8 is a prior art multi-port integrated DC/DC converter with a coupled transformer structure using additional windings;

(9) FIG. 9 is a prior art multi-port converter with a stationary storage port which utilizes the star connection in the secondary of transformers to eliminate third-order harmonics ripples;

(10) FIG. 10 is a schematic diagram of the subject on-board charger (OBC) for an electric vehicle (EV);

(11) FIG. 11 is a block diagram for the subject resonant based topology of the on-board charger for an electric vehicle;

(12) FIG. 12 is a block diagram for pulse widths modulation-based topology for the subject on-board charger for an electric vehicle;

(13) FIG. 13 is a block diagram of the subject on-board charger for electric vehicles which uses dual active bridge converter combined with a tertiary active current doubler rectifier;

(14) FIG. 14 is a block diagram of the subject on-board charger for electric vehicles using a dash tertiary side back-to-back (B2B) switch;

(15) FIG. 15 is a block diagram of the subject on-board charger for electric vehicles with a primary side back-to-back (B2B) switch/relay installation;

(16) FIG. 16 is representative of the integrated DC/DC conversion stage of the subject on-board charger for an electric vehicle;

(17) FIG. 17 is representative of the Delta-equivalent of the star connected network shown in FIG. 16;

(18) FIG. 18 is a diagram representative of the three-level bridge voltage waveforms on the primary, secondary, and tertiary sides of the subject OBC;

(19) FIG. 19 is the block diagram of the subject control methodology with three-loop structure for controlling three reference variable for the Triple Active Bridge (TAB) converter;

(20) FIG. 20 is a schematic diagram representative of the 3-port network that can be modelled as 3-coupled transmission lines;

(21) FIG. 21 is a block diagram of the control architecture of the triple active bridge (TAB) converter with reactive power minimization;

(22) FIG. 22 is an equivalent circuit of a resonant DC/DC converter for the subject on-board charger for electric vehicles;

(23) FIG. 23 is a diagram of a generalized harmonic model;

(24) FIG. 24 is a block diagram representative of the subject control system using both primary side control and synchronous rectification for resonant-based topology;

(25) FIG. 25 is a block diagram of an alternative synchronous rectification structure for rectification control in the subject on-board charger;

(26) FIG. 26 is a block diagram of a subject modular two-stage on-board charger for simultaneous charging by means of various bridge and transformer configurations;

(27) FIG. 27 is a block diagram for a subject modular single-stage converter;

(28) FIG. 28 is a block diagram of the subject bi-directional OBC having three-phase integrated PFC rectifier and capable of simultaneous charging both HV and LV batteries from the grid;

(29) FIG. 29 is representative of the subject topology using three-phase boost PFC rectifier and integrated with the TAB having input parallel-output parallel;

(30) FIG. 30 is representative of the subject alternative topology with the three-phase boost PFC rectifier integrated with a TAB having input split-output parallel;

(31) FIG. 31 is representative of the subject alternative topology with a Vienna-based TAB converter with split DC link;

(32) FIG. 32 is representative of the subject alternative topology with the three-phase boost PFC rectifier integrated with three-phase CLLC;

(33) FIG. 33 is representative of the subject alternative topology with the three-phase boost PFC rectifier integrated with a 3-phase TAB;

(34) FIG. 34 is representative of the subject alternative topology including three single-phase H-bridge PFC rectifiers with TAB converters connected in parallel;

(35) FIG. 35 is representative of the subject alternative topology including three single-phase AC-DC TAB converters connected in parallel;

(36) FIG. 36 is representative of the subject alternative topology including a matrix-based three-phase AC-DC TAB converter;

(37) FIG. 37 is representative of the subject alternative topology with the Swiss-based TAB converter with split capacitors;

(38) FIG. 38 is representative of the subject alternative topology having MMC-based system with the TAB having input parallel-output parallel;

(39) FIGS. 39A and 39B are representative of the sub-module configurations in the subject MMC-based system with half bridge and full bridge, respectively;

(40) FIGS. 40A and 40B are representative of the configurations of the windings using the subject MMC based system with a single star and single delta connections, respectively;

(41) FIG. 41 is representative of the subject control loop configuration implemented with both phase shift-and-duty ratio control;

(42) FIG. 42 is representative of the delta equivalent configuration of the TAB-based converter;

(43) FIG. 43 is block-diagram of the subject control system adapted for the subject OBC;

(44) FIG. 44 is a diagram representative of the comparison of operation range between the phase shift φ control and the duty ratio δ control;

(45) FIG. 45 is a diagram representative of the State-of-Charge SOC map; and

(46) FIG. 46 is a diagram representative of the SOC mapping of the HV and LV batteries in the time domain.

DETAILED DESCRIPTION

(47) Referring to FIGS. 10-38 the subject on-board charger system 10, includes an on-board charger (OBC) 12, also referred to herein as an on-board charging DC/DC converter, and an integrated transformer sub-system 14. The OBC 12 also includes a bi-directional PFC (power factor correction) rectifier 16 through which the on-board charger system 10 is connected to the power grid 18.

(48) The PFC rectifier 16 is included in an alternative current (AC) side 17 of the OBC system 10, while the integrated transformer 14 and the DC/DC converter 12, are included in a direct current (DC) side 19 of the OBC system 10. The AC side 17 and the DC side 19 are interconnected through a DC link 21 having a DC link capacitor C.sub.DC.

(49) The transformer sub-system 14 includes, in the implementation shown in FIG. 10, a primary winding 23, a secondary winding 25, and a secondary winding 27 (also referred to herein as a tertiary winding) at the tertiary side 36 of the OBC (DC/DC converter) 12.

(50) In the topology shown in FIG. 10, the DC side 19 is configured with MOSFET devices interconnected to form three H-bridges sub-systems 20, 22, 24, three shim inductors (also acting as power transfer inductors) L.sub.1, L.sub.2 and L.sub.3, and two DC-blocking capacitors C.sub.1 and C.sub.2, which, in combination, form a triple active bridge (TAB) structure 29.

(51) Topology wise, the inductors L.sub.1, L.sub.2 and L.sub.3 provide sufficient inductive energy to achieve zero-voltage-switching for MOSFETs Q.sub.1-Q.sub.4 (in the bridge sub-system 22), Q.sub.5-Q.sub.8 (in the bridge sub-system 20), and Q.sub.9-Q.sub.11 (in the bridge sub-system 24), respectively. The inductors L.sub.1, L.sub.2, and L.sub.3 can be achieved by the leakage inductance of the transformer sub-system 14 to further enhance the power density of the DC/DC converter 12.

(52) The capacitors C.sub.1 and C.sub.2 serve the purpose of blocking the DC component in the current in different operating modes. The DC component of current arises from different mechanisms, such as: (a) asymmetrical pulse width modulation, (b) parasitic imbalance of the devices, and (c) start-up load transients of the DC/DC converter 12.

(53) A unique control methodology is implemented with the topology shown in FIG. 10, which will be detailed in the following paragraphs. With the innovative and unique control, the topology of the subject OBC system 10 shown in FIG. 10 is able to achieve the power flow in different directions. In fact, in addition to the grid-to-vehicle (G2V), vehicle-to-grid (V2G) and HV-to-LV operations (H2L), the system shown in FIG. 10 can provide a unique grid to both HV and LV batteries operation, i.e. is capable of simultaneously charging both HV and LV batteries, with minimized reactive power and circulating current.

(54) There are various topological alternative implementations of the subject system are contemplated which are applicable in simultaneous charging of the HV battery 26 and LV battery 28, as well as the bi-directional power flow in a three-port network system with the subject power transfer control strategy, detailed in further paragraphs.

(55) There are primarily two categories of alternative embodiments of the subject system 10, including: a) Resonant-based topologies, and b) Pulse width modulation (PWM)-based topologies.

(56) Resonant-based topologies are depicted in FIG. 11, where the charger sub-system (Block 1) 30 and the charger sub-system (Block 2) 32 have either a half bridge or a full bridge configuration. The charger sub-system (Block 3) 34 includes MOSFET devices and/or capacitors, as well as inductances, interconnected to form either a half bridge or a full bridge configuration. The charger sub-system (Block 3) 34 can be a half bridge, a full bridge, or an active current doubler rectifier.

(57) In the circuitry shown in FIG. 11, C.sub.1 and C.sub.2 are the resonant capacitors. When the half bridge configuration is selected for either of the charger sub-systems 30 or 32, there are two options, when the C.sub.1 and C.sub.2 serve as resonant capacitors, or when C.sub.1 and C.sub.2 can be removed from the circuit, while the capacitors 31, 33 in the half bridge 35 are selected to be resonant capacitors with the capacitance of C.sub.1/2 or C.sub.2/2.

(58) The resonant inductors L.sub.1 and L.sub.2 can be realized by the leakage inductances from the transformer sub-system 14 to further enhance the power density of the DC/DC converter 12.

(59) The resonant based topological variations are summarized in Table. I. It is noted that not every variation is capable of grid-to-both batteries (G2B) operation, which will be detailed in further paragraphs.

(60) TABLE-US-00001 TABLE I Summary of resonant based topological variations to the invention Block 1 Block 2 Block 3 Resulted topology G2B capability 1 Full bridge Full bridge Full bridge Variation 1 Yes 2 Full bridge Full bridge Active current doubler Variation 2 Yes 3 Full bridge Full bridge Half bridge Variation 3 No 4 Full bridge Half bridge Full bridge Variation 4 No 5 Full bridge Half bridge Half bridge Variation 5 No 6 Full bridge Half bridge Active current doubler Variation 6 No 7 Half bridge Full bridge Full bridge Variation 7 No 8 Half bridge Full bridge Half bridge Variation 8 No 9 Half bridge Full bridge Active current doubler Variation 9 No 10 Half bridge Half bridge Full bridge Variation 10 No 11 Half bridge Half bridge Half bridge Variation 11 No 12 Half bridge Half bridge Active current doubler Variation 12 No

(61) Pulse width modulation based topological alternative implementations are presented in FIG. 12. Compared with the configuration shown in FIG. 11, the topology presented in FIG. 12 has an additional inductor L.sub.3 at the tertiary side 36.

(62) The capacitors C.sub.1 and C.sub.2 function as the DC-blocking capacitors, inductances L.sub.1, L.sub.2 and L.sub.3 are shim inductors functioning to provide the inductive energy for the zero-voltage-switching of the MOSFET devices in the charger sub-systems 30, 32, 34, also referred to herein as Block 1, Block, Block 3, respectively. Similarly, the inductors L.sub.1, L.sub.2 and L.sub.3 can be realized by the leakage inductances of the transformer sub-system 14 to further improve the power density of the DC/DC converter 12.

(63) In the circuit configuration shown in FIG. 12, Block 2 (charger sub-system 32) can be either a half bridge or a full bridge. The block 3 (sub-system 34) can be a half bridge, a full bridge, or an active current doubler rectifier. Different combinations of the alternative implementations contemplated in the subject system result in different topologies.

(64) For instance, a triple-active-bridge (TAB) converter, shown as FIG. 10 can be implemented by selecting a full bridge for all the Blocks 1, 2, 3 (charger sub-systems 30, 32 and 34).

(65) In the pulse width modulation-based implementation, shown in FIG. 13, one of the implementations uses a dual active bridge converter combined with the tertiary active current doubler rectifier 38 which operates to control the power flow and to enable the power flow management with multiple directions, i.e. grid-to-vehicle (G2V), vehicle-to-grid (V2G), HV-to-LV (H2L), and grid-to-Both (G2LH) batteries operations.

(66) The pulse width modulation-based implementations of the subject system 10 contemplated in the subject system 10, are summarized as Table II. Not every alternative embodiment is capable of grid-to-both batteries operation, as will be detailed in further paragraphs.

(67) TABLE-US-00002 TABLE II Summary of pulse width modulation based topological embodiments of the subject system Block 1 Block 2 Block 3 Resulted topology G2B capability 1 Full bridge Full bridge Full bridge Subject System (FIG. 10) Yes 2 Full bridge Full bridge Active current doubler Variation 1 (FIG. 13) Yes 3 Full bridge Full bridge Half bridge Variation 2 No 4 Full bridge Half bridge Full bridge Variation 3 No 5 Full bridge Half bridge Half bridge Variation 4 No 6 Full bridge Half bridge Active current doubler Variation 5 No 7 Half bridge Full bridge Full bridge Variation 6 No 8 Half bridge Full bridge Half bridge Variation 7 No 9 Half bridge Full bridge Active current doubler Variation 8 No 10 Half bridge Half bridge Full bridge Variation 9 No 11 Half bridge Half bridge Half bridge Variation 10 No 12 Half bridge Half bridge Active current doubler Variation 11 No

(68) The alternative topologies contemplated in the subject OBC system 10 also may include back-to-back (B2B) switches/relays 39. As shown in FIG. 14, a back-to-back switches (or relay) 39 are needed at the tertiary side 36 of the DC/DC converter 12 to cut the power flow during the grid-to-vehicle (G2V) charging or vehicle-to-grid (V2G) discharging operations. Otherwise, when the voltage of the LV battery 28 is lower than the voltage of the HV battery 26, the power flow towards LV battery 28 cannot be zero even with all tertiary side MOSFETs (in Block 3) turned off. Moreover, during HV-to-LV (H2L) power flow operation, the reflected voltage on the primary winding 40 of the transformer sub-system 14 may have an excessive peak value, which may cause the failure of the voltage on the DC link 21 and on the MOSFETs in the Block 1 on the primary side 42. To prevent such failure, a relay (or back-to-back switch) 44 can be implemented at points A, B or C, as shown in FIG. 15.

(69) During the HV-to-LV (H2L) power flow, the back-to-back switch/relay(s) 44 can be turned off to break the electrical connections between the active part (DC/DC converter 12) of the circuit and the primary side circuit (Block 1). It is noted that for both topologies shown in FIGS. 14-15, the inductor L.sub.3 at the tertiary side 36 is included in the circuitry only for the pulse width modulation-based embodiments.

(70) Power Flow Control Methodology for Single-Phase and Three-Phase Integrated OBC and APM

(71) (a) Simultaneous Charging

(72) The control system 45 implemented in the subject charger system 10 is capable of charging two energy storage sources (HV battery, LV battery) 26, 28 from another source, which may be the power system AC grid 18. In addition, the control system 45 has the flexibility for grid-to-vehicle (G2V), vehicle-to-grid (V2G), and HV-to-LV (H2L) operations. Charging can be performed within a wide range of operating conditions of 85 V-265 V single-phase AC, 100 V-1000 V for HV battery, and 8 V-28 V for LV battery.

(73) The integrated DC/DC conversion stage of the subject converter architecture, as shown in FIG. 16, is a three-port network including one input port 60 and two output ports 62, 64. The primary, secondary and tertiary bridge voltages V.sub.pri, V.sub.sec, V.sub.ter, respectively, are bipolar quasi-square waves, respectively, with mutual phase shifts. Since the magnetizing inductance of the transformer sub-system 14 is much greater than the leakage inductance, it can be ignored in the equivalent circuit without any loss in generality.

(74) FIG. 17 represents the delta-equivalent of the star connected network shown in FIG. 16, which is a more convenient model for establishing power flow expressions. The equivalent delta inductances are expressed by

(75) $\begin{array}{cc}{L}_{12}=L_1+L_2+\frac{L_1{L}_2}{L_3};& \left(\mathrm{Eq}.1\right)\\ L_{13}=L_1+L_3+\frac{L_1{L}_3}{L_2};& \left(\mathrm{Eq}.2\right)\\ L_{23}=L_2+L_3+\frac{L_2{L}_3}{L_1}.& \left(\mathrm{Eq}.3\right)\end{array}$

(76) In the model shown in FIGS. 16-17, the voltages V.sub.1, V.sub.2, V.sub.3 (V.sub.pri, V.sub.sec, V.sub.ter) are the reflected amplitudes of the input DC voltage to the DC/DC stage, the HV battery, and the LV battery on the primary side of the transformer, respectively.

(77) The dual-output triple active bridge (TAB) cannot be treated as two independent dual active bridge converters due to the unavoidable power flow between Port-2 and Port-3, shown in FIG. 17.

(78) In order to control the power flow towards the two output ports 62, 64, as well as to regulate the output voltage levels, the subject control system 45 uses up to five independent control variables, i.e. three duty ratios (δ.sub.1, δ.sub.2, δ.sub.3) of three full-bridges (primary, secondary and tertiary) incorporated in Blocks 1, 2, 3 shown in FIG. 15, and the phase angle differences φ.sub.2 and φ.sub.3 between the primary-secondary and primary-tertiary fundamental bridge voltage waveforms, respectively, as shown in FIG. 17. Based on the subject switching scheme, three bridge voltage outputs are defined as follows:

(79) The power transfer expression from port-x to port-y (where, x, y could be 1, 2, 3) is as follows:

(80) $\begin{array}{cc}{P}_{\mathrm{xy}}=\frac{4V_x{V}_y\sin \left({\phi }_y-{\phi }_x\right)\cos \left({\delta }_y\right)\cos \left({\delta }_x\right)}{{\pi }^3{f}_s{L}_{\mathrm{xy}}}& \left(\mathrm{Eq}.4\right)\end{array}$

(81) In order to minimize the circulating active power for a given set of load power levels P.sub.2 and P.sub.3, the power transfer between any two ports (out of the three ports P.sub.12 P.sub.2, P.sub.3) is fixed, and thus must be regulated at their reference values as follows:

(82) $\begin{array}{cc}{P}_{12}=-\frac{2P_2+P_3}{3}& \left(\mathrm{Eq}.5\right)\\ P_{23}=\frac{P_2-P_3}{3}& \left(\mathrm{Eq}.6\right)\\ P_{31}=\frac{P_2+2P_3}{3}& \left(\mathrm{Eq}.7\right)\end{array}$

(83) Three-level bridge voltage waveforms on the primary, secondary and tertiary sides are shown in FIG. 18. From the desired values of the duty ratios δ and phase angle differences φ of the bridge voltages, all the gate pulses can be generated with respect to a reference pulse, presented as follows:
Q1:delay=0;Q3:delay=π−2δ.sub.1  (Eq. 8)
Q5:delay=φ.sub.2+δ.sub.2−δ.sub.1; Q7:delay=π+φ.sub.2−δ.sub.2−δ.sub.1  (Eq. 9)
Q9:delay=φ.sub.3+δ.sub.3−δ.sub.1; Q11:delay=π+φ.sub.3−δ.sub.3  (Eq. 10) where Q2, Q4, Q6 and Q8 (shown in FIGS. 10 and 13) are driven with complementary pulses to those of Q1, Q3, Q5 and Q7, respectively. The duty ratios of all the gate pulses are 50%.

(84) The design of conventional control loops as used in dual active bridge converters is not applicable to this problem due to the cross-couplings in power flow among the ports. Therefore, the subject control methodology with a three-loop structure (for controlling three reference variables), shown in FIG. 19 incorporating power flow decoupling, is employed to achieve the simultaneous charging. Both the power flow expressions of P.sub.12 and P.sub.32 involve φ.sub.2 and V.sub.2 as common variables. The net output power at Port-2 is a quadratic function of V.sub.2 and also is a linear combination of P.sub.12 and P.sub.32.

(85) The above mathematically implies that the phase shift angle φ.sub.2 can be used as a control variable for the Port-2 output voltage. Similarly, the phase shift angle φ.sub.3 can be used as a control variable for the Port-3 output voltage, as both P.sub.13 and P.sub.23 include φ.sub.3 and V.sub.3 as common terms.

(86) Furthermore, in order to account for the cross-coupled power flow in the control loop, the third reference variable shall be the active power flow reference P.sub.23 flowing between Port-2 and Port-3. The corresponding control variable could be either δ.sub.2 or δ.sub.3, as φ.sub.2 and φ.sub.3 are involved in controlling the output voltages.

(87) It is observed that φ.sub.3 has an important role in controlling the Port-2 output voltage (equivalently power P.sub.2), and so does φ.sub.2 in controlling Port-3 power, i.e. P.sub.3. Therefore, the control loops generating the phase angle differences cannot be totally decoupled, which is solved by providing cross-loop gains to the output of the voltage controllers. Assuming X.sub.1 and X.sub.2 to be the voltage controller outputs, sin(φ.sub.2) and sin(φ.sub.3) can be generated as follows:
sin(φ.sub.2)=X.sub.1+G.sub.2X.sub.2  (Eq. 11)
and
sin(φ.sub.3)=X.sub.2+G.sub.1X.sub.1  (Eq. 12)
After satisfying the hard constraints on active power transfer, there still remains the freedom in selection of the remaining two control variables, i.e. δ.sub.1 and δ.sub.3. If no other optimization constraints are applied, these two phase shift angles can be set to zero in order to transfer the maximum power.

(88) (b) Reactive Power Flow Optimization

(89) The subject control system 45 operates to minimize the net circulating reactive power inside the converter 12. As a part of the control variable generation algorithm, peak current stress minimization and reactive power optimization algorithms are incorporated without violating the three independent equality constraints on active power transfer towards two DC outputs. This is beneficial in selecting active semiconductors with less current rating, in minimizing the conduction losses leading to a higher conversion efficiency, as well as easier thermal management.

(90) As evident from the three hard constraints set on the active power transfer among the three ports (presented in previous paragraphs), three control variables are generated from the control loop structure and the remaining two variables have the freedom for optimization constraints. In order to minimize the circulating conduction losses, the main objective of the control approach is to make the net reactive power injected into the network to zero. The three-port network can be modeled as three coupled transmission lines, one of which is depicted in FIG. 20.

(91) The reactive power flow from Port-1 to Port-2 and in the reverse direction can be expressed as follows:

(92) $\begin{array}{cc}{Q}_{12}=\frac{V_1\cos \left({\delta }_1\right)\left[V_2\cos \left({\delta }_2\right)\sin \left({\phi }_2\right)-V_1\cos \left({\delta }_1\right)\right]}{2\pi f_s{L}_{12}}& \left(\mathrm{Eq}.13\right)\\ \mathrm{and}& \\ Q_{21}=\frac{-V_2\cos \left({\delta }_2\right)\left[V_1\cos \left({\delta }_1\right)\sin \left({\phi }_2\right)-V_2\cos \left({\delta }_2\right)\right]}{2\pi f_s{L}_{12}}& \left(\mathrm{Eq}.14\right)\end{array}$

(93) As follows from Eqs. 13-14, for an asymmetric network, the sum of two reactive powers is not equal to zero with no interdependence constraint being applied to δ.sub.1 and δ.sub.2, which essentially means that there will inevitably be a phase difference between any bridge voltage and switch current, which unavoidably increases the peak level of the device current.

(94) In case of a three-port network, there are six different paths for reactive power flow, which must sum to zero and hence, the following condition can be set:
Q.sub.12+Q.sub.13+Q.sub.23+Q.sub.21+Q.sub.32+Q.sub.31=0  (Eq. 15)

(95) Eq. 15 essentially results into a quadratic relation between cos(δ.sub.1) and cos(δ.sub.3) for a given set of φ.sub.2, φ.sub.3, δ.sub.2 generated from the control loop. Therefore, applying reactive power minimization constraints reduces the degrees of freedom for selection of duty ratios from two to one. If no additional constraints are imposed, δ.sub.3 can be set to zero for enabling maximum power transfer and δ.sub.1 can be determined.

(96) Furthermore, an additional constraint arises due to ensuring soft-switching at all the MOSFETs, which requires a minimum reactive power to be flowing from the input port. Assuming a minimum phase lag angle θ required between the fundamental of primary bridge voltage and primary winding current at a particular load power to ensure ZVS (Zero-Voltage-Switching), the minimum reactive power from Port-1 is as follows:
Q.sub.1=Q.sub.12+Q.sub.13>Q.sub.1,min=P.sub.1 tan(θ)  (Eq. 16)
where P.sub.1 denotes the active power input from port-1.

(97) θ can be calculated from the condition that instantaneous inductive energy during switching has to be greater than the energy stored in MOSFET output capacitors. Therefore, there exist four equality constraints and one inequality constraint (for soft-switching), based on which four control variables can be explicitly determined whereas the fifth variable will have a range of solution. The control system block diagram with constraints on reactive power and soft-switching is shown in FIG. 21.

(98) (c) Synchronous Rectification Operation

(99) An efficient bi-directional power flow can be achieved by incorporating synchronous rectification in the secondary MOSFETs (Q5-Q8, shown in FIG. 10) during the grid-to-vehicle (G2V) charging mode and in the primary MOSFETs (Q1-Q4, shown in FIG. 10) during the vehicle-to-grid (V2G) discharging mode of operation. The proposed method accurately predicts the zero crossing instant of the resonant currents based on comprehensive and generalized harmonics modeling of the entire converter. This method helps to achieve synchronous rectification operation without even requiring any high bandwidth secondary side current sensor.

(100) In case of any resonant based variations presented in the previous paragraphs, the secondary side of the converter requires rectification to convert the high frequency AC output of the transformer to DC. Conventionally, diodes are used on the secondary side, which can automatically detect the phase of current by switching at zero crossing of current. The use of diodes as secondary rectification devices leads to higher conduction loss due to on-state voltage drop of the diode and also hinders the bidirectional operation capability of circuit. In order to enable bidirectional power transfer capability and improve efficiency, diodes are replaced by switches, depicted in FIG. 11. The use of the secondary switches leads to the requirement of tracking of zero crossing of secondary side resonant current for synchronous rectification. During SR operation, phase shift has to be introduced between the primary gate signals and secondary gate signals synchronizing with the zero crossing of secondary side resonant current.

(101) In order to regulate the output voltage and control the power flow, primary side control using pulse frequency modulation (PFM) is conducted and the required phase angle difference between primary and secondary gate pulses is calculated using the proposed generalized harmonics modeling approach. The equivalent circuit of an isolated resonant DC/DC converter is present at FIG. 22. Since the secondary side 68 acts just as a rectifier, the gate pulses are generated based on the resonant current polarity. Therefore, the net phase difference between the primary and secondary gate pulses equals to the sum of phase angle of the input impedance and the phase angle difference between secondary i.sub.see and primary i.sub.pri resonant current at the secondary side 68 and primary side 42, respectively. Despite that a resonant circuit is typically analyzed using first harmonic approximation, the subject control scheme incorporates the effects of other higher order odd harmonics in net resonant current expression and thus, determines a more accurate zero crossing angle.

(102) Since the input to the AC equivalent model (shown in FIG. 22) of a resonant converter is a bipolar square wave, i.e., a set of input voltage sources of odd order frequencies connected in series to a passive network, superposition theorem is used to calculate the effect of each harmonic on output current. For each harmonics, the input impedance phase angle and the phase shift between primary and secondary resonant currents can be calculated. According to the diagram shown in FIG. 22, different phase angles and impedances are expressed as follows:

(103) $\begin{array}{cc}{Z}_{\mathrm{in}}\left(k\omega \right)=Z_1\left(k\omega \right)+Z_m\left(k\omega \right).Math.(Z_2\left(k\omega \right).Math.R_{\mathrm{eq}})& \left(\mathrm{Eq}.17\right)\\ \mathrm{where}{Z}_1\left(k\omega \right)=jk\omega L_{\mathrm{pri}}-\frac{j}{k\omega C_{\mathrm{pri}}}& \left(\mathrm{Eq}.18\right)\\ Z_m\left(k\omega \right)=jk\omega L_m& \left(\mathrm{Eq}.19\right)\\ Z_2\left(k\omega \right)=\mathrm{jk}\omega L_{\sec }^{\prime }-\frac{j}{k\omega C_{\sec }^{\prime }}m,& \left(\mathrm{Eq}.20\right)\end{array}$
where k is the order of the harmonics

(104) Furthermore, the phase difference between primary and secondary current is expressed as:

(105) $\begin{array}{cc}{\phi }_k=\mathrm{angle}\left(\frac{Z_m\left(k\omega \right)}{Z_m\left(k\omega \right)+Z_2\left(k\omega \right)+R_{\mathrm{eq}}}\right)& \left(\mathrm{Eq}.21\right)\end{array}$

(106) Therefore, the net current expression flowing on the secondary side is given as the sum of all odd harmonics current:

(107) $\begin{array}{cc}{i}_{\sec }=\underset{k=1}{\overset{\infty }{.Math.}}\frac{4V_{DC}}{k\pi .Math.Z_{tn}\left(k\omega \right).Math.}.Math.\frac{Z_m\left(k\omega \right)}{Z_m\left(k\omega \right)+Z_2\left(k\omega \right)+R_{\mathrm{eq}}}.Math.\sin \left(k\omega t-{\alpha }_k-{\phi }_k\right)& \left(\mathrm{Eq}.22\right)\end{array}$

(108) The secondary side 68 gate pulses (i.e. synchronous rectification) can be generated by detecting the zero crossing (online/offline) of the formulated resonant current for a given set of load resistance and switching frequency. The order of harmonics up to which this sum needs to be carried out possesses a trade-off between accuracy and computational complexity. The generalized harmonic model and the overall control block diagram are demonstrated in FIGS. 23-24, respectively.

(109) The implementation of the synchronous rectification control is not only restricted to digital domain, but also can be performed using analog solutions, as shown in FIG. 25, where the secondary side resonant current sensed by a high bandwidth current sensor can be compared with zero using a high speed comparator, and thus, the comparator output can be fed as gate signals to the secondary side semiconductors after maintaining electrical isolation (if necessary).

(110) Three-Phase Integrated OBC Capable of Simultaneous Charging Both HV and LV Batteries from the Grid.

(111) Various topologies are contemplated herein which constitute the subject on-board charging system 10 that are capable of realizing simultaneous charging by means of alternative triple active bridge and transformer configurations, including a single-stage configuration 100 and a two-stage configuration system 102 of the subject system 10, which can be controlled through pulse frequency modulation (PFM) or pulse width modulation (PWM).

(112) The modular two-stage configurations 102 are presented by a block diagram depicted in FIG. 26, which can be further categorized as single-phase and three-phase configurations. The charger modules, also referred to herein as Blocks 1, 2, 3, 4, 5, or charger sub-systems, which are depicted in FIG. 26, are configured with multiple converters interconnected in parallel to split the power flow, as will be detailed in further paragraphs.

(113) The two-stage charger system 102, as shown in FIG. 26, has a modular configuration, which includes: (a) a Block 1, also referred to herein as charger sub-system 104, which can be a bi-directional three-phase PFC rectifier or three single-phase PFC rectifiers connected in parallel. In an alternative embodiment, the module 104 can be a modular multilevel converter (MMC), which offers modularity and fault tolerance capability. (b) a charger sub-system (Block 2) 106 can be implemented as a single three-phase bridge, or multiple full bridges, or half bridges, depending on the DC link structure. (c) Block 3, also referred to herein as the changer sub-system 108, represents the integrated transformer sub-system, which can be implemented with two or three transformer sets coupled in parallel by means of separate structures, through star or delta connections. (d) Block 4, also referred to herein as a charger sub-system 110, may be implemented as a full bridge rectifier or half bridge rectifier for the HV battery 26. Depending on the transformer configuration in the modular sub-system 108, Block 4 may include two or three parallel full bridge rectifiers or parallelly connected half bridge rectifiers. (e) Charger sub-system 112 (Block 5) is coupled to the transformer sub-system 108 in the module (Block 3) and may be implemented with one or more full-bridge rectifiers or one or more half-bridge rectifiers for the LV battery 28.

(114) The operation of each block 2, 3, 4, 5 is influenced by the preceeding stage. Different combinations of the modular stages 104, 106, 108, 110, 112 (presented in Table III) result in numerous alternative topologies. For instance, the integrated transformers in Block 3 may be parallel connected, delta or star connected, or have a three-phase structure, with alternative bridge configurations (a half-bridge or a full-bridge) as well.

(115) TABLE-US-00003 TABLE III Summary of two-stage alternative topological configurations of the subject OBC T# Block 1 Block 2 Block 3 Block 4 Block 5 Split DC link 1 Three-phase Two full bridges Two separate transformers Two full bridges Two full bridges No Boost PFC or half bridges or half bridges or half bridges in in parallel in parallel parallel 2 Three-phase Two full bridges Two separate transformers Two full bridges Two full bridges Yes Boost PFC or half bridges or half bridges or half bridges in separately in parallel parallel 3 Three-phase Two full bridges Two separate transformers Two full bridges Two full bridges No Buck PFC or half bridges or half bridges or half bridges in in parallel in parallel parallel 4 Three-phase Two full bridges Two separate transformers Two full bridges Two full bridges Yes Buck PFC or half bridges or half bridges or half bridges in separately in parallel parallel 5 Three-phase Two full bridges Two separate transformers Two full bridges Two full bridges No Vienna PFC or half bridges or half bridges or half bridges in separately in parallel parallel 6 Three-phase Two full bridges Two separate transformers Two full bridges Two full bridges Yes Vienna PFC or half bridges or half bridges in parallel separately in parallel 7 Three-phase Three-phase Three separate transformers; primary delta Three full Three full bridges No Boost PFC bridge connected with capacitors; secondary bridges or half or half bridges series connected with capacitors bridges 8 Three-phase Three-phase Single three-phase transformer/three Three full Three full bridges No Boost PFC bridge separate transformers bridges or half or half bridges bridges 9 Three-phase Three-phase Three separate transformers; primary delta Three full Three full bridges No Buck PFC bridge connected with capacitors; secondary bridges or half or half bridges series connected with capacitors bridges 10 Three-phase Three-phase Single three-phase transformer/three Three full Three full bridges No Buck PFC bridge separate transformers bridges or half or half bridges bridges 11 Three Three full Single three-phase transformer/three Three full Three full bridges No single-phase bridges or half separate transformers bridges or half or half bridges Boost PFC bridges bridges 12 Three Three full Single three-phase transformer/three Three full Three full bridges No single-phase bridges or half separate transformers bridges or half or half bridges Buck PFC bridges bridges 13 Modular Two full bridges Two separate transformers Two full bridges Two full bridges No multilevel or half bridges or half bridges or half bridges in converter in parallel in parallel parallel 14 Modular Three full Single three-phase transformer/three Three full Three full bridges No multilevel bridges or half separate transformers bridges or half or half bridges converter bridges bridges 15 Modular Three-phase Three separate transformers; primary delta Three full Three full bridges No multilevel bridge connected with capacitors; secondary bridges or half or half bridges converter series connected with capacitors bridges

(116) The single-stage system 100 is depicted as a modular diagram shown in FIG. 27. Compared to the two-stage system 102, the single-stage system 100 does not need the DC link capacitor. The Block 6, also referred to herein as a module, stage, or charger sub-system 114, represents a single-stage AC-DC converter, which can be implemented as a Swiss-derived converter or a matrix converter. Blocks 2, 3, 4, 5 shown in FIG. 26 correspond to blocks 2, 3, 4, 5 depicted in FIG. 27. Various single-stage derived topologies contemplated in the subject OBC are reflected in Table IV.

(117) TABLE-US-00004 TABLE IV Summary of single-stage alternative topological configurations of the subject OBC. T# Block 6 Block 3 Block 4 Block 5 16 Three single-phase AC- Three separate transformers Three full bridges Three full bridges DC converters in parallel or half bridges or half bridges 17 Matrix-based three-phase Two separate transformers Two full bridges or Two full bridges AC-DC TAB converter half bridges in or half bridges in parallel parallel 18 Swiss-based TAB Two separate transformers Two full bridges or Two full bridges converter with split AC half bridges in or half bridges in capacitors parallel parallel 19 Three single-phase AC- Three separate transformers; Three full bridges Three full bridges DC converters in parallel primary delta connected with or half bridges or half bridges capacitors; secondary series connected with capacitors 20 Three single-phase AC- Single three-phase transformer/ Three full bridges Three full bridges DC converters in parallel three separate transformers or half bridges or half bridges

(118) One of the topologies of the two-stage charger 102 is presented in FIG. 28. This topology includes a bi-directional three-phase PFC rectifier 120 of the module 104, a DC link 21 with the capacitor C.sub.DC, two triple active full bridge converters 122 of the module (stage) 106 connected in parallel with six shim inductors L.sub.1-L.sub.6 of the two integrated transformers 124 of the module (stage) 108. The shim inductors L.sub.1-L.sub.6 are used for transferring power and achieving soft-switching for the MOSFET switches in the DC/DC converters in modules 110, 112, inherently, that may be obtained through leakage inductance control of the integrated transformers 124.

(119) By using the parallel structure (converters 122 and transformers 124), the maximum achievable power transfer capability is increased with reduced conduction losses and voltage/current stress on the circuitry components. The modular design of the three-phase high-power integrated OBC 102 shown in FIG. 28 provides high reliability and further improves the power transfer efficiency. A combined phase shift and duty ratio control is implemented in the configuration shown in FIG. 28 to minimize reactive power and circulating energy, and to enable the power flow in different directions (V2G), (G2V), and (H2L), as well as (G2B), and B2G).

(120) By means of electromagnetic integration of the transformers 124 and by splitting the power flow in parallel paths, the subject system 102 can be used for simultaneous charging of the HV and LV batteries at high-power ratings.

(121) There are numerous alternative embodiments contemplated in the subject OBC system 102.

(122) Architecture I (Three-Phase PFC+Single-Phase DC/DC Converter)

(123) This architecture embodiment, shown in FIG. 29, uses a three-phase PFC rectifier 130 (in the module 104) as the front-end stage prior to the DC link 21, which provides flexibility to the DC/DC stage 110, 112 configuration which is configured with single-phase DC/DC converters 132, 134, respectively. The tertiary winding 136 is integrated into the transformers 124 in the single-phase DC/DC converters 130, 132, to ensure simultaneous charging for the HV and LV batteries 26, 28 from the grid 18, as well as the H2L power flow.

(124) The Topology shown in FIG. 29 has a parallel structure in the DC/DC stage (formed by the modular sub-systems 106, 108, 110, 112). By means of splitting the power into two separate DC/DC converters 132/134, conduction losses and voltage/current stress can be reduced. The topology shown in FIG. 29 is highly symmetrical. This leads to a relatively simple control of the power flow in different directions.

(125) Alternative to the Topology shown in FIG. 29, the Topology 2 shown in FIG. 30 splits the DC link capacitor C.sub.DC into two capacitors. This configuration has the advantage of a lower voltage stress on the DC/DC stage components, which helps to reduce the cost and overall dimensions of the OBC system 102. Moreover, by splitting the DC link voltage, the converter gain of the TAB stage 110, 112 requires less phase shift modulation, which reduces the circulating power.

(126) An alternative embodiment of the subject two-stage system 102 is depicted in FIG. 31, where a three-phase Vienna PFC rectifier 144 is implemented as the front-end stage (module) 104, where the switching losses and voltage stress of each component can be reduced.

(127) Architecture II (Three-Phase PFC+Three-Phase DC/DC Converter)

(128) In this architecture, alternative topologies are contemplated which are focused on the DC/DC stage. Instead of using multiple single-phase converters connected in parallel, a three-phase integrated topology is suggested for two output ports, which can be categorized as a resonance-based and a PWM-based, shown in FIG. 32 and FIG. 33, respectively.

(129) The topology shown in FIG. 32, of the two-stage OBC configuration 102 of the subject system 10 uses a three-phase Boost PFC rectifier 146 (as the front stage 104) and a three-phase bi-directional CLLLC resonant DC/DC converter 148.

(130) The topology, shown in FIG. 33, uses the three-phase Boost PFC rectifier 146 and the three-phase TAB 150.

(131) Architecture III (Three Single-Phase PFC+Single-Phase DC/DC Converter)

(132) In this architecture, single-phase converters are implemented in a modular configuration, as shown in FIG. 34, which provides a high system level reliability. Each module 152, 154, 156 in the topology shown in FIG. 34, connects with one phase input 164, 166, 168, respectively, while having parallelly connected outputs 170, 172, 174. This configuration also has the advantage of flexibility since each module is identical and easy to replace.

(133) Architecture IV (Single-Stage System)

(134) In the architecture of the single-stage charger configuration 100 of the subject system 10, shown in FIGS. 35, 36, 37, the PFC rectifier and the DC/DC converter are combined in a single-stage implementation by removing the DC link capacitor. The AC capacitors are needed in this embodiment for filtering purposes. By means of parallel connection of the outputs, line frequency voltage ripple can be eliminated to a desired extent, which is another advantage in the three-phase system as compared to the single-phase OBC system.

(135) The topology of the configuration 100, shown in FIG. 35 is represented by three single-phase AC/DC TAB converters 158, 160, 162 interconnected in parallel.

(136) The configuration 100 using a matrix-based three-phase AC/DC TAB converter 180 is shown in FIG. 36. The Swiss-based topology embodiment, shown in FIG. 37, achieves all the main advantages of a SWISS Rectifier. The system shown in FIG. 37 uses the SWISS-based TAB converter 182 with split capacitors C.sub.AC can be controlled similar to a DC/DC converter while maintaining the achievable high efficiency at high-power rating. Fundamental knowledge of a buck-type DC/DC converter and a three-phase diode rectifier is sufficient to operate a three-phase PFC rectifier 184 with the sinusoidal input current and a regulated output voltage.

(137) Architecture V (MMC-Based System)

(138) In this architecture, the PFC rectifier stage is replaced with the MMC system 186, which brings modularity and fault tolerance capability. FIG. 38 shows the topology of an MMC-based system. In each module 188 of the MMC system 186, two kinds of circuit topologies are contemplated, including a half bridge 190 and/or a full bridge 192, as shown in FIG. 39A, 39B, respectively.

(139) The MMC-based integrated onboard charger 186 can be interfaced with the propulsion system of an EV, as shown in FIGS. 40A-40B, where MMC provides the flexibility to the winding reconfiguration of the motor. It helps prevent torque production and improve the efficiency and lifespan of the motor. FIG. 40A depicts a single-star configuration of the windings using MMC; while FIG. 40B depicts a single-delta configuration of the windings.

(140) Power Flow Control

(141) A conventional control strategy for a Triple Active Bridge (TAB)-based converter may be supported by the phase shift modulation (PSM) concept, which brings convenience for circuit modelling and control loop design. However, due to the nature of multi-directional power flow, this PSM-based control strategy suffers from high circulating power in the tank, especially when the converter operates at high power rating, which results in high conduction loss and demanding hardware design.

(142) To overcome the high current circulation issue, a combined phase shift—and—duty ratio power flow control strategy is proposed in the subject system.

(143) The subject control methodology is capable of simultaneous charging of both HV and LV batteries from the power system AC grid. Moreover, it can achieve the functionalities, including grid to HV battery (G2H) charging, grid to LV battery (G2L) charging, HV battery to LV battery (H2L) charging, and vehicle to grid (V2G) discharging.

(144) The subject control system 200 which has a three-loop structure is shown in FIGS. 26,27 and 41. In FIG. 41, φ represents the phase shift control variable, and δ represents the duty ratio control variable. The control system 200 is implemented with both φ and δ control functionality, and operates to achieve simultaneous charging while maintaining tight voltage regulations. The computation sub-system 202 provides a real time feedback for δ control.

(145) The topology of a TAB-based converter used in the subject OBC system 100, 102 is equivalent to the delta configuration, as shown in FIG. 42, where equivalent inductances are expressed as,

(146) $\begin{array}{cc}\{\begin{array}{c}{L}_{12}=L_1+L_2+\frac{L_1{L}_2}{L_3}\\ L_{23}=L_2+L_3+\frac{L_2{L}_3}{L_1}\\ L_{13}=L_1+L_3+\frac{L_1{L}_3}{L_2}\end{array}& \left(\mathrm{Eq}.23\right)\end{array}$
where L.sub.1, L.sub.2 and L.sub.3 are the primary, secondary and tertiary power inductors, respectively.

(147) As shown in FIG. 42, both P.sub.12 and P.sub.32 involve the control variable φ.sub.2, which implies that phase angle φ.sub.2 can be used as a control variable for the port-2 output voltage regulation. To minimize the circulating active power, the modulus sum of power transferred between ports P.sub.1, P.sub.2, P.sub.3 is to be minimized:
f(δ.sub.1,δ.sub.2,δ.sub.3,φ.sub.1,φ.sub.2,φ.sub.3)=Min{|P.sub.12|+|P.sub.23|+|P.sub.31|}  (Eq. 24)

(148) A quick solution can be obtained by reducing one freedom of control variable:
φ.sub.1=0, φ.sub.2=φ.sub.3  (Eq. 25)

(149) There are some restrictions on the reactive power control. Because of the three-port structure, reactive power flows between ports P.sub.1, P.sub.2, P.sub.3 are not inherently equal, which needs the boundary condition of δ control variable. φ.sub.1 is set as 0 to obtain maximum power flow capability. By using φ.sub.2=φ.sub.3, no circulating active power flows between port-2 and port-3. δ.sub.3 is utilized for the LV battery side voltage regulation. The control diagram for the subject control system 200 is expressed in FIG. 43.

(150) It is important to note that the subject power flow control strategy is applicable both to the three-phase integrated charging system, and the single-phase, as well as to the subject OBC system using any number of phases and power levels.

(151) FIG. 44 depicts a comparison result of the operation range between φ control and δ control. As presented in FIG. 44, the δ control enables wide operation range of HV and LV batteries, especially under unbalanced loads condition.

(152) The state-of-charge (SoC) map, shown in FIG. 45, describes the relation of charging modes (constant voltage (CV), constant current (CC), and constant power (CP)) and operation modes (G2B, G2V, G2L).

(153) A detailed SoC plot based on the subject control methodology in the time domain is illustrated in FIG. 46, where the LV charging power curve is determined by HV (high voltage) SoC curve.

(154) Although this invention has been described in connection with specific forms and embodiments thereof, it will be appreciated that various modifications other than those discussed above may be resorted to without departing from the spirit or scope of the invention as defined in the appended claims. For example, functionally equivalent elements may be substituted for those specifically shown and described, certain features may be used independently of other features, and in certain cases, particular locations of elements, steps, or processes may be reversed or interposed, all without departing from the spirit or scope of the invention as defined in the appended claims.