Control loop of a digital control device of a rotary electrical machine with excitation for a motor vehicle

Abstract

The invention relates to a control loop (10) which is installed in a voltage regulator of a motor vehicle alternator and controls an output voltage of the latter by adjusting an excitation current of the alternator. The control loop includes, at the input, means for measuring (31) the output voltage by sampling, generating a measurement signal (Um), error-calculation means (13) generating an error signal (e) equal to a difference between the measurement signal (Um) and a set value (Uo), means for processing the error signal (e) including an amplifier (14) and generating a control signal (Ysat) and, at the output, means (35) for generating a control signal (PWM) controlling excitation control means in accordance with the control signal (Ysat). According to the invention, the processing means also include a phase-advance filter (24).

Claims

1. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle of the type which can function as a generator supplying an output voltage (U.sub.b+) adjusted by an excitation current (I.sub.e), said digital control device (2) comprising means (11) for controlling said excitation current (I.sub.e) and said control loop (10) comprising, as input, means (31) for measurement by sampling of said output voltage (U.sub.b+) generating a measurement signal (U.sub.m), means (13) for error calculation generating an error signal (e) equal to a difference between said measurement signal (U.sub.m) and a set value (U.sub.0), means (14) for processing of said error signal (e) comprising an amplifier (14) and generating a control signal (Y.sub.sat), and, as output, means (35) for generation of a control signal (PWM) which controls said control means (11) according to said control signal (Y.sub.sat), wherein said processing means (14, 24) additionally comprise a phase advance filter (24), said phase advance filter (24) having a transfer function at Z which has the form: $\mathrm{FT}\left(z\right)=\frac{1-\left(1-1/a\right).Math.Z^{-1}}{1-\left(1-1/b\right).Math.Z^{-1}}$ where a is a first predetermined coefficient and b is a second predetermined coefficient such that a>b, and said phase advance filter (24) comprises means for implementation of a recurrence equation, said recurrence equation having the form:
Y.sub.n+1=Y.sub.nZ.sup.1Y.sub.nZ.sup.11/b+X.sub.nX.sub.nZ.sup.1+X.sub.nZ.sup.11/a where X and Y are respectively a digital input and a digital output of said phase advance filter (24), and a and b are respectively said first and second predetermined coefficients.

2. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 1, wherein said means (25, 26) for implementation comprise a multiplier (25) by a predetermined factor at the input of said phase advance filter (24), and a divider (26) by said predetermined factor at the output.

3. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 2, wherein said means (25, 26, 27, 28, 29) for implementation additionally comprise an adder (27) and elements (28, 29) for multiplication by values which are the inverse of said first and second predetermined coefficients.

4. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 3, wherein said phase advance filter (24) is in series with said amplifier (14).

5. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 3, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

6. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 2, wherein said phase advance filter (24) is in series with said amplifier (14).

7. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 2, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

8. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 1, wherein said phase advance filter (24) is in series with said amplifier (14).

9. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 8, wherein said means (14, 24, 34) for processing additionally comprise an integrator (34) in parallel with said amplifier (14) and said phase advance filter (24).

10. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 9, wherein said processing means (14, 17, 24, 34) additionally comprise a saturation block (17) which generates a disconnection signal (Cmd) controlling a switch (36) which disconnects said integrator (34) of said error calculation means (13) in the case of detection of a state of saturation of said control signal (Y.sub.sat).

11. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 10, wherein said integrator (34) is a low-pass filter (34).

12. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 10, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

13. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 9, wherein said integrator (34) is a low-pass filter (34).

14. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 13, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

15. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 9, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

16. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 8, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

17. Control loop (10) of a digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle according to claim 1, wherein said phase advance filter (24) has a nominal cut-off frequency (f.sub.c) which is substantially equal to 22 Hz, a transfer function in an open loop of said digital control device with a gain margin substantially equal to 22 dB, and a phase margin substantially equal to 80.

18. Digital control device (2) of a rotary electrical machine with excitation (1) for a motor vehicle, of the type which can function as a generator, comprising a control loop (10) according to claim 1.

19. Rotary electrical machine with excitation (1) for a motor vehicle of the type which can function as a generator, comprising a digital control device (2) according to claim 18.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) FIG. 1 is a schematic representation of a rotary electrical machine with excitation known in the prior art, provided with a digital control device comprising a control loop, and of its use in the on-board network of a motor vehicle.

(2) FIG. 2 is a process diagram of a control loop of the digital control device shown in FIG. 1, of a proportional integral type known in the prior art.

(3) FIG. 3 is an analogue representation of a type of phase advance filter implemented in the control loop according to the invention.

(4) FIGS. 4a and 4b show a frequential response of a digital embodiment of the phase advance filter shown in FIG. 3 (gain and phase respectively).

(5) FIG. 5 is a process diagram of a form of digital implementation of a phase advance filter implemented in a preferred embodiment of the control loop according to the invention.

(6) FIG. 6 is a process diagram of a preferred embodiment of the control loop according to the invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

(7) The rotary electrical machine with excitation represented schematically in FIG. 1 is, by way of non-limiting example, a three-phase alternator 1 provided with a digital control device 2.

(8) The stator 3 of the alternator 1 comprises three windings which are subjected to the rotary field created by the inductor 4 through which an excitation current I.sub.e passes.

(9) The alternating current which is produced in the stator 3 is rectified by a rectifier block 5 and filtered by a capacitor 6, such that the alternator 1 supplies a direct output voltage U.sub.b+ to the battery 7 and to the on-board network of the vehicle 8 which supplies power to the loads 9 (a connection by means of a power cable being schematised by a reactor L and a resistor R).

(10) The output voltage U.sub.b+ of the alternator 1 is kept constant when the load 9 and the speed of rotation are varied by means of a control loop 10 which acts on control means 11 of the excitation current I.sub.e on the basis of measurements 12 by sampling of this output voltage U.sub.b+.

(11) The control means 11 of the excitation current I.sub.e are generally constituted by power transistors 11 which function with switching and are controlled by a rectangular variable PWM duty cycle signal.

(12) In the most recent alternators 1 known in the prior art, the control loop 10 is mostly a proportional integral control loop equipped with an anti-saturation system with calculated feedback of the type shown in FIG. 2.

(13) The control loop 10 comprises at its input measurement means which are generally constituted by an analogue-digital converter in order to sample the output voltage U.sub.b+ of the alternator 1 and generate a measurement signal U.sub.m which is compared with a set value U.sub.o.

(14) Error calculation means 13 generate with a first operator Diff_1 an error signal e which is equal to a difference between the measurement signal U.sub.m and the set value U.sub.o.

(15) In the parallel structure represented in FIG. 2, the error signal e is amplified, firstly by a first amplifier 14 with a predetermined proportional gain K.sub.p, and secondly integrated by an integrator 15.

(16) An output voltage S.sub.a of the first amplifier 14 and an output voltage S.sub.i of the integrator 15 are added 16 in order to produce an intermediate control signal Y.

(17) A saturation block 17 makes it possible to adapt the format of the data of the control loop 10 to that of means for generation of the PWM control signal at the output, by supplying a control signal Y.sub.sat from the intermediate control signal Y.

(18) This control loop 10 of a known type additionally comprises an anti-saturation system with calculated feedback 18, the functioning of which is as follows:

(19) Non-Saturated Mode

(20) A value Y.sub.diff represents a difference between an error production before saturation Y and after saturation Y.sub.sat created by a second operator Diff_2 19.

(21) When the loop 10 is not saturated, the value Y.sub.diff is zero and does not disrupt the functioning of the proportional integral loop 10 (with a second integrator K.sub.i gain amplifier 20 in series with the integrator 15 having a first transfer function with the form FT=1/s). The anti-saturation system 18 is considered to be disconnected.

(22) Mathematically:
IF (Y=Y.sub.sat) THEN e.sub.i=e
where e.sub.i is an intermediate error signal at the input of the second amplifier 20 preceding the integrator 15.

(23) Saturated Mode

(24) The value Y.sub.diff is non-zero in saturated mode.

(25) The value Y.sub.diff in saturated mode attenuates to a greater or lesser extent (according to a saturator gain K.sub.lim of an additional amplifier 21) the loop error s.sub.i generated by the integral part 15, 20 via a difference created by a third operator Diff_3 22.

(26) Mathematically:
IF (YY.sub.sat) THEN e.sub.i=eK.sub.lim(YY.sub.sat)

(27) It should be noted that the structure of the anti-saturation system with calculated feedback type 18 comprises two difference operators (Diff_2 19 and Diff_3 22) and an amplifier K.sub.lim applied to a pure integrator 15.

(28) The problem encountered in this type of circuit known according to the prior art is that the use of an integral part 15, 20 in the control loop 10 makes it possible to reduce the control voltage drop when the regulator gain is decreased, but the phase margin and the gain margin are affected.

(29) For the purpose of solving this problem, the inventive body has studied the possibility of adding phase advance filtering in in the control loop 10 which controls the alternator 1, if the loop comprises in particular an integral part 15, 20, 23, as will be explained in association with FIG. 6.

(30) An analogue representation of a type of phase advance filter studied is shown in FIG. 3.

(31) The transfer function of this advance filter with the form (R1=R2=R is selected for simplified calculation):

(32) $\frac{\mathrm{Vs}}{\mathrm{Ve}}=\frac{1}{2}\left[\frac{1+j}{1+j\frac{}{2}}\right]$
where =RC

(33) It will be noted that with a very low frequency there is attenuation of a value of , i.e. 6 dB (20log 0.5 equal is 6 dB).

(34) This attenuation value can be adjusted if lesser or greater attenuation is required according to the control of the system by selecting appropriate values of R1 and R2.

(35) The low-frequency gain of this circuit is in fact provided by the equation R2/(R1+R2).

(36) It will also be noted that this phase advance filter is broken down into two basic filters:

(37) a derivative part (mathematical part situated at the numerator) which has a first cut-off frequency f.sub.c1=;

(38) an integral part (mathematical part situated at the denominator) which has a second cut-off frequency f.sub.c2=/ greater than the first cut-off frequency f.sub.c1.

(39) Within the context of a digital embodiment 24, it is thus possible to have an approach similar to the analogue study by using a transfer function at Z of the type:

(40) $\mathrm{FT}\left(z\right)=\frac{1-\left(1-1/a\right).Math.Z^{-1}}{1-\left(1-1/b\right).Math.Z^{-1}}$

(41) For the digital creation of a phase advance filter 24, the first predetermined coefficient a is greater than the second predetermined coefficient b.

(42) FIGS. 4a and 4b show an example of frequential response of a filter 24 of this type for a=2048, b=1024, and a sampling frequency f.sub.e=100 kHz using the equation

(43) $Z\left(f\right)=e^{j2\frac{f}{\mathrm{fe}}}$

(44) The frequential response of the digital phase advance filter 24 is in fact virtually similar to that of the analogue phase advance filter (same characteristics, i.e. a derivative part and an integral part).

(45) A form of digital implementation 24 shown in FIG. 5 can be provided by analysing a recurrence equation deduced from the transfer function at Z with the form:
Y.sub.n=Y.sub.nZ.sup.1Y.sub.nZ.sup.11/b+X.sub.nX.sub.nZ.sup.1+X.sub.nZ.sup.11/a

(46) A multiplier 25 by a predetermined factor M at the input makes it possible to increase the precision of calculation of the phase advance filter 24; the result at the output of the filter 24 is then divided by this predetermined factor M by a divider 26.

(47) As shown clearly in FIG. 5, the phase advance filter 24 according to the invention is also implemented by means of an adder 27 and elements of multiplication 28, 29 by values which are the inverse of the first and second predetermined coefficients a, b.

(48) Thanks to the multiplier 25 and the divider 26, truncation errors generated by these values are minimised.

(49) According to a preferred embodiment of the invention, the different functional blocks of which are shown in FIG. 6, the digital phase advance filter 24 is added in series with the amplifier 14 of the control loop 10.

(50) The description of these functional blocks is as follows:

(51) input signal 12 representing the voltage of the battery 7 or the voltage of the B+ terminal of the alternator 1;

(52) analogue filtering 30 (voltage anti-ripple, anti-aliasing filter), associated with the analogue-digital converter 31 and voltage divider 30 in order to adapt the voltage level for the analogue-digital converter 31;

(53) analogue-digital converter 31;

(54) means 13 for calculation of error between the measurement signal U.sub.m and the set value U.sub.o;

(55) digital order 32 which generates the set value U.sub.o required;

(56) anti-aliasing filter 33 associated with the decimation induced by the generation of the PWM control signal;

(57) first amplifier 14 of the proportional part of the control loop 10 (proportional gain K.sub.p designed to guarantee the stability of the regulator device 2 connected to the alternator 1 connected to the battery 7);

(58) adder block 16 between the proportional part 14 and the integral part 23;

(59) saturation block 17 making it possible to adapt the format of the data of the control loop 10 to that of the means 35 for generation of the PWM control signal between a minimum value Y.sub.min and a maximum value Y.sub.max;

(60) means 35 for generation of the PWM control signal (controlling the means 11 for control of the excitation current I.sub.e of the alternator 1), carried out by a comparison between a triangular reference signal (also known as a sawtooth signal) and the control signal Y.sub.sat obtained from the saturation block 17;

(61) switch 36 which connects or disconnects the integral part 23 according to a disconnection signal Cmd generated by the saturation block 17;

(62) second amplifier 20 (with an integrator gain K.sub.i which is designed to guarantee the stability of the control device 2 connected to the alternator 1 connected to the battery 7);

(63) very low-frequency low-pass cut-off frequency low-pass filter 34 which forms the integral part 23 of the control loop 10;

(64) disconnection signal Cmd of the switch 36 representing the saturation of the control loop 10 generated by the saturation block 17;

(65) PWM control signal controlling the power electronics 11 controlling the excitation current of the alternator 1;

(66) phase advance filter 24 (designed to guarantee the stability of the control subassembly connected to the alternator 1 and connected to the battery 7).

(67) This phase advance filter 24 has the essential characteristic of having a significant positive phase on a given frequency band. Its nominal cut-off frequency f.sub.c is designed to obtain a maximum phase advance in the control loop 10 in order to obtain, on the transfer function in an open loop of the control system, a maximum gain margin and phase margin.

(68) In the preferred embodiment of the invention, the control loop 10 is a proportional integral control loop 14, 23 which, unlike the control loops known in the prior art (such as the one shown in FIG. 2) additionally comprises an anti-saturation system with conditional detection 36, which makes it possible to optimise the time of return to the non-saturated mode.

(69) The integral part 23 of the control loop 10 comprising the second amplifier 20 and the integrator 34 is connected or disconnected by the saturation block 17 according to the state of saturation of the control signal Y.sub.sat.

(70) For this purpose, the saturation block 17 generates a disconnection signal Cmd controlling the switch 36 which applies to the input of the second amplifier 20 either the error signal e or a zero voltage by earthing 37.

(71) The detection of the saturation in the saturation block 17 is implemented by a numerical algorithm which uses the following signals:

(72) Y: intermediate control signal at the input of the saturation block 17;

(73) Y.sub.sat: control signal at the output of the saturation block 17;

(74) Cmd: disconnection control signal.

(75) The saturation detection algorithm is:
IF (Y=Y.sub.sat) THEN Cmd=0
OTHERWISE Cmd=1

(76) The functioning of this anti-saturation system in the proportional integral control loop 10 is then as follows:

(77) Non-Saturated Mode

(78) When the disconnection signal Cmd is in a zero logic state, the non-saturated mode is detected. The switch 36 connects the error signal e to the input of the integral part 23 (i.e. with the second integrator gain K.sub.1 amplifier 20 in series with the low-pass filter 34 which carries out the integration function 15). The anti-saturation system is considered to be disconnected.

(79) Mathematically:
IF (Y=Y.sub.sat) THEN Cmd=0 and e.sub.1=e
where e.sub.i is the intermediate error signal at the input of the second amplifier 20 preceding the low-pass filter 34.

(80) Saturated Mode

(81) When the control signal Cmd is in the logic state 1, the saturated mode is detected. The switch 36 then connects the input of the integral part 23 to a zero voltage in order to stop the development of output voltage s.sub.i of the integral part 23.

(82) Mathematically:
IF (YY.sub.sat) THEN Cmd=1 and e.sub.i=0.

(83) The output voltage s.sub.i of the integral part 23 remains fixed at a constant value during the saturated mode.

(84) In fact, a pure integration operator carries out a non-limited operation relative to the time [0, +[ defined by the mathematical function:

(85) $s_i=K_i.Math.{}_0^{+}\left(e_i\right).Math.dt$

(86) As a result, s.sub.i is equal to the value of s.sub.i at the moment of transition to saturated mode.

(87) However, a pure numerical integrator with a first transfer function with the form (transformed into Z):

(88) $\mathrm{FT}0\left(z\right)=\frac{c}{1-Z^{-1}}$
can be replaced by a second transfer function digital low-pass filter 34 with the form:

(89) $\mathrm{FT}1\left(z\right)=\frac{d}{1-\left(1-d\right).Math.Z^{-1}}$
with identical behaviour.

(90) For example, lines on the Bode plane carried out by the transformation

(91) $Z\left(f\right)=e^{j.Math.2\frac{f}{\mathrm{fe}}}$
and parameters

(92) $c=\frac{1}{2^{20}}$
and

(93) 0$d=\frac{1}{2^{20}}$
with a sampling frequency of f.sub.e=100 KHz show that beyond 30 MHz, the behaviour of the low-pass filter 34 and that of the pure integrator 15 are identical.

(94) The inventive body has been able to determine that the implementation in an alternator 1 which outputs 300 A of a digital regulator device 2 comprising a control loop 10 as described above with a phase advance filter 24 with a nominal cut-off frequency f.sub.c substantially equal to 22 Hz made it possible to obtain a maximum gain margin substantially equal to 22 dB, and a maximum phase margin substantially equal to 80.

(95) It will be appreciated that the above description would apply in similar terms to rotary electrical machine models with excitation other than the three-phase alternator represented in FIG. 1.

(96) The numerical values indicated correspond to experimental developments and computer simulations carried out by the applicant company, and are provided purely by way of example.

(97) The location of the phase advance filter 24 in the proportional part 14 of the control loop 10 is also only an example corresponding to a preferred embodiment of the invention; other locations in the control loop 10 can be provided as an alternative, and would provide similar advantages in terms of phase and gain margins for high-power machines 1.

(98) The invention thus incorporates all the possible variant embodiments which would remain within the scope defined by the following claims.