Dual-band phased array antenna with built-in grating lobe mitigation

Abstract

A dual-Band phased array antenna with built-in grating lobe mitigation includes an array of radiating elements capable of working at both bands and arranged at distances small enough, avoiding grating lobes with respect to the lower band within the desired field of view. The radiating elements are arranged in planar subarrays that can be steered independently from each other and each of the subarrays has a different boresight normal vector, so that grating lobes in the upper band is mitigated.

Claims

1. A dual-band phased array antenna with built-in grating lobe mitigation, comprising: an array of radiating elements configured to operate at both upper and lower bands of the dual bands, wherein the radiating elements of the array are arranged at distances d for avoiding grating lobes with respect to the lower band, the distances d are less than $.Math.\frac{1}{1+\sin \frac{}{2}},$ is a wavelength of the lower band, and is a grating lobe-free scan angle, wherein the radiating elements are arranged in independently steerable planar subarrays, wherein each of the independently steerable subarrays has a different boresight normal vector to mitigate grating lobes in the upper band while coherently adding up signals of the independently steerable planar subarrays to form a beam of the antenna.

2. The dual-band antenna of claim 1, wherein the subarrays are arranged in a polyhedral surface of the antenna.

3. The dual-band antenna of claim 1, wherein the subarrays are arranged lying tangentially on a surface of a virtual sphere.

4. The dual-band antenna of claim 1, wherein the subarrays are arranged, as seen in a boresight direction of the antenna, a rectangular, a circular or a quadratic shape.

5. The dual-band antenna of claim 1, wherein the array of radiating elements is arranged on a mechanically steerable gimbal system.

6. The dual-band antenna of claim 1, wherein the distance d between the radiating elements is smaller than the wavelength .

7. The dual-band antenna of claim 1, wherein the subarrays are arranged in a concave or convex surface of the antenna.

8. The dual-band antenna of claim 7, wherein an angle between the boresight normal vectors of two subarrays located at opposite edges of the antenna is 6 degrees.

9. The dual-band antenna of claim 7, wherein an angle between the boresight normal vectors of two subarrays located at opposite edges of the antenna is 12 degrees.

10. The dual-band antenna of claim 1, wherein an angle between the boresight normal vectors of any two subarrays is 6 degrees.

11. The dual-band antenna of claim 1, wherein an angle between the boresight normal vectors of any two subarrays is 12 degrees.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) The invention may be more fully understood by the following more detailed description with corresponding figures wherein:

(2) FIG. 1 shows the antenna pattern of an array antenna with

(3) $\left(\frac{d}{}<0,5\right),$
d being the distance between neighboring radiating elements,

(4) FIG. 2 shows the antenna pattern of an array antenna with

(5) $\left(\frac{d}{}>0,5\right),$

(6) FIG. 3 shows an exemplary embodiment of the invention with 97 planar subarrays,

(7) FIG. 4 shows an excerpt from the array of FIG. 3 indicating the design and normal vectors of the subarrays,

(8) FIG. 5 shows three other embodiments of the array antenna according to the invention,

(9) FIG. 6 shows the computer simulation results indicating the pattern with a planar subarray arrangement according to the prior art,

(10) FIG. 7 shows the computer simulation results indicating the pattern using a subarray arrangement according to the present invention.

DETAILED DESCRIPTION

(11) It is well known in phased array theory that the antenna pattern for sufficiently large arrays can be assumed to be the product of the element pattern and the array factor as in equation Eq 1, shown for a linear array, but not limited to linear arrays:

(12) $\begin{array}{cc}E\left(\right)=\underset{\underset{\mathrm{Element}\mathrm{Pattern}}{}}{E_{\mathrm{RE}}\left(\right)}\underset{\underset{\mathrm{Array}\mathrm{Factor}}{}}{\underset{n}{.Math.}{A}_n{e}^{-i2\frac{d}{}\left(\sin -\sin {}_0\right)n}}& \mathrm{Eq}1\end{array}$

(13) The first term E.sub.RE() in Eq 1 is called element pattern, whereas the sum is commonly known as array factor. In this second term the individual signals with amplitude A.sub.n and Phase

(14) $2\frac{d}{}\left(\sin -\sin {}_0\right)n$
are summed. d designates the distance between neighboring radiating elements. The phase depends on the position n*d within the array, the wavelength , the desired direction and the steering direction .sub.0. The array factor will have maximal amplitude when the phase in the exponential term becomes a multiple of 2 as noted in Eq 2:

(15) $\begin{array}{cc}2\frac{d}{}\left(\sin -\sin {}_0\right)=k2k& \mathrm{Eq}2\end{array}$

(16) If

(17) $\frac{d}{}$
is smaller than 0.5, Eq 2 is solvable only for k=0 and only one major lobe exists in the whole scanning range /2<theta</2 that is the so-called main lobe 10 as shown in FIG. 1 where the patterns according Eq 1 in dB above isotropic radiation is plotted. In cases where

(18) $\frac{d}{}$
becomes larger than 0.5 as for e.g. operating the same antenna at higher frequencies solutions with values of k different from 0 are additionally possible, which results in secondary lobes or grating lobes. The direction of the grating lobes are given as solutions of Eq 2:

(19) The directions of the grating lobes are defined according to Eq 3

(20) $\begin{array}{cc}{}_k={\sin }^{-1}\left(\sin {}_0+\frac{k}{d}\right);.Math.k.Math.<\mathrm{Int}\left[.Math.\frac{d\left(1-\sin \left({}_0\right)\right)}{}.Math.\right]& \mathrm{Eq}3\end{array}$

(21) As an example for

(22) 0$\frac{d}{}=3/2$
the pattern of an array as in FIG. 1 with a three times higher operating frequency is shown in FIG. 2, where three grating lobes 20 can clearly be identified beside the main lobe 10. The directions of the grating lobes 20 for the above example

(23) $\frac{d}{}=3/2$
according to Eq 3 are at: .sub.GL={1.42, 0.395, 0.951}.

(24) This may be easily extended to 2 dimensional arrays, as known from the literature, too.

(25) Let us now consider two linear arrays one (index I) tilted by +/2 and the second (index r) by /2, so that both array's normal vectors are tilted by . Both arrays are electronically steered so that their main beams are looking in the same direction .sub.0. The first array has to be steered to .sub.0/2 and the second to .sub.0+/2. According to Eq 2 are the directions of resulting beams:
.sub.0l=sin.sup.1(sin(.sub.0/2))+/2Eq 4
.sub.0r=sin.sup.1(sin(.sub.0+/2))/2Eq 5

(26) So .sub.0l=.sub.0r=.sub.0 and the resulting signals received or transmitted by the arrays will add up coherently.

(27) The grating lobe behavior is different as it is shown in Eq 6 and Eq 7:

(28) $\begin{array}{cc}{}_{1l}={\sin }^{-1}\left(\sin \left({}_0-/2\right)+\frac{}{d}\right)+/2& \mathrm{Eq}6\\ {}_{1r}={\sin }^{-1}\left(\sin \left({}_0+/2\right)+\frac{}{d}\right)-/2& \mathrm{Eq}7\end{array}$

(29) Now it is obvious that .sub.1l.sub.1r, so that the first grating lobe will direct to different solid angles and therefore will have less integration gain as the main beam putting both arrays together. As a result, the ratio between main lobe directivity and first grating lobe directivity will improve. The same is true for all grating lobes entering the real space.

(30) The effect can even be improved having more than two subarrays each tilted against each other. If the arrays are arranged in a two-dimensional grid, and each array has a different normal vector from each other, the resulting grating lobe will be widened up in two dimensions with a significant improvement of the main lobe to grating lobe ratio, especially for large arrays.

(31) In the following several concrete examples of antennas implementing the above described principle are shown.

(32) The array of FIG. 3 approximately is of a circular shape and consists of 97 planar subarrays 100 advantageously arranged in columns and lines. The phase centers of each subarray is indicated by respective dots 101. Each of the subarrays 100 is directed to a different solid angle. Each subarray contains 64 radiating elements 110 (shown as individual dots) advantageously arranged in columns and lines. The 3-D arrangement of the individual subarrays 100 becomes visible from FIG. 4, which shows an enlarged section of FIG. 3 as marked by the square Q in the middle of FIG. 3. FIG. 4 shows nine subarrays 100 each comprising of 64 radiating elements 110. For each subarray 100 the respective normal vectors 120 are illustrated in a 3-D representation.

(33) The face of each subarray is squinting in a different direction. In the exemplary embodiment of FIG. 3 the normal vectors of the subarrays vary gradually from about 3 degree from the left to +3 degree to the right, as well as from the lower to the upper subarrays. The sectional view along A-A shows the resulting convex arrangement of the subarrays within the same line (for a better understanding of the underlying design principle the angles between neighboring subarrays are shown in an excessive way).

(34) In an advantageous embodiment each subarray may be arranged according to a tangential plane touching a virtually taut sphere at its phase centers 101. Thereby a multi-facetted surface of the antenna is built where each facet corresponds to one of the subarrays.

(35) In other words, the antenna surface thus created looks like the spherical segment of a mirror ball. The grid constants of the subarray radiating elements are preferably approximately half the wavelength of the lower operating band avoiding grating lobes in this operation band (the resulting pattern of each subarray is shown in FIG. 1), whereas the pattern in the upper operating band (from known art) will have grating lobes as expected (see FIG. 2).

(36) The signals of each radiating element within a subarray are coherently summed after phase shifting in order to steer the beam, either analog by an appropriate radio frequency combiner or digitally using an analog digital converter behind each radiating element. In the advantageous version of an AESA antenna additionally TRMs are used.

(37) The phase centers 101 of the subarrays shown as white dots in FIG. 3 are then connected for further signal combining.

(38) To form a beam with the exemplary phased array antenna, each subarray has to be steered to a slightly different direction, according to its squint angle and the desired beam direction. In the upper operating band where grating lobes appear each grating lobe will then point to a different direction as described in Eq 6 and Eq 7. As a result of this subarray arrangement the grating lobes will be suppressed by more than 15 dB compared to a planar array at a scanning angle up to +/15 deg.

(39) FIG. 5 shows three further embodiments of the antenna design according to the invention. The examples are based on a two-dimensional antenna, the subarrays of which are arranged in lines and columns similar to the example shown in FIG. 3.

(40) In each example a cross-sectional view along one column of arrays is shown.

(41) V1: a convex arrangement of the facets/subarrays 100 (e.g. part of the surface of a mirror ball),

(42) V2: concave arrangement of the facets/subarrays 100,

(43) V3: alternating/irregular arrangement of the facets/subarrays 100.

(44) The related normal vector 120 directions are also shown for each subarray.

(45) In addition, other arrangements of the subarrays are possible. For example, regular or irregular polyhedral arrangements of subarrays may be used. In another example the polyhedral surface of the antenna may approximate a section of an ellipsoid or the like.

(46) The squint angles between the subarrays may be fairly small, in particular if the number of subarrays or the overall seize of the phased array antenna is large. In principle the squint angles are based on an optimization task and are pending on the used array design, size and steering direction. In the exemplary embodiment of FIG. 3 the squint angles are within the interval [3,+3] degree for the north-south and west-east direction using the cardinal directions. For larger arrays the angles might even be less than 3 degree, for smaller arrays the angles have to be increased e.g. [6, +6] degree. In summary, the maximum squint angle depends on the design of the array, number of subarrays and the maximum steering angle of a subarray, so that all subarrays are still able to focus on the same target. The maximum steering angle of the antenna is reduced by the maximum squinting angle of any subarray with respect to the master subarray compared to a planar arrangement. Here, the master subarray is defined as the center for the angle measurement for all other subarrays.

(47) A computer simulation shows this behavior of the grating lobe suppression with a dual-band antenna according to the invention compared to an antenna without the implemented invention using the same number and size of subarrays.

(48) As illustrated in FIG. 6, for a planar subarray arrangement according to prior art grating lobes 200 exist beside the main lobe 10. By contrast, using the inventive dual-band phased array antenna the grating lobes 210 are highly suppressed (see FIG. 7) e.g. about 15 dB at 0.35 Theta/rad compared to the prior art antenna.

(49) Without using the invention the grating lobes 200 are highly disturbing the signal reception and are decreasing the detection quality. However, by usage of the invention these grating lobes are significantly reduced as required.

(50) This written description uses examples of the subject matter disclosed to enable any person skilled in the art to practice the same, including making and using any devices or systems and performing any incorporated methods. The patentable scope of the subject matter is defined by the claims, and may include other examples that occur to those skilled in the art. Such other examples are intended to be within the scope of the claims.

LIST OF ABBREVIATIONS

(51) AESA active electronically scanned array

(52) Eq equation

(53) GL grating lobe

(54) ML main lobe

(55) RE radiating element

(56) RF radio frequency

(57) TRM transmit receive module