DIGITAL MULTI-BAND PREDISTORTION LINEARIZER WITH NON-LINEAR SUBSAMPLING ALGORITHM IN THE FEEDBACK LOOP

Abstract

A concurrent multi-band linearized transmitter (CMLT) has a concurrent d a multi-band predistortion block (CDMPB) and a concurrent multi-band transmitter (CMT) connected to the CDMPB, The CDMPB can have a plurality of digital baseband signal predistorter blocks (DBSPBs), an analyzing and modeling (A&M) stage, and a signal observation feedback loop. Each DBSPB can have a plurality of inputs, each corresponding to a single frequency band of the multi-band input signal, and its output corresponding to a single frequency band; each output connect corresponding to an input of the CMLT. The A&M stage can have a plurality of outputs connected to and updating the parameters of the DBSPBs, and a plurality of inputs connected to either both outputs of the signal observation loop or the output of the subsampling loop and to outputs of the DBSPBs. The A&M stage can perform signals' time alignment, reconstruction of signals and compute parameters of DBSPBs.

Claims

1. A concurrent multi-band linearized transmitter comprising: a concurrent digital multi-band predistortion block; a concurrent multi-band linearized transmitter connected to said concurrent digital multiband predistortion block.

2. The concurrent multi-band linearized transmitter of claim 1, wherein said concurrent digital multi-band predistortion block further comprises: a plurality of digital baseband signal predistorter blocks; an analyzing and modelling stage; and a signal observation feedback loop,

3. The concurrent multi-band linearized transmitter of claim 2, wherein said plurality of digital baseband signal predistorter blocks further comprises: a plurality of inputs, each input corresponding to a single frequency channel; a plurality of outputs, each output corresponding to a single frequency channel, and each output connected to an input of said concurrent multi-band transmitter.

4. The concurrent multi-band linearized transmitter of claim 2, wherein said analyzing and modeling stage further comprises: a plurality of outputs connected to and updating the parameters of said digital baseband signal predistorter block; a plurality of inputs connected to both (a) said outputs of said signal observation feedback loop, and (b) to said outputs of said digital baseband signal predistorter blocks.

5. The concurrent multi-band linearized transmitter of claim 2, wherein said analyzing and modeling stage is further adapted to: perform time alignment of complex baseband signals from sampling said outputs of said concurrent multi-band transmitter; and perform the reconstruction of the complex baseband signals from sampling said outputs of said concurrent multi-band transmitter.

6. The concurrent multi-band linearized transmitter of claim 2, wherein said signal observation feedback loop further is further adapted to: down-convert samples of the RF signals at said output of the concurrent multi-band transmitter of claim 1; and extract from said down-converted samples a baseband equivalent for all frequency channels.

7. The concurrent multi-band linearized transmitter of claim 2, wherein said signal observation feedback loop further comprises for each channel an RF filter; A signal down conversion block; and an analog-to-digital converter (ADC).

8. The concurrent multi-band linearized transmitter of claim 2, wherein said signal observation feedback loop further comprises: a single subsampling-based receiver to down-convert samples output from a concurrent multi-band transmitter.

9. The concurrent multi-band linearized transmitter of claim 8, wherein said single subsampling -based receiver further comprises: an RF filter; a track and hold (T&H) block; and an analog-to-digital converter (ADC).

Description

BRIEF DESCRIPTION OF THE DRAWINGS

[0017] The present invention will become more fully understood from the detailed description and the accompanying drawings, wherein:

[0018] FIG. 1 is a block diagram of the dual-band digital predistortion architecture according to an exemplary embodiment of the present invention.

[0019] FIG. 2 is an alternate embodiment illustrating a detailed block diagram of the architecture of FIG. 1, using subsampling based feedback loop.

[0020] FIG. 3 is one embodiment illustrating a detailed block diagram of the subsampling receiver architecture of FIG. 2.

[0021] FIG. 4A is the fundamental signal representation at the input of the dual-band transmitter.

[0022] FIG. 4B is the fundamental signal representation at the output of the dual-band transmitter and intermodulation terms at the output of the nonlinear transmitter.

[0023] FIG. 5 is dual-band RF signal and the frequency position of the subsampling harmonics.

[0024] FIG. 6 is a flowchart illustrating the steps of the execution of finding the possible subsampling frequencies.

[0025] FIG. 7A is the power spectrum of the predistortion results for 880 MHz, using dual-band digital prediction technique with dual-branch feedback loop. Spectrums marked with circles are the output without linearization, those marked with diamonds are the output after linearization, and those marked with solid rectangles are the inputs signals.

[0026] FIG. 7B is the power spectrum of the predistortion results for 1978 MHz, using dual-band digital prediction technique with dual-branch feedback loop, using the same codes as described for FIG. 7A.

[0027] FIG. 8 is the Power spectrum of the output of the dual-band nonlinear transmitter including the two fundamental RF signals and their inter-modulation products and harmonics.

[0028] FIG. 9 is the Power spectrum of the captured signal using an ADC operating at 619.8 MHz sampling frequency.

[0029] FIG. 10A is the power spectrum of the predistortion results for 880 MHz. using dual-band digital prediction technique with subsampling feedback loop.

[0030] FIG. 10B is the power spectrum of the predistortion results for 1978 MHz, using dual-band digital prediction technique with subsampling feedback loop.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0031] The following description of the preferred embodiment(s) is merely exemplary in nature and is in no way intended to limit the invention, its application, or uses.

[0032] Broadly, an embodiment of the present invention provides multiple branch digital predistortion linearization architecture and digital signal processing algorithms for impairments-free operation and linearized multi-band transmitter.

[0033] Referring to FIG. 1, the system block diagram of the dual-band linearization architecture 100 is displayed. The input signals, x1 and x2, 105 are fed into two distinct predistorters blocks 110. The predistorted signals 115 are converted from digital to analog 120 and up-converted 125 to RF frequencies. Then the two RF signals are combined 130 and amplified by the power amplifier 135.

[0034] For digital predistortion linearization and identify the inverse model, the sample of the RF signal are captured using dual-band coupler 140. Then the RF signals are bandpass filtered 145, frequency down converted 150, digitized using analog-to-digital converters 155. The digital output samples 160, the input signals 105 and predistorted signals 115 are used in the analyzing stage 165 for nonlinear model identification and reverse modeling.

[0035] The feedback path of the dual-band linearizer requires the use of two down-conversion stages 150, as well as bandpass fitters 145 to remove most of the imperfections caused by the power amplifier. The predistorted inputs, x.sub.pd1 and x X.sub.pd2, 115 as well as the output of each band of the PA, y.sub.1 and y.sub.2, 160 are used to generate the predistorter signal processing model 110. The processing model equations of the linearization processing algorithm 165 for prediction and compensation of the distortions and intermodulations is as follows:

[00001]$\begin{array}{cc}{x}_{\mathrm{pd}.Math..Math.1}\left(n\right)=\underset{m=0}{\overset{M-1}{.Math.}}.Math..Math.\underset{k=0}{\overset{K-1}{.Math.}}.Math..Math.\underset{j=0}{\overset{k}{.Math.}}.Math..Math.c_{1,j,k,m}.Math.x_1\left(n-m\right).Math.{.Math.x_1\left(n-m\right).Math.}^{k-j}.Math.{.Math.x_2\left(n-m\right).Math.}^j.Math..Math.x_{\mathrm{pd}.Math..Math.2}\left(n\right)=\underset{m=0}{\overset{M-1}{.Math.}}.Math..Math.\underset{k=0}{\overset{K-1}{.Math.}}.Math..Math.\underset{j=0}{\overset{k}{.Math.}}.Math..Math.c_{2,j,k,m}.Math.x_2\left(n-m\right).Math.{.Math.x_2\left(n-m\right).Math.}^{k-j}.Math.{.Math.x_1\left(n-m\right).Math.}^j& \left(1\right)\end{array}$

[0036] Where x.sub.1(n) and x.sub.2(n) are the input signals, x.sub.pd1(n) and x.sub.pd2(n) are the predistorted signals to the input of the dual-band transmitter, c.sub.1,j,k,m and c.sub.2,j,k,m are the identified model's coefficients, and finally M is the order of the memory effect and K is the order of nonlinearity.

[0037] Concurrent multi-band receiver architectures require a bandpass filter 145, down-conversion stage 150, and ADC 155 for the translation of each RF frequency bands to baseband. Using subsampling with a high speed ADC allows the elimination of all these components; however, the user needs to make sure that the signals don't overlap in the subsampled spectral domain.

[0038] Sampling multi-bands at the same time also eliminates the time delay taken between different band paths caused by the filters. FIG. 2 displays the dual-band predistortion architecture with a subsampling feedback loop 200. At the feedback loop, it consists of optional bandpass filters 245, a track and hold 250, an analog-to-digital converter 255, a digital conversion unit 260, and analyzing stage 270.

[0039] Sampling the band-limited RF signal at frequency rates much lower than the carrier frequency, but higher than signal bandwidth folds the RF signal to the lower frequencies, where these replicates of the RF signal at baseband or intermediate frequencies can he used to reconstruct the baseband signal. To make sure that there is no aliasing between the replicas, the subsampling rate should be chosen in the following range:

[00002]$\begin{array}{cc}\frac{2.Math..Math.f_U}{n}{f}_s\frac{2.Math..Math.f_L}{n-1}.Math..Math.\mathrm{where}.Math..Math.1n.Math.\frac{f_U}{B}.Math..Math..Math.f_{s}2B.Math..Math.\mathrm{Nyquist}.Math..Math.\mathrm{rate}& \left(2\right)\end{array}$

[0040] where f.sub.L and f.sub.U are the lower and upper frequencies of the band-limited RF signal, B=f.sub.Uf.sub.L is the signal bandwidth, and n is an integer value.

[0041] FIG. 3 shows a general block diagram of subsampling-based receiver 300. It consists of RF bandpass filter 305, low-noise amplifier (LNA) 310, subsampling receiver 345 including the track and hold (T&H) 320, and ADC 325 followed by baseband digital signal processing (DSP) unit 330. The T&H 320 is required to expand the analog bandwidth of the receiver and defines the RF range of receiver operation. The sampling clock of the T&H 320 and ADC 325 are chosen from (2) to avoid any aliasing with the other RF signals.

[0042] In dual-band operation transmitter with nonlinearity, the first arid second bands will produce intermodulation, cross modulation and harmonic products. FIG. 4 shows the power spectrum of the input signal, and the output signal when passed through a third order nonlinear system. The input signals as two-tone signal around carrier frequencies of .sub.1 and .sub.2 (FIG. 4(a)) the output signals are around carrier frequencies of .sub.1 and .sub.2 and the 3rd order intermodulation frequencies of 2.sub.1.sub.1 and 2.sub.2.sub.1 (FIG. 4(b)). The unwanted intermodulation can he classified into three groups of cross-modulation, in-band intermodulation, and out-of-band intermodulation terms. This latter is usually filtered out before transmission using RF filter to avoid signal quality degradation and interference with the signals at the adjacent channels.

[0043] Now considering two RF signals at carrier frequencies of .sub.2 and .sub.2, with their respective bandwidths B1 and B2 as shown in FIG. 5, the subsampling frequency, fs, must be chosen to ensure that the two signals do not overlap in the subsampled domain. Taking into account the subsampling theorem for sampling the multi-band signals and following the neighbor and boundary constraints, an iterative process is used to find all the valid subsampling frequencies for the two fundamental frequencies.

[0044] The, out-of-band intermodulation-modulation, and harmonics generated by the fundamental signals are not required for the predistortion application; therefore, an iterative subsampling algorithm has been developed to subsample the RF signals without any overlap with the other unwanted RF signals. FIG. 6 is the flowchart of the developed iterative subsampling algorithm to find the valid subsampling frequencies so that the replicas of the wanted RF signals have no overlap with the harmonics and intermodulation frequency terms.

[0045] Referring to FIG. 7, there is shown the measured output spectrum of the dual-band transmitter 170 for three cases: 1) without using the dual-band digital pre-compensator 110, 2) using the dual-band digital pre-compensator 110 3) the input signal. The output spectrum of case-2 with digital pre-compensator shows that the digital pre-compensator 110 can compensate for the cross-modulation and in-band inter-modulation terms introduced by the transmitter nonlinearity.

[0046] Referring to FIG. 8, there is an example of the power spectrum at the output of the concurrent dual-band nonlinear transmitter which contains two fundamental RF frequencies and their corresponding harmonics and inter-modulation products.

[0047] As an example for the application of this invention, FIG. 9, points up the power spectrum after the subsampling feedback loop which shows the two fundamental RF signals at 260.2 MHz and 118.6 MHz when the subsampling frequency of 619.8 is used following is determination by the developed iterative subsampling algorithm illustrated in FIG. 6. The spectrum in FIG. 9 shows that the harmonics and inter-modulation terms have no interference with the two desired fundamental signals.

[0048] Referring to FIG. 10, there is shown the measured output spectrum of the dual-band transmitter 280 for three cases: 1) without using the dual-band digital pre-compensator 210, 2) using the dual-band digital pre-compensator 210, 3) the input signal. The output spectrum of case-2 with digital pre-compensator shows that the digital pre-compensator 210 can compensate for the cross-modulation and in-band inter-modulation terms introduced by the transmitter dynamic nonlinearity with memory effects.