Class D Audio Amplifier and Method for Reading a Current Supplied by the Amplifier

Abstract

A circuit includes a final stage that includes an H-bridge comprising first and second half-bridges. A read circuit is configured to read a load current supplied by a class-D audio-amplifier to a load. The read circuit is configured for estimating the load current by reading a current at an output by the first or second half-bridge by measuring a drain-to-source voltage during an ON period of a power transistor of the H-bridge. A sensing circuit is configured to detect a first drain-to-source voltage from a transistor of the first half-bridge and a second drain-to-source voltage from a corresponding transistor of the second half-bridge. The sensing circuit is also configured to compute a difference between the first drain-to-source voltage and the second drain-to-source voltage and to perform an averaging operation on the difference to obtain a sense voltage value to be supplied to an analog-to-digital converter.

Claims

1. A circuit comprising a final stage that includes an H bridge comprising a first half-bridge and a second half-bridge; a read circuit configured to read a load current supplied by a class-D audio-amplifier to a load, the read circuit being configured for estimating the load current by reading a current at an output by the first or second half-bridge by measuring a drain-to-source voltage during an ON period of a power transistor of the first half-bridge or second half-bridge; and a sensing circuit comprising a circuit portion configured to detect a first drain-to-source voltage from a transistor of the first half-bridge and a second drain-to-source voltage from a corresponding transistor of the second half-bridge, the sensing circuit comprising a first sub-circuit configured to compute a difference between the first drain-to-source voltage and the second drain-to-source voltage, which are detected by the circuit portion, and a second sub-circuit configured to perform an averaging operation on the difference to obtain a sense voltage value to be supplied to an analog-to-digital converter.

2. The circuit according to claim 1, wherein the power transistor comprises a low-side transistor of the first half-bridge or the second half-bridge.

3. The circuit according to claim 1, wherein the first sub-circuit comprises a differential amplifier.

4. The circuit according to claim 1, wherein the second sub-circuit comprises a low-pass filter.

5. The circuit according to claim 1, wherein the circuit portion comprises respective protection switches set between drain and source nodes of the transistors of the first half-bridge and of the second half-bridge and inputs of the first sub-circuit.

6. The circuit according to claim 1, wherein the circuit portion comprises a sensing network includes an auxiliary transistor with a gate and a drain, wherein the gate of the auxiliary transistor is coupled to the gate of a respective transistor of the first half-bridge or second half-bridge and wherein a drain of the auxiliary transistor is coupled to the drain of the respective transistor of the first half-bridge or second half-bridge.

7. The circuit according to claim 6, wherein the circuit portion further comprises a sense resistance between a source of the auxiliary transistor and ground of a corresponding half-bridge.

8. The circuit according to claim 7, wherein the sense resistance is sized so that the sense resistance has a higher value than a switch-on resistance of the auxiliary transistor, such as to render a sum of the switch-on resistance of the auxiliary transistor and of the sense resistance approximately equal to the sense resistance.

9. The circuit according to claim 7, wherein the sense resistance is sized so that the sense resistance has a lower value than a switch-on resistance of the auxiliary transistor, such as to render a sum of a switch-on resistance of the auxiliary transistor and of the sense resistance approximately equal to the switch-on resistance.

10. The circuit according to claim 7, further comprising a reference power MOSFET configured to generate a reference drain-to-source voltage, the reference power MOSFET being electrically and thermally coupled to the power transistor that supplies the current at the output with a known aspect ratio and current.

11. A method for reading a load current supplied by the class-D audio-amplifier to the load using the circuit according to claim 1.

12. A method for reading a load current supplied by a class-D audio amplifier to a load by estimating the load current, the method comprising: reading a current supplied by a power transistor of a first half-bridge or a second half-bridge of an H bridge of a final stage of the amplifier; detecting a first drain-to-source voltage from a transistor of the first half-bridge; detecting a second drain-to-source voltage from a corresponding transistor of the second half-bridge; computing a difference between the first detected drain-to-source voltage and the second detected drain-to-source voltage; and performing an averaging operation on the difference to obtain a sense voltage value.

13. The method according to claim 12, further comprising obtaining the load current by comparing the sense voltage value with a reference drain-to-source voltage.

14. The method according to claim 12, further comprising obtaining the load current by comparing the sense voltage value with a voltage on a reference resistor.

15. The method according to claim 12, wherein reading the current comprises reading the current supplied by a low-side transistor of the first half-bridge or the second half-bridge of the H bridge of a final stage of the amplifier.

16. The method according to claim 12, wherein reading the current comprises reading a current supplied by a power transistor of the first half-bridge and also reading a current supplied by a power transistor of the second half-bridge.

17. The method according to claim 12, further comprising supplying the sense voltage value to an analog-to-digital converter.

18. A circuit comprising a final stage that includes an H bridge comprising a first half-bridge and a second half-bridge; a read circuit configured to read a load current supplied by a class-D audio-amplifier to a load, the read circuit being configured to estimate the load current by reading a current at an output by the first or second half-bridge by measuring a drain-to-source voltage during an ON period of a low-side transistor of the first half-bridge or second half-bridge; and a sensing circuit comprising a differential amplifier and a low-pass filter, the sensing circuit configured to detect a first drain-to-source voltage from a transistor of the first half-bridge and a second drain-to-source voltage from a corresponding transistor of the second half-bridge, wherein the differential amplifier is configured to compute a difference between the first drain-to-source voltage and the second drain-to-source voltage, wherein the low-pass filter is configured to perform an averaging operation on the difference to obtain a sense voltage value.

19. The circuit according to claim 18, wherein sensing circuit comprises respective protection switches set between drain and source nodes of the transistors of the first half-bridge and of the second half-bridge and inputs of the differential amplifier.

20. The circuit according to claim 18, wherein the sensing circuit comprises a sensing network that includes an auxiliary transistor with a gate and a drain, wherein the gate of the auxiliary transistor is coupled to a gate of a respective transistor of the first half-bridge or second half-bridge and wherein a drain of the auxiliary transistor is coupled to a drain of the respective transistor of the first half-bridge or second half-bridge.

21. The circuit according to claim 20, wherein the sensing circuit further comprises a sense resistance between a source of the auxiliary transistor and ground of a corresponding half-bridge.

22. The circuit according to claim 21, wherein the sense resistance is sized so that the sense resistance has a higher value than a switch-on resistance of the auxiliary transistor, such as to render a sum of the switch-on resistance of the auxiliary transistor and of the sense resistance approximately equal to the sense resistance.

23. The circuit according to claim 22, wherein the sense resistance is sized so that the sense resistance has a lower value than a switch-on resistance of the auxiliary transistor, such as to render a sum of a switch-on resistance of the auxiliary transistor and of the sense resistance approximately equal to the switch-on resistance.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

[0031] Various embodiments will now be described, purely by way of example, with reference to the annexed drawings, wherein:

[0032] FIG. 1 illustrates a full-H bridge 11 of a final stage of a power audio amplification circuit;

[0033] FIG. 2 illustrates a circuit diagram;

[0034] FIG. 3 illustrates a time plot corresponding to the circuit diagram of FIG. 2;

[0035] FIG. 4 illustrates a reference power MOSFET;

[0036] FIG. 5 illustrates a plot that represents an output current and a corresponding sampled current;

[0037] FIG. 6 is a schematic illustration of a sensing circuit of the amplifier apparatus described;

[0038] FIGS. 7 and 8 show time plots of quantities in an amplifier apparatus that uses the sensing circuit of FIG. 6;

[0039] FIG. 9 is a schematic illustration of a variant embodiment of the sensing circuit of FIG. 6;

[0040] FIG. 10 shows time plots of quantities in an amplifier apparatus that uses the sensing circuit of FIG. 9;

[0041] FIG. 11 shows the sensing circuit of FIG. 9 according to a first operating configuration; and

[0042] FIG. 12 shows the sensing circuit of FIG. 9 according to a second operating configuration.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

[0043] In the ensuing description numerous specific details are provided in order to enable maximum understanding of the embodiments provided by way of example. The embodiments may be implemented with or without specific details, or else with other methods, components, materials, etc. In other circumstances, well-known structures, materials, or operations are not illustrated or described in detail so that various aspects of the embodiments will not be obscured. Reference, in the course of this description, to an embodiment or one embodiment means that a particular feature, structure, or characteristic described in connection with the embodiment is comprised in at least one embodiment. Hence, phrases such as in an embodiment, in one embodiment, or the like that may be present in various points of the present description do not necessarily refer to one and the same embodiment. Moreover, the particular features, structures, or characteristics may be combined in any convenient way in one or more embodiments.

[0044] The notation and references are provided herein merely for convenience of the reader and do not define the scope or meaning of the embodiments.

[0045] The idea underlying the solution described herein is to exploit the information of current of both of the half-bridges, and hence to detect a first drain-to-source voltage, preferably from a low-side transistor of the first half-bridge, and a second drain-to-source voltage from a corresponding low-side transistor of the second half-bridge. The sampling operation is replaced by an operation of averaging on a difference between the first detected drain-to-source voltage and the second detected drain-to-source voltage, obtained, for example, via a differential amplifier, to obtain a sense voltage value to be supplied to an analog-to-digital converter. The averaging operation is preferably carried out via a low-pass filtering of the aforethe difference between the signal present on the MOSFET, for example the low-side MOSFET, of the H bridge during its ON period, and the signal present on the corresponding transistor of the other half-bridge during its ON period.

[0046] Operation in the case of sensing performed on the low-side power transistors is described with reference to FIGS. 6, 7, and 8.

[0047] FIG. 6 shows a block diagram, where a sensing circuit 50 comprises a differential preamplifier 51, the input pins of which are connected, through switches 52P, a first one to the drain and a second one to the source of the low-side MOSFET 13b, and, through switches 52M, the first one to the drain and the second one to the source of the low-side MOSFET 23b of the half-bridge 22. Downstream of the switches 52P and 52M respective first and second sense voltages V.sub.SP and V.sub.SM are formed, which correspond to the drain-to-source voltages detected, supplied to the differential inputs of the preamplifier 51. The switches 52P connected to the first half-bridge 12 are driven by a protection signal SWPRP similar to that of FIG. 3, whereas the second switches 52M are driven via a protection signal SVVPRM that has ON states, which are complementary in the case of out-of-phase modulation (FIG. 7), whereas in the case of in-phase modulation (FIG. 8) anyway coincide with the ON interval of the respective low-side transistor (t.sub.aM for the second half-bridge). Hence, the protection switches 52P and 52M represent a circuit portion, or network, for detecting a first drain-to-source voltage V.sub.SP from a transistor 13b of the first half-bridge 12 and a second drain-to-source voltage V.sub.SM from a corresponding transistor 23b of the second half-bridge 22 via respective protection switches 52P, 52M set between the drain and source nodes of the transistors of the first half-bridge 12 and of the second half-bridge 22 and the preamplifier 51. This preamplifier 51 provides a module for computing a difference between the first detected drain-to-source voltage V.sub.SP and the second detected drain-to-source voltage V.sub.SM.

[0048] Connected to the output of the preamplifier 51 is a low-pass filter 53, at the output of which a resulting sense voltage V.sub.SENSE is formed. The low-pass filter 53 may also be integrated in the differential amplifier 51 and hence not be a block cascaded thereto. The low-pass filter 53 provides a module for performing a continuous-time averaging of the aforethe difference to obtain a sense voltage value V.sub.SENSE to be supplied to an analog-to-digital converter 54.

[0049] In this regard, illustrated in FIGS. 7 and 8 are plots representing, as a function of time t, the first output current I.sub.OUTP, the second output current I.sub.OUTM, the first sense voltage V.sub.SP, the second sense voltage V.sub.SM, and the difference between the first sense voltage and the second sense voltage, V.sub.SM-V.sub.SP. FIG. 7 shows the waveforms for out-of-phase modulation, and FIG. 8 shows the waveforms for in-phase modulation. As may be noted, the aforethe difference V.sub.sm-V.sub.SP, which corresponds, through the amplification ratio, to the resulting sense voltage V.sub.SENSE, determines, through the low-pass filter 53, an average sense signal with value V.sub.S, both for the in-phase case and for the out-of-phase case. Consequently, the sensing circuit 50 provides a continuous-time reading method that enables a greater accuracy in so far as the averaging operation eliminates the contribution of the ripple with zero mean value of the current in the finite inductance of the LC filter 14 or 24.

[0050] It may be noted that it is moreover possible to read the current even in the absence of the LC filter 14 or 24 in the circuit n of FIG. 1, a solution that eliminates the band limitations in so far as this modification causes the load current I.sub.LOAD to be equal to the first output current I.sub.OUTP and to the negative value of the second output current I.sub.OUTM not only at low frequencies.

[0051] Saturation of the reading system for high values of duty cycle is intrinsically eliminated.

[0052] Finally, it is possible to eliminate the circuits for generation of the signals for the sample and hold circuit (SWSH in FIG. 2).

[0053] FIG. 9 illustrates a preferred alternative embodiment for generation of the first sense voltage V.sub.SP and of the second sense voltage V.sub.SM to be supplied to the differential inputs of the preamplifier 51, which envisages replacement of each switch 52P and 52M with a respective sensing network 62 shown in FIG. 9. This sensing network 62 comprises an auxiliary transistor 64, of the same type as the transistor 13b or 23b, in general with an appropriate scale ratio that will guarantee thermal matching with the corresponding low-side transistor 13b or 23b. The scale ratio is chosen as a compromise between occupation of area and matching. The smaller the area ratio (or R.sub.DSON ratio) between the MOSFET 13b or 23b and the auxiliary transistor 62, the better, in general, the matching. With reference to what is referred to in what follows, this is valid when it is desired to operate in a partitioning mode, described hereinafter. In the case, instead, of operation in a sensing mode, once again described in what follows, a good coupling between the MOSFET 13b or 23b and the auxiliary transistor 62 is no longer necessary.

[0054] The gate and drain of the auxiliary transistor 64 are connected, respectively, to the gate and drain of the low-side transistor 13b. A sense resistance R.sub.SP, sized for example as described hereinafter to obtain two different operating modes, is set between the source of the auxiliary transistor 64 and the ground GNDP of the half-bridge 12. Between the sensing resistances R.sub.SP and R.sub.SM, respectively, of the two half-bridges and the preamplifier 51 a further amplification stage may possibly be inserted.

[0055] Thanks to the added sensing network 62, during the OFF step of the low-side transistor 13b the sense signals V.sub.SP and V.sub.SM have a value of o V instead of the value of the supply voltage VDD, as represented in FIG. 10, which shows the first output current I.sub.OUTP and the first sense voltage V.sub.SP as a function of time t, compatibly with the operation described with reference to FIGS. 7 and 8.

[0056] As further advantage, the reading electronics is automatically protected from the high voltage, and hence it is not necessary to resort to further protective circuitry, i.e., the switches 52P and 52M.

[0057] According to the mutual sizing between the switch-on resistance R.sub.DSONAUXP of the auxiliary transistor 64 and the sense resistance R.sub.SP, it may possible to choose two different operating modes, taking into account that for the circuit of FIG. 9 we have:

[00001]$R_{\mathrm{DSONLP}}.Math.<<R_{\mathrm{DSONAUXP}}+R_{\mathrm{SP}}$$V_{\mathrm{SP}}=-I_{\mathrm{OUTP}}.Math.\left(R_{\mathrm{DSONLP}}.Math.R_{\mathrm{SP}}\right)/\left(R_{\mathrm{DSONLP}}+R_{\mathrm{DSONAUXP}}+R_{\mathrm{SP}}\right)-I_{\mathrm{OUTP}}.Math.\left(R_{\mathrm{DSONLP}}.Math.R_{\mathrm{SP}}\right)/\left(R_{\mathrm{DSONAUXP}}+R_{\mathrm{SP}}\right)$

[0058] In a first, sensing, mode the sense resistance R.sub.SP is sized so that it has a value much higher than the switch-on resistance R.sub.DSONAUXP of the auxiliary transistor 64, in particular such as to render the sum of the switch-on resistance R.sub.DSONAUXP of the auxiliary transistor 64 and of the sense resistance R.sub.SP approximately equal to the sense resistance R.sub.SP in the relation appearing above that expresses the sense voltage V.sub.SP. In this way, the first sense voltage V.sub.SP is approximately equal to I.sub.OUTP.Math.R.sub.DSONLP, and sensing of the drain-to-source voltage V.sub.DSLP of the low-side transistor 13b is carried out.

[0059] In a second, partitioning, mode, the sense resistance R.sub.SP is sized so that it has a value much lower than the switch-on resistance R.sub.DSONAUXP, in particular such as to render the sum of the switch-on resistance R.sub.DSONAUXP of the auxiliary transistor 64 and of the sense resistance R.sub.SP approximately equal to the switch-on resistance R.sub.DSONAUXP in the relation appearing above that expresses the sense voltage V.sub.SP. Hence, the first sense voltage V.sub.SP is approximately equal to I.sub.OUTP9 (R.sub.DSONLP/R.sub.DSONAUXP).Math.R.sub.SP), and thus the sensing operation is carried out on a partitioning of known value of the first output current I.sub.OUTP.

[0060] Of course, similar arguments apply in a dual way to the low-side transistor 23b of the second half-bridge 22.

[0061] In greater detail, with reference to the diagram of FIG. 11, in the first, sensing, mode the drain-to-source voltage V.sub.DSLP of the low-side transistor 13b is measured.

[0062] To obtain the information of current, as described with reference to FIG. 4, it is necessary to compare the measured drain-to-source voltage V.sub.DSLP of the low-side transistor 13b with the reference voltage V.sub.DSREF resulting from a known current, the reference current I.sub.REF, that flows in the reference MOS transistor 13c coupled to the low-side transistor 13b according to the relation:

I.sub.OUT(I.sub.REF.Math.(R.sub.DSREF/R.sub.DSONLP)).Math.(V.sub.SP/V.sub.DSREF)

[0063] where (I.sub.REF.Math.(R.sub.DSREF/R.sub.DSONLP)) is a term the values of which are known design values and (V.sub.SP/V.sub.DSREF) is a term the values of which are measured. More specifically, of the term (V.sub.SP/V.sub.DSREF) only the value of the ratio is measured, for example using an ADC, where the full-scale is regulated by the reference drain-to-source voltage V.sub.DSREF, and the voltage to be converted is the sense voltage V.sub.SP, so that the output of the converter depends in actual fact only upon the ratio between the two values.

[0064] The aforethe first, sensing, mode presents the following advantages:

[0065] no thermal matching between the low-side transistor 13b (or 23b) and the auxiliary transistor 64 is required;

[0066] the layout is simplified;

[0067] the signal to be amplified is the maximum one available in so far as the drain-to-source voltage V.sub.DSLP of the low-side transistor is not partitioned; and

[0068] the reading electronics is simplified.

[0069] Since the non-linearity of the drain-to-source voltage V.sub.DSLP of the low-side transistor as a function of current causes a non-linearity in the reading, the first, sensing, mode is preferable in the cases of relatively low drain-to-source voltages, in which the effect of non-linearity is negligible.

[0070] With reference to the diagram of FIG. 12, in the second, partitioning, mode a partitioning of the current that flows in the low-side transistor 13b is measured.

[0071] To obtain the information of current, it is necessary to compare the sense voltage V.sub.SP with a reference constituted by the known current I.sub.REF that flows in a reference resistance 17 of value R.sub.REF coupled to the sense resistance R.sub.SP, i.e., connected between the generator 16 of reference current I.sub.REF and ground GNDP, on which there is a reference voltage drop V.sub.REF, according to the relation:

I.sub.OUT(I.sub.REF.Math.(R.sub.REF/R.sub.SP).Math.(R.sub.DSONAUXP/R.sub.DSONLP)).Math.(V.sub.SP/V.sub.REF),

[0072] where (I.sub.REF.Math.(R.sub.REF/R.sub.SP).Math.(R.sub.DSONAUXP/R.sub.DSONLP)) is a term the values of which are known design values, and (V.sub.SP/V.sub.REF) is a term the values of which, or the value of their ratio, are/is measured. It should be noted that to obtain a measurement of I.sub.OUT independent of temperature and process, the measurement must depend upon ratios of resistances of the same type, so that the process or temperature variations cancel out, the measurements amounting only to ratios of areas. Hence, R.sub.REF/R.sub.SP is a ratio between values of two resistors, R.sub.DSONAUXP/R.sub.DSONLP is a ratio between values of two MOSFETs in the ohmic region (the sense resistance R.sub.SP must be much lower than the switch-on resistance R.sub.DSONLP of the low-side transistor 13b so that the power MOSFET 13b and the auxiliary MOS transistor 62 work as far as possible in the same condition, i.e., the same gate-to-source voltage V.sub.Gs and the same drain-to-source voltage V.sub.SP).

[0073] This second, partitioning, mode presents, as compared to the first mode, the advantage that the effect of the non-linearity of the drain-to-source voltage of the low-side transistor is limited, and hence the measurement can be very linear.

[0074] On the other hand, accurate thermal and electrical matching is required between the low-side transistor, for example 13b, and the auxiliary transistor 64, so as to guarantee a well-controlled partitioning. For this reason, the layout of the power transistors is more complex. There is also required good thermal and electrical matching between the sense resistors R.sub.SP and R.sub.REF. The sense resistance R.sub.SP may be of a very small value so as to be negligible as compared to the resistance of the auxiliary switch, this possibly resulting in a complex layout of the sense resistance, whilst the signal to be amplified may be very small, thus rendering the reading electronics more critical.

[0075] The second, partitioning, mode is hence preferable in the cases where the effect of the non-linearity of the drain-to-source voltage is not acceptable. Instead, the first, sensing, mode in any case presents advantages in terms of complexity of the layout and simplicity of the reading circuit.

[0076] Hence, the advantages of the solution described emerge clearly from the foregoing description.

[0077] The class-D audio amplifier comprising a circuit for reading a current supplied by the amplifier to the load described herein advantageously provides a continuous-time reading method that enables a greater accuracy in so far as the operation of averaging eliminates the contribution of the zero-mean ripple of the current in the finite inductance of the LC filter.

[0078] Moreover, this amplifier advantageously enables reading of the current even in the absence of the LC filter, thus eliminating the band limitations.

[0079] Furthermore, this amplifier advantageously enables intrinsic elimination of the saturation of the reading system for high values of duty cycle.

[0080] Finally, this amplifier advantageously enables elimination of the circuits for generation of signals for the sample and hold circuit.

[0081] In addition, the use of a sensing network in the amplifier described enables automatic protection of the reading electronics from high voltage.

[0082] Moreover, advantageously, via simple sizing of a sense resistance, the sensing network is readily configurable for use in sensing mode with simpler layout and reading electronics, or in partitioning mode, which is less sensitive to non-linear behaviours. In particular, dependence of the measurement upon the non-linearity of the drain-to-source current-voltage characteristic of the MOSFET of the half-bridge, acquired on which is the output current, is eliminated.

[0083] Of course, without prejudice to the principle of the invention, the details and the embodiments may vary, even considerably, with respect to what has been described herein purely by way of example, without thereby departing from the sphere of protection, which is defined in the annexed claims.

[0084] The class-D audio amplifier apparatus comprising a circuit for reading a load current supplied by the amplifier apparatus to a load described herein may envisage reading, for estimating the load current, a current supplied at output by a half-bridge by measuring a drain-to-source voltage (V.sub.DSLP) during an ON period of the high-side (n-channel or p-channel) power MOSFETs, instead of carrying out the measurement on the low-side ones, even though it is in general more convenient to operate with a circuit referenced to ground, rather than with a circuit referenced to the supply voltage. In the case of high-side p-channel transistors a complementary equivalent solution is used, with the sources connected to the supply voltage.