Abstract
The invention relates to a balanced antenna system comprising a radiator connected via a matching circuit to a balun. In certain embodiments, the radiator comprises a first radiating element and a second radiating element and the matching circuit comprises a first impedance-matching circuit connected to the first radiating element and a second impedance-matching circuit connected to the second radiating element. The first and second matching circuits may be identical and are connected through the balun to a single port. To minimize the component count, the design of the matching circuit and balun is co-optimized.
Claims
1. A balanced antenna system comprising a radiator connected to an integrated matching circuit and balun, wherein i) the radiator comprises a first radiating element and a second radiating element each having a respective feed line; (ii) the matching circuit comprises a first impedance-matching circuit having an input and an output and a second impedance-matching circuit having an input and an output; (iii) the balun comprises first and second inputs for input of first and second balanced signals, circuitry to convert the first and second balanced signals to a single unbalanced signal, and a single output for output of the unbalanced signal; (iv) the feed line of the first radiating element is connected to the input of the first impedance-matching circuit and the feed line of the second radiating element is connected to the input of the second impedance-matching circuit; (v) the output of the first impedance-matching circuit is connected to the first input of the balun and the output of the second impedance-matching circuit is connected to the second input of the balun; and (vi) the first and second impedance-matching circuits and the balun are integrated and produce a phase difference of substantially 180 (degrees) between the feed lines of the first and second radiating elements across an operating bandwidth of the antenna system.
2. The balanced antenna system according to claim 1 wherein the balun is configured to convert unbalanced signals to balanced signals by cancelling or choking an outside current.
3. The balanced antenna system according to claim 1 wherein the balun comprises one of a current balun, a coax balun or a sleeve balun.
4. The balanced antenna system according to claim 1 wherein the balun comprises a wideband LC balun, configured for impedance transformation.
5. The balanced antenna system according to claim 1 wherein the balun comprises a first filter and a second filter.
6. The balanced antenna system according to claim 5 wherein a first impedance-matching circuit is provided between the first filter and a first radiating element and a second impedance-matching circuit is provided between the second filter and a second radiating element.
7. The balanced antenna system according to claim 6 wherein the balun comprises a high pass filter comprising the first filter, a low pass filter comprising the second filter and a T-junction.
8. The balanced antenna system according to claim 6 wherein the balun comprises a high pass filter comprising the first filter and a band pass filter comprising the second filter, connected in parallel.
9. The balanced antenna system according to claim 5 wherein the balun comprises a high pass filter comprising the first filter, a low pass filter constituting comprising the second filter and a T-junction.
10. The balanced antenna system according to claim 9 wherein the high pass filter and/or the low pass filter each comprise one or more than one inductor or capacitor.
11. The balanced antenna system according to claim 10 wherein the high pass filter comprises one or more than one capacitor connected in series and no, one or more than one inductor connected in parallel and the low pass filter comprises one or more than one inductor connected in series and no, one or more than one capacitor connected in parallel.
12. The balanced antenna system according to claim 9 wherein a second high pass filter and/or a second low pass filter is provided and at least one switch is provided to enable the second high pass filter and/or the second low pass filter to be activated in place of the respective high pass filter and/or low pass filter.
13. The balanced antenna system according to claim 12 wherein the second high pass filter and/or the second low pass filter comprises an L-C circuit and the second high pass filter comprises three capacitors connected in series and two inductors connected in parallel and the second low pass filter comprises three inductors connected in series and two capacitors connected in parallel.
14. The balanced antenna system according to claim 5 wherein the balun comprises a high pass filter comprising the first filter and a band pass filter comprising the second filter, connected in parallel.
15. The balanced antenna system according to claim 1 wherein the first and/or second impedance-matching circuits are reconfigurable to enable the respective first and/or second radiating elements to be tuned to different frequencies.
16. The balanced antenna system according to claim 15 wherein the first and/or second impedance-matching circuits comprise an L-C circuit comprising a variable capacitor.
17. The balanced antenna system according to claim 15 wherein the first and/or second impedance-matching circuit comprises a first inductor, a capacitor and a second inductor.
18. The balanced antenna system according to claim 17 wherein the first inductor is connected in parallel with the capacitor and the second inductor is connected in series with the capacitor.
19. The balanced antenna system according to claim 18 wherein the first inductor is connected to a ground plane and the capacitor is tunable.
20. The balanced antenna system according to claim 15 wherein at least one alternative component is provided for inclusion in the first and/or second impedance-matching circuit and at least one switch is provided to enable the at least one alternative component to be activated in place of another component.
21. The balanced antenna system according to claim 1 wherein the first radiating element is constituted by a first strip which is substantially U-shaped and is provided on a first side of a substrate at a first end thereof.
22. The balanced antenna system according to claim 21 wherein the U-shaped strip is located in one half of a first end portion of the substrate and is orientated such that its open end faces inwardly towards the central region of the first end portion.
23. The balanced antenna system according to claim 22 wherein a feed line is provided at a start of the U-shaped strip closest to the centre of the substrate and which extends along the length of the substrate.
24. The balanced antenna system according to claim 22 wherein the second radiating element is substantially similar to the first radiating element and is also provided on the first side of the substrate but orientated in an adjacent half of the first end portion of the substrate, opposite to the first radiating element, such that an open end of the second strip faces the open end of the first strip.
25. The balanced antenna system according to claim 24 wherein a gap is provided between the respective feed lines of the first and second radiating elements and between the respective ends of the first and second strips.
26. The balanced antenna system according to claim 21 wherein a ground plane is provided on a second side of the substrate, opposite the first side.
27. An antenna structure for MIMO applications comprising at least one balanced antenna system according to claim 1 and at least one further antenna.
28. The antenna structure according to claim 27 wherein the at least one further antenna comprises a balanced or unbalanced antenna and is reconfigurable.
29. The antenna structure according to claim 27 wherein a first antenna is located at a first end of the structure and a second antenna is located at a second end of the structure.
30. The antenna structure according to claim 27 wherein the at least one further antenna comprises a reconfigurable antenna comprising two or more mutually coupled radiating elements and two or more impedance-matching circuits configured for independent tuning of the frequency band of each radiating element, wherein each radiating element is arranged for selective operation in each of the following states: a driven state, a floating state and a ground state.
31. The antenna structure according to claim 27 wherein the balanced antenna system is isolated from the further antenna by provision of a slot in a ground plane of the antenna structure.
32. The balanced antenna system according to claim 1, wherein the integrated matching circuit and balun comprises at least one variable impedance component actively reconfiguring the integrated matching circuit and balun for different frequencies across the operating bandwidth of the antenna system.
Description
BRIEF DESCRIPTION OF THE DRAWINGS
(1) Certain embodiments of the present invention will now be described with reference to the accompanying drawings in which:
(2) FIG. 1A shows a front view of a balanced antenna according to a first embodiment of the present invention;
(3) FIG. 1B shows a rear view of the balanced antenna of FIG. 1A;
(4) FIG. 2 shows a circuit diagram for a balun configuration for use with the balanced antenna shown in FIGS. 1A and 1B;
(5) FIG. 3 shows a schematic representation of a balanced antenna system according to an embodiment of the present invention;
(6) FIG. 4 shows a circuit diagram for an impedance-matching circuit configuration for an embodiment of the present invention;
(7) FIGS. 5A and 5B show a circuit diagram for a balanced antenna system according to an embodiment of the present invention, including multiple balun configurations and multiple impedance-matching configurations;
(8) FIG. 6 shows a graph of return loss against frequency for a first configuration of the circuit shown in FIGS. 5A and 5B, over a range of varactor capacitance from 1.11 pF to 10 pF;
(9) FIG. 7 shows a graph of return loss against frequency for a second configuration of the circuit shown in FIGS. 5A and 5B, over a range of varactor capacitance from 0.21 pF to 10 pF;
(10) FIG. 8 shows a graph of return loss against frequency for a third configuration of the circuit shown in FIGS. 5A and 5B, over a range of varactor capacitance from 0.38 pF to 10 pF;
(11) FIG. 9A shows a front view of a combined MIMO antenna system comprising the balanced antenna of FIGS. 1A and 1B with a chassis antenna;
(12) FIG. 9B shows a rear view of the combined MIMO antenna system of FIG. 9A;
(13) FIG. 10 shows a graph of S parameters for the MIMO antenna system of FIGS. 9A and 9B when connected to the circuit shown in FIGS. 5A and 5B;
(14) FIG. 11A illustrates the current distribution through the MIMO antenna system of FIGS. 9A and 9B when the balanced antenna is driven;
(15) FIG. 11B illustrates the current distribution through the MIMO antenna system of FIGS. 9A and 9B when the chassis antenna is driven;
(16) FIG. 12A illustrates the average current distribution through the MIMO antenna system of FIGS. 9A and 9B when the balanced antenna is driven;
(17) FIG. 12B illustrates the average current distribution through the MIMO antenna system of FIGS. 9A and 9B when the chassis antenna is driven;
(18) FIG. 13A illustrates the far field current distribution of the MIMO antenna system of FIGS. 9A and 9B when the balanced antenna is driven at 700 MHz;
(19) FIG. 13B illustrates the far field current distribution of the MIMO antenna system of FIGS. 9A and 9B when the chassis antenna is driven at 700 MHz;
(20) FIG. 14A shows a front view of another combined MIMO antenna system comprising the balanced antenna of FIGS. 1A and 1B with a side-mounted chassis antenna;
(21) FIG. 14B shows a rear view of the combined MIMO antenna system shown in FIG. 14A;
(22) FIG. 15 shows a graph of S parameters for the MIMO antenna system of FIGS. 14A and 14B when connected to the circuit shown in FIGS. 5A and 5B;
(23) FIG. 16A shows a front view of another combined MIMO antenna system comprising the balanced antenna of FIGS. 1A and 1B with a second, side-mounted balanced antenna;
(24) FIG. 16B shows a rear view of the combined MIMO antenna system shown in FIG. 16A;
(25) FIG. 17 shows a graph of S parameters for the MIMO antenna system of FIGS. 16A and 16B when connected to the circuit shown in FIGS. 5A and 5B;
(26) FIG. 18A shows a front view of another combined MIMO antenna system comprising two balanced antennas with a side-mounted chassis antenna;
(27) FIG. 18B shows a rear view of the combined MIMO antenna system shown in FIG. 18A;
(28) FIG. 19 shows a graph of S parameters for the MIMO antenna system of FIGS. 18A and 18B when connected to the circuit shown in FIGS. 5A and 5B;
(29) FIG. 20 shows a circuit diagram for an alternative balun configuration for use in embodiments of the present invention;
(30) FIG. 21A shows a front view of a balanced antenna according to a further embodiment of the present invention;
(31) FIG. 21B shows a back view of the balanced antenna of FIG. 21A;
(32) FIG. 21C shows an enlarged plan view of the radiating element shown in FIG. 21A;
(33) FIG. 22 shows a balun and matching circuit arrangement for the antenna of FIGS. 21A through 21C;
(34) FIG. 23 shows a graph of phase against frequency for the outputs Z.sub.b1 and Z.sub.b2 of the circuit of FIG. 22;
(35) FIG. 24 shows simulated reflection coefficients against frequency for when the varactors in the matching circuit of FIG. 22 are varied from 10 pF to 0.2 pF;
(36) FIG. 25A shows a front perspective view plus an enlarged detail view of a balanced antenna according to another embodiment of the present invention;
(37) FIG. 25B shows a rear view of the balanced antenna of FIG. 25A;
(38) FIG. 25C shows a front plan view of the radiating elements of the balanced antenna of FIGS. 25A and 25B;
(39) FIG. 26 shows a circuit diagram comprising a balun and matching circuit arrangement for the antenna of FIGS. 25A through 25C;
(40) FIG. 27 shows simulated reflection coefficients against frequency for when the varactors in the circuit of FIG. 26 are varied from 10 pF to 0.1 pF;
(41) FIG. 28 shows an alternative circuit diagram comprising a balun and matching circuit arrangement for the antenna of FIGS. 21A through 21C;
(42) FIG. 29 shows simulated reflection coefficients against frequency for when the varactors in the circuit of FIG. 28 are varied from 10 pF to 0.1 pF;
(43) FIG. 30 shows a front and rear view of a further balanced antenna according to an embodiment of the present invention;
(44) FIG. 31 shows a circuit diagram comprising a balun and matching circuit arrangement for the antenna of FIG. 30;
(45) FIG. 32 shows simulated reflection coefficients against frequency for when the varactors in the circuit of FIG. 31 are varied from 10 pF to 0.28 pF;
(46) FIG. 33A shows a front perspective view of another balanced antenna according an embodiment of the present invention;
(47) FIG. 33B shows a rear perspective view of the balanced antenna of FIG. 33A;
(48) FIG. 34 shows circuit diagrams for driving each of the radiating elements in the antenna of FIGS. 33A and 33B;
(49) FIG. 35 shows measured reflection coefficients against frequency for when the varactors in the circuit of FIG. 34 are varied from approximately 15.4 pF to 0.4 pF; and
(50) FIG. 36 shows a view similar to that of FIG. 33B but wherein the ground plane has been modified to include a slot.
DETAILED DESCRIPTION OF CERTAIN EMBODIMENTS
(51) With reference to FIGS. 1A and 1B, there is illustrated a balanced antenna system 10 according to a first embodiment of the present invention. The balanced antenna system 10 is reconfigurable, as will be described in more detail below, and is designed for use in a portable product such as a mobile phone, laptop or PDA.
(52) The balanced antenna system 10 is provided on a microwave substrate 12 (e.g. a printed circuit board, PCB) having a surface area of approximately 11640 mm.sup.2 and a thickness of approximately 1.15 mm so that the system can easily be accommodated in a conventional mobile phone.
(53) As illustrated in FIG. 1A, a first radiating element 14 is provided on a first side 16 of the substrate 12, at a first end portion 18 thereof. The first radiating element 14 is formed from a substantially U-shaped first strip layer 20 which is located in one half of the first end portion 18 of the substrate 12 and is orientated such that its open end 22 faces inwardly towards the central region of the first end portion 18. A short feed line 24 is provided at a start of the first strip 20 closest to the centre of the substrate 12 and extends along the length of the substrate 12.
(54) A second radiating element 26, which is substantially similar to the first radiating element 14, is also provided on the first side 16 of the substrate 12 and is located in an adjacent half of the first end portion 18 of the substrate 12. The second radiating element 26 is therefore formed from a substantially U-shaped second strip layer 28 which is also orientated such that its open end 30 faces inwardly towards the central region of the first end portion 18. Thus, the second radiating element 26 is orientated in an opposite direction to the first radiating element 14. A short feed tine 32 is again provided at a start of the second strip 28 closest to the centre of the substrate 12 and extends along the length of the substrate 12.
(55) A gap 34 is provided between the respective feed lines 24, 32 of the first and second radiating elements 14, 26 and between the respective ends 36 of the first and second strips 20, 28. Accordingly, the first and second radiating elements 14, 26 form a dipole antenna 37. In the embodiment shown in FIG. 1A, the first and second radiating elements 14, 26 together extend over an area of approximately 4010 mm.sup.2.
(56) As shown in FIG. 1B, a ground plane 38 is provided on a second side 40 of the substrate 12, opposite to the first side 16. The ground plane 38 is substantially rectangular and occupies substantially the whole of the substrate 12 surface from a second end 42 thereof (opposite to the first end portion 18) to a position substantially opposite the feed lines 24, 32. The ground plane 38 has a size of approximately 10040 mm.sup.2.
(57) The balanced antenna system 10 also includes a balun and two matching circuits which are connected to the first and second radiating elements 14, 26 and which are not shown in FIGS. 1A and 1B for reasons of clarity but which would be provided on the first side 16 of the substrate 12, opposite to the ground plane 40.
(58) An example of a suitable balun 50 is shown in FIG. 2 by way of an LC circuit diagram. This particular balun 50 has a wideband configuration and is substantially as described by Iizuka and Watanabe in published Japanese patent application number 2005-198167. Thus, the balun 50 comprises a first (unbalanced) port Z.sub.u forming an input to a T-junction 52 which is configured to split an electrical signal received at the first port Z.sub.u into a high path 54 and a low path 56. The high path 54 is arranged to feed into a first high pass filter (HPF) and the low path is arranged to feed into a second low pass filter (LPF).
(59) The high pass filter (HPF) is constructed from an L-C circuit having three capacitors connected in series C.sub.H1, C.sub.H2, C.sub.H1 and two inductors L.sub.H1, L.sub.H1 connected in parallel from respective branches provided between the capacitors. Each of the inductors L.sub.H1 is connected to the ground plane 40 and the output from the capacitors C.sub.H1, C.sub.H2, C.sub.H1 constitutes an impedance Z.sub.bH.
(60) The low pass filter (LPF) is constructed from an L-C circuit having three inductors connected in series L.sub.L1, L.sub.L2, L.sub.L1 and two capacitors C.sub.L1, C.sub.L1 connected in parallel from respective branches provided between the inductors. Each of the capacitors C.sub.L1 is connected to the ground plane 40 and the output from the inductors L.sub.L1, L.sub.L2, L.sub.L1 constitutes an impedance Z.sub.bL. Together, Z.sub.bH and Z.sub.bL form a balanced output Z.sub.b.
(61) As illustrated in FIG. 3, the balanced antenna system 10 according to the present embodiment of the invention comprises the balun 50 shown in FIG. 2, used to feed the balanced dipole antenna 37 shown in FIG. 1A, with an impedance matching circuit 60 provided therebetween. It will be noted that in this illustration, an input Port 1 is shown which feeds into the first (unbalanced) port Z.sub.u of the balun 50 to drive the balanced antenna system 10. More specifically, the high pass filter with impedance Z.sub.bH from the balun 50 is connected to the matching circuit 60 and to Feed 1 (i.e. feed line 24 of FIG. 1A) of the dipole antenna 37 and the low pass filter with impedance Z.sub.bL from the balun 50 is connected to the matching circuit 60 and to Feed 2 (i.e. feed line 32 of FIG. 1A) of the dipole antenna 37.
(62) As will be explained in more detail below, by incorporating the matching circuit 60 between the dipole antenna 37 and the balun 50, the system can be made reconfigurable and can be used to provide a wide tuning range of from 470 MHz to 2200 MHz, which can cover DVB-H, all GSM and UMTS2100 frequency bands.
(63) FIG. 4 shows a circuit diagram for a particular impedance-matching circuit 60 which can be used in an embodiment of the present invention. It will be noted that in this illustration, the paths for the high pass filter with impedance Z.sub.bH and the low pass filter with impedance Z.sub.bL from the balun 50 to Feed 1 and Feed 2 respectively, have been separated out such that the impedance-matching circuit 60 is split into a first matching circuit 62 and a second matching circuit 64. In this particular embodiment, the first and second matching circuits 62, 64 are identical and are configured to provide impedance transformation to each leg of the balanced dipole antenna 37.
(64) More specifically, each of the first and second matching circuits 62, 64 comprise an inductor L.sub.2 connected in parallel to the ground plane 40 and a capacitor C.sub.1/C.sub.2 and inductor L.sub.1 connected in series. The capacitors C.sub.1/C.sub.2 are variable so as to allow the impedance of the first and second matching circuits 62, 64 to be adjusted to tune the antenna 37 over a range of frequencies.
(65) While the embodiment shown in FIG. 4 can enable tuning over a certain range of frequencies it has been discovered that a greater tuning range can be achieved by including multiple balun configurations and multiple impedance-matching configurations into an integrated tuning circuit 70, such as illustrated in FIGS. 5A and 5B. In this embodiment, input Port 1 (unbalanced feed-line) is provided at the left-hand side of the circuit 70 and the unbalanced feed-line is converted to two balanced feed-lines via the balun configuration 72. The high pass filter is connected to a first impedance matching circuit 74 and to the balanced dipole antenna 37 via Feed 1, and the low pass filter is connected to a second impedance matching circuit 76 and to the balanced dipole antenna 37 via Feed 2.
(66) The balun configuration 72 comprises a first high pass filter 78 and a second high pass filter 80 which are identical in construction to the high pass filter described above in relation to FIG. 2 but which have different values for each of the capacitors and inductors. A first (signal pole double throw) switch 82 is provided to select one of the first or second high pass filters 78, 80 to be employed at any one time. A first low pass filter 84 and a second low pass filter 86 are also provided which are identical in construction to the low pass filter described above in relation to FIG. 2 but which again have different values for each of the capacitors and inductors. A second (signal pole double throw) switch 88 is provided to select one of the first or second low pass filters 84, 86 to be employed at any one time.
(67) The first impedance matching circuit 74 is of the form described above in relation to FIG. 4 but includes three different inductors (L7, L8, L9) which can be selected as the inductor in parallel (L.sub.2) and three further different inductors (L1, L2, L3) which can be selected as the inductor in series (L.sub.1). Various switches are provided in order to activate the desired combination of components for the first impedance matching circuit 74. As above, a tunable capacitor C.sub.1 (i.e. varactor) is provided between the two sets of inductors. The tunable capacitor C.sub.1 can be tuned from 0.2 pF to 10 pF.
(68) Similarly, the second impedance matching circuit 76 is of the form described above in relation to FIG. 4 but also includes three different inductors (L14, L15, L16) which can be selected as the inductor in parallel (L.sub.2) and three further different inductors (L4, L5, L6) which can be selected as the inductor in series (L.sub.1). Various switches are provided in order to activate the desired combination of components for the second impedance matching circuit 76. As before, a tunable capacitor C.sub.2 (i.e. varactor) is provided between the two sets of inductors. The tunable capacitor C.sub.2 can be tuned from 0.2 pF to 10 pF.
(69) The circuit shown in FIGS. 5A and 5B was designed using a simulation tool (CST Microwave Studio) in which the antenna structure of FIGS. 1A and 1B was simulated using the transient solver to produce a 1-Port S-Parameter file representing the antenna response and this was used as a starting point for designing the matching networks. The values of the components within the first and second matching circuits 74, 76 were calculated using standard formulas available within the literature. Microwave Office (an RF/microwave design platform available from AWR Corporation) was then used to adjust the value of each inductor in order to optimize the return loss performance of the antenna. The capacitors C.sub.1 and C.sub.2 were fixed to 10 pF respectively during this phase of the design process. It is noted that the same simulation tool was used for all simulations described herein.
(70) The Applicants discovered during these simulations that, in order to obtain a single resonance across the whole of the desired operating band (470 MHz to 2200 MHz), with at least 6 dB return loss, three different configurations for each of the matching networks 74, 76 were required along with the two different configurations for the balun 72. These different configurations have been integrated into the circuit shown in FIGS. 5A and 5B but could be provided as three separate circuits if desired.
(71) Table 2 below lists the required logic states for the switches shown in FIG. 5 in order to produce the required spectrum coverage. In this embodiment, a single pole double throw switch, such as switch 82 in FIGS. 5A and 5B, is in an on position, represented by 1 and an off position, represented by 0. For a double pole double throw switch as illustrated in FIGS. 5A and 5B, a 0 represents the situation where nodes labelled 1 and 2 are connected and nodes labelled 4 and 5 are connected while a 1 represents the situation where nodes labelled 1 and 3 are connected and nodes labelled 4 and 6 are respectively connected. As illustrated in FIG. 5, each of the switches is shown in a default position.
(72) TABLE-US-00002 TABLE 2 Operating logic states Modes X Y Z Operating band OUTPUT (MHz) A 1 1 0 470-640 B 1 0 0 630-1520 C 0 0 1 1500-2200
(73) Thus, the three states described in Table 2 and denoted as Mode A, B, and C relate to the three relatively narrow bands of operation of the antenna.
(74) In Mode A operation, it is possible to move the resonant frequency from 470 MHz to 640 MHz by varying the varactors C.sub.1 and C.sub.2 together from 10 pF to 1.11 pF as illustrated in FIG. 6.
(75) FIG. 7 illustrates Mode B operation in which the resonant frequency can be tuned from 630 MHz to 1520 MHz by varying the varactors C.sub.1 and C.sub.2 together from 10 pF to 0.21 pF.
(76) Similarly, FIG. 8 illustrates Mode C operation when the varactors C.sub.1 and C.sub.2 together are varied from 10 pF to 0.38 pF and the resonant frequency is moved from 1500 MHz to 2200 MHz.
(77) Thus, the simulation results using ideal components show that the present antenna structure has wide tuning range of 1730 MHz. Accordingly, the resonant frequency of the antenna can be tuned to cover DVB-H, GSM710, GSM850, GSM900, GPS1575, GSM1800, PCS 1900 and UMTS2100 bands.
(78) It will be noted that although the varactors C.sub.1 and C.sub.2 have each been varied together in the above examples, in other embodiments, tuning may achieved by setting each varactor to a different value.
(79) An antenna structure 90 suitable for Multiple-Input-Multiple-Output (MIMO) applications is illustrated in FIGS. 9A and 9B in accordance with another embodiment of the present invention. The antenna structure 90 comprises the balanced antenna system 10 described above in combination with a two-port chassis antenna 100. As explained above MIMO devices utilise multiple uncorrelated channels/antennas to increase the capacity of a communication link without the need of additional spectrum or transmitter power. The uncorrelated antennas should be terminated at an optimal location to induce a required (or best) antenna-to-antenna isolation. The best isolation values are often achieved when the antennas are spaced by the largest available distance (e.g. one at the top edge of the substrate 12 and the other at the bottom edge). In this particular embodiment, the antennas are separated along the length of the substrate 12 by a distance of approximately 100 mm.
(80) In the present application, many examples of MIMO systems according to the present invention are described. For the first three cases, it will be noted that the MIMO system comprises two separate antennas. While in the forth case, the MIMO system comprises three separate antennas.
(81) In the MIMO system shown in FIGS. 9A and 9B, the two-port chassis antenna 100 is of the type described in detail in GB0918477.1 and comprises a pair of non-resonant coupling elements 112, 114, that are capable of simultaneous dual-band operation. The coupling elements 112, 114 occupy a relatively small volumetric space of 4057 mm.sup.3 and are located on the top of a chassis 116 on the same side as the second side 40 of the substrate 12, while the dipole antenna 37 of the balanced antenna system 10 is located on the opposite first side 16.
(82) More specifically, the two coupling elements 112, 114 are mounted in close proximity to each other and are driven over the ground plane 38. The first coupling element 112 is constituted by an L-shaped metal patch having a planar portion which constitutes the chassis 116, parallel to the ground plane 38, and an orthogonal portion 118, orthogonal to the ground plane 38. As described above, the planar portion 116 is provided on the opposite side of the substrate from the ground plane 38 and is laterally spaced therefrom. The orthogonal portion 118 extends from an edge of the planar portion 116 furthest from the ground plane 38 such that the orthogonal portion 118 is spaced from the ground plane 38 by a so-called first gap 120. In this particular embodiment the first gap 120 is less that 10 mm.
(83) The second coupling element 114 is also constituted by a metal patch which, in this case, forms a planar rectangle. The second radiating element 114 is also orientated orthogonally to the ground plane 38 and is located within the first gap 120. Thus, the second radiating element 114 is effectively enclosed on two adjacent sides by the L-shaped first coupling element 12. In the embodiment shown, the second coupling element 114 is approximately half the length of the first coupling element 112 and is slightly inset from the edge of the first coupling element 112. The distance between the ground plane 38 and the second coupling element 114 forms a so-called second gap 122. The distance between the second coupling element 114 and the orthogonal portion 118 of the first coupling element 12 determines the amount of mutual coupling therebetween. This distance is therefore referred to as the mutual gap 124.
(84) Although not shown, each radiating element 112, 114 is connected, respectively, to a first and second control module via a second and third feed port 126, 128. The second feed port 126 (Port 2) extends between the orthogonal portion 118 of the first coupling element 112 and a first control module (not shown), and is located approximately one third of the distance along the length of the first coupling element 112. The third feed port 128 (Port 3) is located adjacent to the second feed port 126 and connects to an adjacent second control module (not shown). As described in GB0918477.1 each coupling element 112, 114 can be selectively driven independently, allowed to float, or tied to the ground state through operation of the respective control modules. Thus, it is possible to dynamically tune the operating frequency of each coupling element 112, 114 by selecting different modes of operation in a similar manner to that described above in relation to the tuning of the balanced dipole antenna 37.
(85) It is possible to operate the MIMO antenna structure 90 by driving the balanced antenna system 10 through Port 1 and using Port 2 and Port 3 to operate each of the coupling elements 112, 114 of the chassis antenna 100. For purposes of demonstration, Port 1 was connected to a circuit similar to that shown in FIGS. 5A and 5B and only Port 2 was activated in connection with the chassis antenna 100. More specifically, Port 2 was connected to a circuit of the type described in GB0918477.1 and Port 3 was simply connected to a 50 ohm load but could have been tuned separately.
(86) FIG. 10 shows a graph of S parameters for the MIMO antenna structure of FIGS. 9A and 9B under the circumstances described above. Thus, the return loss for the balanced antenna system 10 was found to be approximately 11.62 dB while the return loss for Port 2 of the chassis antenna 100 was approximately 35.92dB. It is also apparent from FIG. 10 that the S21 return loss (which is a measure of the isolation between the balanced antenna 10 and the chassis antenna 100) is approximately 30.53 dB.
(87) The low isolation between the balanced antenna 10 and the chassis antenna 100 can be explained from the current distribution plots of FIGS. 11A and 11B, which are taken when either the balanced antenna 10 is driven (FIG. 11A) or the chassis antenna 100 is driven (FIG. 11B). Thus, it can be seen that in each case the current is very much concentrated around the antenna being driven, with little or no current flowing in the other antenna.
(88) FIGS. 12A and 12B also corroborate this result through an illustration of the average current distribution when either the balanced antenna 10 is driven or the chassis antenna 100 is driven.
(89) As illustrated in FIGS. 13A and 13B, the directivity of the current distribution for the MIMO antenna system is different when the balanced antenna is driven (FIG. 13A) and when the chassis antenna is driven (FIG. 13B). This therefore explains why the coupling is low between the two antennas.
(90) It is noted that the present simulations are calculated using ideal components. In practice, integrating the matching circuit of FIGS. 5A and 5B with the antenna structure 90 will have an effect on the efficiency and gain of the antenna structure. This effect has been calculated such that, after the antenna structure 90 has been connected to the matching circuit, the balanced antenna has a total efficiency of 1.038 dB and a Realized Gain of 0.7664 dB while the chassis-antenna has a total efficiency of5.06 dB and a Realized Gain of 3.072 dB. These results therefore reflect the addition loss of the matching circuit.
(91) FIGS. 14A and 14B show, respectively, a front view and a rear view of another combined MIMO antenna system 140 comprising the balanced antenna 10 of FIGS. 1A and 1B and the chassis antenna 100 of FIGS. 9A and 9B but, in this case, the chassis antenna 100 is mounted on the side rather than the end of the substrate 12. The chassis antenna 100 is still mounted towards the opposite end of the substrate 12 (and on the opposite side of the substrate) when compared to the balanced antenna 10, so as to provide the largest possible separation between the two antennas in this configuration. As illustrated, the antennas are separated along the length of the substrate 12 by a distance of approximately 60 mm.
(92) As above, Port 1 was connected to the balanced antenna 10 and driven by the circuit of FIGS. 5A and 5B, Port 2 was connected to the large coupling element 112 (driven by a matching circuit of the type described in GB0918477.1) and Port 3 was connected to the small coupling element 114 and was simply tied to a 50 ohm load.
(93) FIG. 15 illustrates that the return loss for the balanced antenna 10, incorporating a matching network of FIGS. 5A and 5B, is about 22.16 dB while the return loss for the chassis antenna 100 is about 46.98 dB. It is also clear from FIG. 15 that the S12 isolation between the balanced antenna 10 and the chassis-antenna 100 is about 22.62 dB.
(94) FIGS. 16A and 16B show, respectively, a front view and a rear view of another combined MIMO antenna system 150 comprising the balanced antenna 10 of FIGS. 1A and 1B plus a second balanced antenna 152 (which is identical to that of FIGS. 1A and 1B) and which is mounted towards the opposite end of the substrate 12 (and on the same side of the substrate 12) when compared to the balanced antenna 10, but is orientated orthogonally to the balanced antenna 10 is located along the top edge as viewed. As illustrated, the antennas are separated along the length of the substrate 12 by a distance of approximately 60 mm.
(95) In this embodiment, Port 1 is connected as described above and Port 2 is connected to the second balanced antenna 152 and to a further optimized circuit similar to that shown in FIGS. 5A and 5B.
(96) FIG. 17 illustrates the S parameters for the MIMO antenna system 150 when each of the antennas are connected to the respective matching networks. The return loss for the balanced antenna 10 is about 11.12 dB while the return loss for the second balanced antenna 152 is about 9.52 dB. It can also be seen that the S12 isolation between the two balanced antennas is about 17.65 dB in this embodiment.
(97) FIGS. 18A and 18B show, respectively, a front view and a rear view of another combined MIMO antenna system 160 comprising two balanced antennas 10 (which are identical to those described in relation to FIGS. 1A and 1B) plus the chassis antenna 100 described in relation to FIGS. 9A and 9B. However, in this particular embodiment the substrate 162 has an area of approximately 9066 mm.sup.2, the chassis antenna 100 is mounted in the middle of the long side of the substrate 162, and the two balanced antennas 10 are mounted towards each of the two corners of the substrate 162 furthest from the chassis antenna 100. The two balanced antennas 10 are also orientated at an inwardly inclined angle of approximately 45 with respect to the length of the substrate 162. The two balanced antennas 10 are each separated from the chassis antenna 100 in a direction along the short dimension of the substrate 162 by about 30.64 mm and are separated from each other along the long dimension of the substrate 162 by about 19.29 mm.
(98) In this embodiment, Port 1 and Port 2 are connected to each of the balanced antennas 10 and their respective matching circuits and Port 3 is connected to the large coupling element 112 of the chassis antenna 100 (and its optimized matching circuit). The small coupling element 114 is left as an open circuit in this embodiment.
(99) FIG. 19 illustrates that the return loss for the first of the balanced antennas 10, incorporating the matching network, is about 28.32 dB while the return loss for the second of the balanced antennas 10, incorporating the matching network, is about 12.14 dB, and the return loss for the large coupling element 112, incorporating the matching network, is about 23.41 dB. Fit is also shown that the isolations between these antennas are about 10.885 dB between the two balanced antennas, 16.88 dB between the chassis antenna and the second balanced antenna and 17.07 dB between the chassis antenna and the first balanced antenna.
(100) FIG. 20 shows a circuit diagram for an alternative balun 170 which may be used in embodiments of the present invention. The balun 170 comprises an LC circuit substantially as described by Yeh, Liu and Chiou in Compact 28-GHz Subharmonically Pumped Resistive Mixer MMIC Using a Lumped-Element High-Pass/Band-Pass Balun, published in IEEE Microwave and Wireless Components Letters, Vol. 15, No. 2, February 2005.
(101) As shown, the balun 170 comprises a high pass filter (first filter) 172 and a band pass filter (second filter) 174. A first (unbalanced) port Z.sub.u is connected to the high pass filter 172 and the band pass filter 174 via a T-junction. The high pass filter 172 comprises a capacitor C and an inductor L, and the output from which constitutes an impedance Z.sub.b1. The band pass filter 174 comprises three inductors and two compactors, and the output from which constitutes an impedance Z.sub.b2. In this embodiment, it should be noted that the inductors L are all identical but, in the band pass filter 174, one of the capacitors (constituting a shunt capacitor labelled 2C) is double the value of other capactors C.
(102) It will be noted that the balun 170 is essentially an out-of-phase power splitter which includes one high pass filter 172 and one band pass filter 174 connected in parallel. Although this balun 170 can provide wide bandwidth operation (and has fewer components than the balun 50 described above, resulting in less loss), in practice, the balun 170 may provide less than 180 degrees of phase difference between the unbalanced outputs Z.sub.b1 and Z.sub.b2. Thus, in embodiments where a 180 degree phase difference is required it may be more convenient to employ a balun of the type shown in FIG. 2 and in embodiments where a 180 degree phase difference is not required it may more convenient to employ a balun of the type shown in FIG. 20.
(103) It has also been found that where the balun 170 is employed in embodiments of the invention, it is possible to obtain the desired tuning range of about 470 to 2200 MHz by employing only one balun 170 configuration and only two configurations for each of the first and second matching circuits. Thus, a simpler circuit can be employed when compared to the embodiment shown in FIGS. 5A and 5B.
(104) A balanced antenna system 200 according to a further embodiment of the present invention is illustrated in FIGS. 21A through 21C. As described above, the balanced antenna system 200 is reconfigurable, as will be described in more detail below, and is designed for use in a portable product such as a mobile phone, laptop or PDA.
(105) The balanced antenna system 200 is provided on a microwave substrate 202 (e.g. a printed circuit board, PCB) having a length L.sub.1 of approximately 110 mm, a width W of approximately 40 mm and a thickness H of approximately 5 mm so that the system can easily be accommodated in a conventional mobile phone.
(106) As best illustrated in FIG. 21C, a first radiating element 204 is provided on a first side 206 of the substrate 202, at a first end portion 208 thereof. The first radiating element 204 is constituted by a substantially L-shaped first strip layer which is located in one half of the substrate 202 towards the first end portion 208 and is orientated such that its open side 212 faces inwardly towards the central region of the first end portion 208. A short feed line 214 is provided at a start of the first radiating element 204 closest to the centre of the substrate 202 and extends along the length of the substrate 202.
(107) A second radiating element 216, which is substantially similar to the first radiating element 204, is also provided on the first side 206 of the substrate 202 and is located in an adjacent half of the first end portion 208 of the substrate 202. The second radiating element 216 is therefore constituted by a substantially L-shaped second strip layer which is also orientated such that its open side 220 faces inwardly towards the central region of the first end portion 208. Thus, the second radiating element 216 is orientated in an opposite direction to the first radiating element 204. A short feed line 222 is again provided at a start of the second radiating element 216 closest to the centre of the substrate 202 and extends along the length of the substrate 202.
(108) A gap 224 is provided between the respective feed lines 214, 222 of the first and second radiating elements 204, 216 and between the respective ends 226 of the first and second strips. Accordingly, the first and second radiating elements 204, 216 form a large dipole antenna 227. In the embodiment shown in FIG. 21C, the first and second radiating elements 204, 166 have a long side l.sub.1 extending approximately 70 mm along the length of the substrate 202 and a short side l.sub.2 extending approximately 19 mm along the width of the substrate 202. The width w of each of the first and second strips is approximately 1 mm and the gap 224 has an extent d of approximately 2 mm. Each of the feed lines 214, 222 has a length l.sub.3 of approximately 10 mm.
(109) As shown in FIG. 21B, a ground plane 228 is provided on a second side 230 of the substrate 202, opposite to the first side 206. The ground plane 228 is substantially rectangular and occupies substantially the whole of the substrate 202 surface from a second end 232 thereof (opposite to the first end portion 208) to a position substantially opposite the free ends of the feed lines 214, 222. The ground plane 228 has a length L.sub.2 of approximately 100 mm and extends over the entire width W of the substrate 202.
(110) The balanced antenna system 200 also includes a balun and two matching circuits which are connected to the first and second radiating elements 204, 216 and which are not shown in FIGS. 21A through 21C for reasons of clarity but which would be provided on the first side 206 of the substrate 202, opposite to the ground plane 228.
(111) FIG. 22 shows a circuit diagram of a suitable balun 240 and matching circuit arrangement 250 for the antenna 200 of FIGS. 21A through 21C. The balun 240 comprises one inductor L5 (of 1 nH) and one capacitor C3 (of 0.1 pF) which are connected in parallel branches from a T-junction 242 which splits an unbalanced signal Z.sub.u. The matching circuit arrangement 250 comprises a first matching circuit 252 connected to the inductor L5 of the balun 240 and which terminates in a balanced signal Z.sub.b1 which in practice is fed into the feed line 214 of the first radiating element 204, and a second matching circuit 254 connected to the capacitor C3 of the balun 240 and which terminates in a balanced signal Z.sub.b2 which in practice is fed into the feed line 222 of the second radiating element 216. The first and second matching circuits 252, 254 each comprise an inductor L3, L4 (each of 3.5 nH), a capacitor C1, C2 (each of 10 pF) and a second inductor L1, L2 (each of 9.4 nH). In each case the capacitor C1, C2 may be replaced by a varactor capable of varying from 10 pF to 0.2 pF.
(112) As shown in FIG. 23, the integration of the first and second matching circuits 252 and 254 and the balun 240 has been optimised to produce a required phase difference of about 180 in order to transfer the balanced feeds Z.sub.b1 and Z.sub.b2 to the unbalanced feed Z.sub.u within a required operating bandwidth.
(113) FIG. 24 shows simulated reflection coefficients against frequency for the large balanced antenna 200 when fed by the circuit of FIG. 22. More specifically, the balanced antenna is configured to provide three resonances simultaneously and the varactors (denoted as C.sub.1 and C.sub.2) in the matching circuit arrangement 250 can be varied from 10 pF to 0.2 pF to move the three resonant frequencies simultaneously to cover the low-band (from 700 MHz to 1010 MHz), the mid-band (from 1620 MHz to 2490 MHz) and the high-band (from 2740 MHz to over 3000 MHz), while maintaining a return loss above 6 dB. Accordingly, the present antenna structure 200 is able to provide high efficiency within the required operating bands even when taking into account the loss of real components. Further optimisation of the antenna structure or the addition of further matching circuits could be used to cover any remaining frequency bands of interest.
(114) FIGS. 25A, 25B and 25C show a reconfigurable balanced dipole antenna 300 according to another embodiment of the present invention, which has a single tunable resonant frequency. The balanced antenna 300 is provided on a microwave substrate 302, Taconic TLY-3-0450-C5, which has a permittivity of 2.33, loss tangent of 0.0009, a thickness of 1.143 mm, a length L.sub.1 of approximately 114 mm and a width W of approximately 40 mm. The antenna 300 comprises two metallic radiating elements 304 mounted at a first end 306 of the substrate 302 and which extend substantially over the width W of the substrate 302, such that they occupy a total area of approximately 40 mm14 mm. The radiating elements 304 have a metal thickness of 0.01778 mm and are mounted at a height H of 5 mm above the substrate 302, for example using spacers (not shown). Thus, the antenna 300 can easily be accommodated in a conventional mobile phone.
(115) As shown in FIG. 25B, a metal ground plane 307 is provided on a rear of the substrate 302. The ground plane 307 occupies an area of 10040 mm.sup.2 and terminates at a position opposite the end of the region where the radiating elements 304 are disposed.
(116) The radiating elements 304 are symmetrically arranged on either side of a central longitudinal axis of the substrate 302 such a gap d of 2 mm is provided between each radiating element 304. Although each radiating element 304 is substantially rectangular, an L-shaped cut-out 308 is provided adjacent the first end 306 such that an inner portion 310 of each rectangle is missing at the first end 306 and a transverse slit 312 is provided a short distance from the first end 306, which extends from the missing inner portion 310 to a position close to but spaced from the edge of the substrate 302. At an outer edge of each radiating element 304, at an end opposite to the first end 306, there is provided a further cut-out in the shape of a small rectangle 314. A feed line 315 is provided adjacent an inner edge of each of the radiating elements 304, at the end opposite to the first end 306, for connecting the radiating elements 304 to a control circuit as will be described below. The dimensions of all of the features of the balanced antenna 300 are given in Table 3 below.
(117) TABLE-US-00003 TABLE 3 Dimensions for the antenna shown in FIGS. 25A to 25C H 5 mm w.sub.4 12 mm W 40 mm w.sub.5 14 mm L.sub.1 114 mm l.sub.1 2 mm L.sub.2 100 mm l.sub.2 10 mm w.sub.1 14 mm l.sub.3 2 mm w.sub.2 5 mm l.sub.4 2 mm w.sub.3 17 mm l.sub.5 12 mm
(118) FIG. 26 shows a circuit diagram comprising a balun 320 and matching circuit 322 for the antenna 300 of FIGS. 25A through 25C. In practice, the balun 320 and matching circuit 320 are provided on the substrate 302, opposite to the ground plane 307 and are connected to the radiating elements 304 via the feed lines 315. In this embodiment, the balun 320 comprises an inductor of 10.4 nH and a capacitor of 1.9 pF connected in parallel. The matching circuit 322 comprises two identical circuits, each of which is connected between a branch of the balun 320 and one of the radiating elements 304, and comprises an inductor of 1.3 nH connected in parallel with a varactor C.sub.1 of up to 10 pF, which in turn is connected in series with an inductor of 41 nH.
(119) As illustrated in FIG. 27, by varying the varactors C.sub.1 of FIG. 26 from 10 pF to 0.1 pF it is possible to tune the frequency of the antenna 300 from approximately 700 MHz up to 2434 MHz, with at least 6 dB return loss. For the DVB-H band or beyond 2500 MHz, further matching circuits could be provided. Thus, although the antenna 300 has high dissipated loss on the lumped elements in the circuit and low efficiency at low frequency such as 700 MHz, its size is suitable for MIMO applications in small terminals, for example, Watch Cell Phones.
(120) In a further embodiment of the present invention, there is provided a reconfigurable balanced antenna of the same structure as illustrated in FIGS. 21A to 21C and having the dimensions detailed in table 4 below.
(121) TABLE-US-00004 TABLE 4 Dimensions for the antenna shown in FIGS. 21A to 21C H 5 mm L.sub.1 110 mm W 40 mm L.sub.2 100 mm l.sub.1 50 mm l.sub.2 19 mm l.sub.3 10 mm d 2 mm w 1 mm
(122) Thus, the antenna comprises L-shaped dipole arms, 50 mm40 mm in size, with a 1 mm track width, a metal thickness of 0.01778 mm and has a total size of 11040 mm.sup.2 and ground plane size of 10040 mm.sup.2. The antenna was constructed on a microwave substrate material, Taconic TLY-3-0450-05, which has a permittivity of 2.33, loss tangent of 0.0009 and a thickness of 1.143 mm.
(123) FIG. 28 shows a circuit diagram for the antenna described above. This circuit comprises a balun 330 and matching circuit 332 which would be provided on the substrate 202, opposite to the ground plane 228 and would be connected to the radiating elements 204, 216 via the feed lines 214, 222. In this embodiment, the balun 330 comprises an inductor of 12 nH and a capacitor of 2 pF connected in parallel. The matching circuit 332 comprises two identical circuits, each of which is connected between a branch of the balun 330 and one of the radiating elements 204, 216, and comprises an inductor of 7.3 nH connected in parallel with a varactor C.sub.1 of up to 10 pF, which in turn is connected in series with an inductor of 26 nH.
(124) As illustrated in FIG. 29, by varying the varactors C.sub.1 of FIG. 28 from 10 pF to 0.1 pF it is possible to tune three separate frequencies to simultaneously cover from 633 MHz to over 3000 MHz with at least 6 dB return loss. More specifically, it is possible to tune the low-band resonance frequency from 648 MHz to 1616 MHz, the mid-band resonance frequency from 1704 MHz to 2560 MHz and the high-band resonance frequency from 2280 MHz to over 3000 MHz while maintaining a return loss above 6 dB. For the DVB-H band, further matching circuits may be employed. Thus, despite some dissipated loss in the lumped elements and substrate, leading to low efficiency at low frequencies, the exceptional tuning range makes this antenna suitable for MIMO applications in small terminals, particularly when constructed from lower loss materials.
(125) FIG. 30 shows a front and rear view of a further reconfigurable balanced antenna 400 according to an embodiment of the present invention. The antenna 400 is similar to that described above and shown in FIGS. 21A through 21C but this time has the dimensions detailed in Table 5 below.
(126) TABLE-US-00005 TABLE 5 Dimensions for the antenna showed in FIG. 30 H 5 mm L.sub.1 110 mm W 40 mm L.sub.2 100 mm l.sub.1 70 mm l.sub.2 19 mm l.sub.3 10 mm d 2 mm w 1 mm
(127) Thus, the antenna 400 comprises L-shaped dipole radiating elements 404, 70 mm40 mm in size, with a 1 mm track width, a metal thickness of 0.01778 mm and has a total size of 11040 mm.sup.2 and ground plane 406 of 10040 mm.sup.2. The antenna was constructed on a microwave substrate material 402, Taconic TLY-3-0450-C5, which has a permittivity of 2.33, loss tangent of 0.0009 and a thickness of 1.143 mm. A port 408 is provided on the ground plane 406 for driving the radiating elements 404 via a suitable circuit.
(128) FIG. 31 shows such a circuit diagram for the antenna 400. This circuit comprises a balun 410 and matching circuit 412 which would be provided on the substrate 402, opposite to the ground plane 406 (but drivable through the port 408) and would be connected to the radiating elements 404 via feed lines. In this embodiment, the balun 410 comprises an inductor of 13.2 nH and a capacitor of 2.3 pF connected in parallel. The matching circuit 412 comprises two identical circuits, each of which is connected between a branch of the balun 410 and one of the radiating elements 404, and comprises an inductor of 1.9 nH connected in parallel with a varactor C.sub.1 of up to 10 pF, which in turn is connected in series with an inductor of 9.4 nH.
(129) As illustrated in FIG. 32. by varying the varactors C.sub.1 of FIG. 31 from 10 pF to 0.28 pF it is possible to tune three separate frequencies to simultaneously cover from 705 MHz to over 3000 MHz with at least 6 dB return loss. More specifically, it is possible to tune the low-band resonance frequency from 705 MHz to 951 MHz, the mid-band resonance frequency from 1692 MHz to 2457 MHz and the high-band resonance frequency from 2826 MHz to over 3000 MHz while maintaining a return loss above 6 dB. The simulation results, using ideal components, therefore show that the antenna 400 has three-band behaviour. The low-band and mid-band can cover most of the existing cellular services, since the low-band can be tuned to cover LTE700, GSM850 and EGSM900 and the mid-band can be tuned to cover PCN, GSM1800, GSM1900, PCS and UMTS. The applicants have also found that incorporating real components provides at least 3.77 dB of total efficiency at 684 MHz. It is therefore believed that the antenna 400 can address the low efficiency problem that may be associated with some of the previous embodiments at low frequencies.
(130) FIGS. 33A and 33B show, respectively, a front and rear view of another balanced antenna 500 according an embodiment of the present invention and which is suitable for Multiple-Input-Multiple-Output (MIMO) applications. The antenna 500 essentially comprises the antenna 400 of FIG. 30 combined with a two-port chassis-antenna 100 as shown in FIGS. 9A and 9B and as described above. Thus, the two-port chassis antenna 100 is of the type described in detail in GB0918477.1 and comprises a pair of non-resonant coupling elements 112, 114, that are capable of simultaneous dual-band operation.
(131) The MIMO antenna 500 has a total size of 11840 mm.sup.2 and a ground plane 502 of 10040 mm.sup.2. The chassis antenna 100 occupies a small volume of 4047 mm.sup.3 and is mounted off a second end of the substrate 402, opposite to the end where the radiating elements 404 are disposed. As shown in FIG. 33A, the radiating elements 404 are provided on an additional U-shaped substrate 504 which is wider and longer than the radiating elements 404 themselves, to provide mechanical support. The balanced antenna element 400 and the coupling elements 112, 114 in the chassis antenna 100 are supported by Rohacell material. A first port, Port 1, is provided through the ground plane 502 for driving the radiating elements 404 via a suitable circuit. Similarly, a second port, Port 2, is provided on an edge of the substrate 402 for driving coupling element 114 and a third port, Port 3, is provided through the second end of the ground plane 502 for driving coupling element 112.
(132) FIG. 34 shows circuit diagrams for the elements provided between each of the ports and their respective radiating and coupling elements 404, 112, 114. Thus, for the balanced antenna 400, Port 1 is connected to a balun 510 comprising an inductor of 13 nH and a capacitor of 3 pF connected in parallel. Each arm of the balun 510 is then connected to one of the radiating elements 404 via an identical matching circuit 512. Each matching circuit 512 comprises an inductor of 1.9 nH connected in parallel with a varactor C.sub.1 of up to 10 pF, which in turn is connected in series with an inductor of 5.6 nH. For the chassis-antenna 100, a first L-network matching circuit 514 is connected between Port 2 and coupling element 114 and a second L-network matching circuit 516 is connected between Port 3 and coupling element 112. The first matching circuit 514 comprises a parallel inductor of 5.1 nH connected to a varactor C.sub.2 of up to 10 pF, which in turn is connected in series to an inductor of 27 nH. The second matching circuit 516 comprises a parallel inductor of 2.4 nH connected to a varactor C.sub.3 of up to 10 pF, which in turn is connected in series to an inductor of 2 nH.
(133) The MIMO antenna 500 was simulated in CST Microwave Studio and the s4p file representing the frequency response of the antenna was then used to determine the optimum component values detailed above for each matching circuit using Microwave Office, from Applied Wave Research.
(134) The antenna 500 was also demonstrated with the four varactor diodes C.sub.1, C.sub.1, C.sub.2, C.sub.3 being of the type MV34003-150A, having a capacitance variable from 0.409 pF to 15.435 pF (which was broader than the range described above) for an applied voltage of 0 V to 15 V. A dc bias line, incorporating a 10 k resistor, was attached to the anode of each varactor to supply positive voltage. The resistor was employed for damping any residual RF signals appearing on the dc line. The negative voltage was supplied from an inner conductor of an SMA connector (i.e. a coaxial RF connector having a 50 Ohm impedance) by using a bias-tee ZX85-122G-S+, from Mini-Circuits.
(135) FIG. 35 shows measured reflection coefficients against frequency for when the varactors in the circuit of FIG. 34 are varied from approximately 15.4 pF to 0.4 pF. Thus, it can be seen that the antenna 500 can be operated in three simultaneous bands. Ports 2 and 3, of the chassis-antenna 100, were open circuit during the measurement. The voltage across the varactors was varied from 0 V to 15 V and the resulting resonant frequencies varied from 646 MHz to 848 MHz for the low band, 1648 MHz to 2074 MHz for the mid-band and 2512 MHz to over 3000 MHz for high band, while maintaining a return loss above 6 dB. Table 6 below shows the measured reflection coefficient for both ports of the chassis-antenna 100 while the varactor voltages are varied from 0 V to 15 V. During measurements on each port, the other two ports are left open circuit. Unlike the balanced antenna 400 each port of the chassis antenna 100 drives a single resonance. The frequencies of each port vary from 597 MHz to 1124 MHz and 1586 MHz to 2332 MHz, respectively. Thus, the antenna 500 provides MIMO operation from 646 MHz to 848 MHz and 1648 MHz to 2074 MHz. It is also likely that the frequency tuning range could be increased, if the capacitance tuning range of the varactors was wider.
(136) The instantaneous bandwidth at various frequencies for ports 2 and 3 are also shown in Table 6. The bandwidth of port 1 for the same frequencies is also shown. The smallest of these thus represent the instantaneous MIMO bandwidth. It can be seen that port 2 gives a considerably narrower bandwidth than port 3. Table 6 shows that the minimum MIMO bandwidth is 14 MHz (at 771 MHz centre frequency) and the maximum MIMO bandwidth is 93 MHz (at 1812 MHz centre frequency).
(137) TABLE-US-00006 TABLE 6 Measured refection coefficients for Ports 2 and 3 (Chassis Antenna) as varactors varied from 0 V to 15 V respectively Bandwidth for Refl. Port 1 at same Voltage Freq. Coeff. Bandwidth@ frequency (V) (MHz) (dB) 6 dB (MHz) (MHz) Port 2 0 597 19.72 39 1 644 21.30 42 17 3.5 771 32.71 34 14 7 921 18.32 44 12 1075 12.03 41 15 1124 10.62 35 Port 3 0 1586 21.60 236 1 1631 25.97 289 73 3.5 1812 15.89 233 93 7 2067 18.41 224 61 12 2300 25.11 204 15 2332 23.91 189
(138) Table 7 below gives the measured S parameters for the MIMO antenna 500. It is clear from these results that isolation is good as S21 is at least 15 dB over all of the bands.
(139) TABLE-US-00007 TABLE 7 Measured S parameters for the MIMO antenna of FIG. 33A-B Frequency Voltage (V) Ports (MHz) S11 (dB) S22 (dB) S21 (dB) 0* & 1.3.sup.+ 1 & 2 646 16.85 32.14 20.15 1.72* & 2.53.sup.+ 1 & 2 710 16.25 27.21 17.17 6.85* & 5.6.sup.+ 1 & 2 850 10.85 36.53 18.25 0* & 1.1.sup.+ 1 & 3 1645 14.62 25.36 26.57 1.9* & 2.1.sup.+ 1 & 3 1710 16.01 18.71 25.40 12* & 6.4.sup.+ 1 & 3 2050 13.09 14.54 15.58 *applied voltage to the varactors of the balanced antenna
(140) FIG. 36 shows a view similar to that of FIG. 33B but wherein the ground plane 502 has been modified to include a U-shaped slot 520 around Port 1. For certain applications it is desirable for the MIMO antenna 500 to have a total efficiency of at least 4.5 dB for the main transmission and reception antenna (e.g. the chassis antenna 100) and 5.5 dB for the second reception antenna (e.g. the balanced antenna 400). However, it was noted that the realized gain of the chassis antenna 100, when integrated with the balanced antenna 400 in accordance with FIGS. 33A and 33B, dropped at least 5 dB at low frequencies. In order to address this problem, the applicants have proposed to isolate the matching circuits of the balanced antenna 400 and the chassis antenna 100 by introducing the slot 520 in the ground plane 502, as shown in FIG. 36.
(141) Table 8 below shows the simulated reflection coefficient, radiation efficiency, total efficiency and realized gain for the MIMO antenna 500 including the slot 520, as shown in FIG. 36. In this embodiment, the realized gain for the chassis-antenna 100 has been improved by 6.28 dB (79.6%), since it is now 7.89 dB at 687 MHz, without affecting other parameters, simply by including the slot 520.
(142) TABLE-US-00008 TABLE 8 Simulated reflection coefficient, radiation efficiency, total efficiency and realized gain for the antenna of FIG. 36 Balanced Antenna Chassis-Antenna Capacitor Simulated Simulated Simulated Simulated Simulated Simulated Simulated Simulated Simulated For Balanced Frequency Refl. Coe. Rad. Effic. Tot. Effic. Rlzd. Gain Refl. Coe. Rad. Effic. Tot. Effic. Rlzd. Gain Isolation Antenna (MHz) (dB) (dB) (dB) (dB) (dB) (dB) (dB) (dB) (dB) 10 pF 687 20.16 3.34 3.38 2.10 12.92 3.36 3.59 1.61 48.92 1722 12.64 1.24 1.70 1.80 10.01 0.45 0.70 4.49 28.99
(143) It is clear from the above that embodiments of the present invention can provide a reconfigurable balanced antenna which can be tuned over a wide range of frequencies (e.g. from 646 MHz to over 3000 MHz) and which can be incorporated along with another antenna into a MIMO antenna structure which has good antenna isolation. The balanced antenna may be able to cover the existing cellular service bands known as DVB-H, GSM710, GSM850, GSM900, GPS1575, GSM1800, PCS1900 and UMTS2100 and is an ideal candidate for MIMO applications, especially in small terminals such mobile devices, laptops and PDAs.
(144) It will be appreciated by persons skilled in the art that various modifications may be made to the above-described embodiments without departing from the scope of the present invention. In particular, features described in relation to one embodiment may be incorporated into other embodiments also.
