Welding current source

Abstract

A welding current source for providing a welding current and a welding voltage at an output in order to carry out an arc welding process includes an input-side rectifier, an inverter, which is operated with a switching frequency, a transformer having a primary winding and at least two secondary windings, at least two rectifiers arranged between the secondary windings and the output, and at least one capacitor and one load resistor at the output. At least one current-limiting reactor is arranged on the second secondary winding and the load resistor for discharging the capacitor, which can be charged by the current-limiting reactor, the current-limiting reactor, and the capacitor are dimensioned in such a way that the maximum value of the no-load voltage at the output is greater than the voltage corresponding to the transmission ratio of the primary winding to the secondary winding of the transformer.

Claims

1. A welding current source for supplying of a welding current and a welding voltage at an output for the performance of an arc welding process, comprising: an input-side rectifier, an inverter operated at a switching frequency, a transformer with a primary winding and at least first and second secondary windings, the second secondary winding having a greater number of turns than the first secondary winding, at least first and second rectifiers, the first rectifier being arranged between the first secondary winding and the output and the second rectifier being arranged between the second secondary winding and the output, at least a first capacitor and one load resistor at the output, at least a current-limiting inductor arranged on the second secondary winding before the second rectifier, and a further current-limiting inductor, wherein the load resistor for discharging the first capacitor that can be charged via the current-limiting inductor, the current-limiting inductor, and the first capacitor are dimensioned such that a maximum value of a no-load voltage at the output is greater than a voltage corresponding to the transmission ratio of the primary winding to the second secondary winding of the transformer, and wherein the second secondary winding of the transformer has a center tapping, and the second secondary winding of the transformer has first and second terminal connections connected, respectively, to the current-limiting inductor and the further current-limiting inductor.

2. The welding current source in accordance with claim 1, wherein the maximum value of the no-load voltage at the output is 5% to 30% higher than the voltage corresponding to the transmission ratio of the primary winding to the second secondary winding of the transformer.

3. The welding current source in accordance with claim 1, wherein the load resistor at the output is dimensioned such that a time constant of an RC-element comprising said load resistor and the first capacitor at the output is between 1 and 20 times a reciprocal of the switching frequency of the inverter.

4. The welding current source in accordance with claim 1, wherein a resonant frequency of a resonant circuit comprising the current-limiting inductor and the first capacitor at the output is between 3 times and 20 times the switching frequency of the inverter.

5. The welding current source in accordance with claim 1, wherein the current-limiting inductor and the further current-limiting inductor form a coupled current-limiting inductor.

6. The welding current source in accordance with claim 1, further comprising a second capacitor at the output, wherein the first capacitor and the second capacitor at the output are connected to ground.

7. The welding current source in accordance with claim 1, wherein the first capacitor at the output is at least 10 nF.

8. The welding current source in accordance with claim 1, wherein the current-limiting inductor has an inductance between one fifth of a reciprocal of the switching frequency (f.sub.S) and five times the reciprocal of the switching frequency (f.sub.S).

9. The welding current source in accordance with claim 1, wherein the number of turns of the second secondary winding is at least twice as large as the number of turns of the first secondary winding of the transformer.

10. The welding current source in accordance with claim 1, wherein the second secondary winding can transmit a power greater than 250 W.

11. The welding current source (1) in accordance with claim 1, wherein the switching frequency is between 20 kHz and 200 kHz.

12. The welding current source in accordance with claim 1, wherein the maximum value of the no-load voltage at the output is between 90V and 113V DC.

Description

(1) The objective invention is explained in more detail below with reference to FIGS. 1 to 7. The figures show advantageous configurations of the invention in an exemplary, schematic and non-restrictive manner. Here:

(2) FIG. 1 shows a block diagram of a primary clocked welding current source;

(3) FIG. 2 shows a diagrammatic circuit of a first embodiment of an inventive welding current source;

(4) FIG. 3 shows the U/I characteristic of an inventive welding current source;

(5) FIG. 4 shows the time courses of current and voltage of an inventive welding current source;

(6) FIG. 5 shows a simplified circuit diagram to explain the no-load voltage profile;

(7) FIG. 6 shows the time courses of current and no-load voltage for the circuit arrangement in FIG. 5; and

(8) FIG. 7 shows a diagrammatic circuit of a further embodiment of an inventive welding current source.

(9) FIG. 1 shows a block diagram of a primary clocked welding current source 1 for supplying of a welding current I and a welding voltage U at an output 2 for the performance of an arc welding process, for example a TIG welding process, a rod electrode welding process, or a MIG/MAG welding process. The input voltage U.sub.AC is rectified to the intermediate circuit voltage U.sub.ZK by means of an input-side rectifier 3. A power factor correction filter (PFC filter), which is not described in any further detail, or also a so-called booster, can also be used for purposes of increasing, smoothing and/or stabilising the intermediate circuit voltage U.sub.ZK. A downstream inverter 4, which is embodied, for example, in the form of a full bridge, generates an alternating voltage U.sub.1 with the switching frequency f.sub.S, which is applied to the primary winding 6 of a transformer 5. This alternating voltage U.sub.1 is approximately a square-wave voltage and is transformed down to the voltage U.sub.2 by the transformer 5 in order to achieve at the same time a high current transformation ratio, and thus a high welding current I. Accordingly, the secondary winding 7 of the transformer 5 has a high current carrying capacity in order to be able to provide welding currents I of the order of several 100 A, for example up to 600 A. The current on the secondary winding 7 of the transformer 5 is rectified via an output-side rectifier 9 and is provided at the output 2. In addition, a capacitor C.sub.B and a load resistor R.sub.B are arranged at the output 2 so as to dampen disturbances.

(10) A second secondary winding 8 of the transformer 5 can be seen in the diagrammatic circuit of a first embodiment of an inventive welding current source 1 as shown in FIG. 2. The voltage U.sub.3 of the second secondary winding 8 is fed via a further output-side rectifier 10 to the output 2 of the welding current source 1. This second secondary winding 8 is embodied with a larger number of turns N.sub.3 than the number of turns N.sub.2 of the first secondary winding 7. This results in a higher voltage U.sub.3 on the second secondary winding 8 than on the first secondary winding 7. In no-load operation, the voltage U.sub.3 on the second secondary winding 8 is therefore dominant at the output 2, and specifies the no-load output voltage U.sub.LL of the welding current source 1.

(11) To save costs, however, the second secondary winding 8 is embodied with less power output than the first secondary winding 7, which can deliver the high welding currents I. For this purpose, in accordance with the invention a current-limiting inductor L.sub.LR is arranged on the second secondary winding 8, which causes a voltage drop at high currents I, and thus limits the current I.sub.L of the second secondary winding 8. The deployment of the L.sub.LR current-limiting inductor means that the high no-load voltage U.sub.LL is provided by the second secondary winding 8, and the very high currents I in the welding operation are provided by the first secondary winding 7. The increased no-load voltage U.sub.LL instigates improved ignition behaviour, but can only be specified within coarse ranges by virtue of the transmission ratios of the transformer 5. For the high current transformation ratio, the first secondary winding 7, which is designed for welding operation at high currents, often has only one winding with a no-load voltage U.sub.2LL of 45 V, for example. If the second secondary winding 8 is embodied with two windings, it has a no-load voltage U.sub.3LL of 90 V. With three windings, however, the second secondary winding 8 would already have a no-load voltage U.sub.3LL of 135V, as a result of which current safety standards, which only allow a maximum value of 113V DC, would not be fulfilled. In order, nevertheless, to fulfil the safety standards, complex and cost-intensive safety measures would be required in the design of the circuit.

(12) The oscillating circuit ensuing from the current-limiting inductor L.sub.LR and capacitor C.sub.B is dimensioned such that during voltage jumps at the transformer 5, which occur with every change in polarity of the square-wave voltage U.sub.1, the capacitor C.sub.B at the output 2 is charged by a polarity reversal process with the oscillating circuit frequency f.sub.01, f.sub.02. After charging the capacitor C.sub.B at the output 2, the said polarity reversal process is interrupted by the output-side rectifier 10. The load resistor R.sub.B for discharging the capacitor C.sub.B is dimensioned such that the maximum value of the no-load voltage U.sub.LL at the output 2 is higher than the voltage U.sub.3 corresponding to the transmission ratio of the primary winding 6 to the second secondary winding 8 of the transformer 5. With the charging and partial discharge of the capacitor C.sub.B utilised in this way, a maximum no-load voltage U.sub.LL at the output 2 is achieved that is greater than the voltage U.sub.3 corresponding to the transmission ratio, but still less than the welding voltage U permitted by safety regulations.

(13) FIG. 3 shows an example of an output characteristic 13 of an inventive welding current source 1 with a maximum pulse width of the inverter 4. The output characteristic 13 can be divided into three sections 14, 15, and 16. Of these, the first section 14 is located in no-load operation, and represents the maximum voltage increase caused by the charging and partial discharge of the capacitor C.sub.B. The energy stored in the capacitor C.sub.B facilitates a simple ignition of the arc. In the second section 15 of the output characteristic 13, the welding voltage U corresponds to the rectified voltage U.sub.3 of the second secondary winding 8. With increasing welding current I, the voltage drop at the current-limiting inductor L.sub.LR increases and the welding voltage U decreases accordingly. In this current range, the welding current I is transmitted from the second secondary winding 8. In the third section 16 of the output characteristic 13, the welding voltage U is the rectified voltage U.sub.2 of the first secondary winding 7. Here, the current-limiting inductor L.sub.LR causes such a high voltage drop that further increases in current can only be provided via the first secondary winding 7. Accordingly, high welding currents I are primarily transmitted by the first secondary winding 7.

(14) In the output characteristic 13 as illustrated, the maximum value of the no-load voltage U.sub.LL at the output 2 is about 15% higher than the voltage corresponding to the transmission ratio of the primary winding 6 to the second secondary winding 8 of the transformer 5. In general, a range between 5% and 30% is recommended for the increase in the no-load voltage U.sub.LL, since here a sufficient voltage increase can be achieved for improved ignition behaviour, as can the necessary energy storage in the capacitor C.sub.B at the output 2.

(15) FIG. 4 shows idealised profiles for the voltage U.sub.1 on the primary side 6 of the transformer 5, the current I.sub.L through the current-limiting inductor L.sub.LR, and the no-load voltage U.sub.LL at the output 2 of welding current source 1. These are idealised in that circuit elements that are not relevant to the invention, such as circuits for damping that occur in many real circuits (so-called snubber circuits), are not taken into consideration. Similarly, any oscillations caused, for example, by parasitic capacitances of the rectifiers 10, are also neglected. As can be seen from the time profile of the current I.sub.L through the current-limiting inductor L.sub.LR, after a voltage change in U.sub.1, a polarity reversal process is initiated, which instigates the charging of the capacitor C.sub.B at the output 2. With the zero crossing of I.sub.L the output-side rectifier 10 enters a blocking mode, as a result of which the resonant oscillation with the resonant frequency f.sub.02 is interrupted at the end of the charging time t.sub.LAD. The capacitor C.sub.B is then discharged via the load resistor R.sub.B during the discharge time t.sub.ENT, as can be seen in the time profile of the no-load voltage U.sub.LL at the output 2. This sequence is repeated once more within the period T.sub.S of the voltage U.sub.1 on the primary side 6 of the transformer 5. The period T.sub.S corresponds to the reciprocal of the switching frequency f.sub.S of the inverter 4.

(16) From FIG. 4 it can also be discerned that the time constant T.sub.RC of the RC-element, consisting of the said load resistor R.sub.B and the capacitor C.sub.B at the output 2, is approximately 2 times the reciprocal of the switching frequency f.sub.S of the inverter 4. Depending on the size of the current-limiting inductor L.sub.LR and capacitor C.sub.B, time constants T.sub.RC between 1 and 20 times the reciprocal of the switching frequency f.sub.S of inverter 4 can be advantageous. The said time constant T.sub.RC is used to configure the discharge of the capacitor C.sub.B, and thus the maximum value of the no-load voltage U.sub.LL of the welding current source 1. Safety regulations regarding the level of the welding voltage U can thus be fulfilled.

(17) FIG. 5 shows a simplified circuit diagram to explain the profile of the no-load voltage U.sub.LL. The input voltage U.sub.3, ideally assumed to be of square-wave form, is present in a circuit arrangement consisting of the current-limiting inductor L.sub.LR, a diode D with a parasitic capacitance C.sub.D as well as the capacitor C.sub.B and load resistor R.sub.B. The no-load output voltage U.sub.LL is formed at the capacitor C.sub.B and the load resistor R.sub.B.

(18) FIG. 6 shows the time courses of the input voltage U.sub.3, the current I.sub.L, and the no-load voltage U.sub.LL of the circuit as shown in FIG. 5. Strictly speaking, FIG. 6 shows a positive alteration of the input voltage U.sub.3 in the steady state, and its effect on the current I.sub.L and the no-load voltage U.sub.LL. Immediately after the positive alteration of the input voltage U.sub.3, the diode D is non-conducting. Thus an oscillating circuit is excited by the alteration in voltage, by way of the current-limiting inductor L.sub.LR and the parasitic capacitance C.sub.D of the diode D, as well as the capacitor C.sub.B at the output. From the profile of the current I.sub.L it can be seen that the frequency of the free oscillation is approx.

(19) $f_{01}=\frac{1}{2\pi \sqrt{L_{\mathrm{LR}}\frac{C_D.Math.C_B}{C_D+C_B}}}$

(20) and in the example of embodiment illustrated lasts for about a ¼-period of the free oscillation. This corresponds to the time period t.sub.L1 in FIG. 6. This discharges C.sub.D to the extent such that a voltage is applied to the diode D in the forward direction. The diode D becomes conducting, and the oscillating circuit is now formed by the current-limiting inductor L.sub.LR and the capacitor C.sub.B at the output 2. The frequency of the free oscillation is now approx.

(21) $f_{02}=\frac{1}{2\pi \sqrt{L_{LR}.Math.C_B}}$

(22) and continues in the example of embodiment for about a ¼-period, the time period t.sub.L2 of the free oscillation now present. The current I.sub.L then crosses zero, causing the diode D to switch back into the non-conducting state. Over the time period t.sub.LAD=t.sub.L1+t.sub.L2 the capacitor C.sub.B at the output 2 is charged and at the same time the no-load voltage U.sub.LL increases. Subsequently, with the diode D in blocking mode, an oscillating circuit is again formed with the parasitic capacitance C.sub.D of the diode D and the oscillation frequency f.sub.01. Here, however, just a decaying oscillation of the energy takes place between the current-limiting inductor L.sub.LR and the parasitic capacitance C.sub.D of the diode D together with the capacitor C.sub.B at the output 2. This polarity reversal process, however, does not contribute significantly to either the charging or the discharging of the capacitor C.sub.B at the output 2. Rather, a discharge of the capacitor C.sub.B takes place in the period t.sub.ENT via the load resistor R. This is also evident from the falling no-load voltage U.sub.LL in the period t.sub.ENT.

(23) Furthermore, the resonant frequency f.sub.02 of the resonant circuit consisting of the current-limiting inductor L.sub.LR and capacitor C.sub.B at the output 2 is approx. 5 times the switching frequency f.sub.S of the inverter 4. The polarity reversal process and thus the charging of the capacitor C.sub.B are thus executed sufficiently quickly. In the current and voltage profiles shown in FIG. 4, the recharging is completed after approx. 1/10 of the period T.sub.S of the transformer alternating voltage U.sub.1. In general, a resonant frequency f.sub.02 of the resonant circuit in terms of the current-limiting inductor L.sub.LR and the capacitor C.sub.B, of between 3 and 20 times the switching frequency f.sub.S of the inverter 4 can be recommended here. Here the polarity reversal process is sufficiently fast, but is nevertheless within a frequency range that can easily be controlled by the circuitry.

(24) FIG. 7 shows a diagrammatic circuit of a further embodiment of an inventive welding current source 1. Snubber circuits that may be necessary to suppress high-frequency signals, smoothing inductors at the output, together with other circuit components that are not relevant to the invention, are not shown in this diagrammatic circuit. The secondary windings 7, 8 of the transformer 5 of the embodiment shown in FIG. 7 are embodied with centre tappings 11, 12. A full-wave rectification can be implemented by means of only two diodes D.sub.11, D.sub.12, or D.sub.21, D.sub.22, as a result of which the extra components and associated costs for a bridge rectification are saved. For purposes of cost reduction it can be advantageous to use identical diodes for the diodes D.sub.11, D.sub.12, D.sub.21, D.sub.22.

(25) Both terminal connections of the second secondary winding 8 of the transformer 5 are connected to a coupled current-limiting inductor L′.sub.LR. By virtue of the arrangement of the coupled current-limiting inductor L′.sub.LR on a common magnetic core, any asymmetries of the electrical properties of the individual inductor windings are compensated for, and saturation of the transformer 5 is prevented. A more complex form of production and/or selection and the associated additional costs are prevented.

(26) In the example of embodiment shown in FIG. 7, the capacitance at the output 2 is formed by two capacitors C.sub.B1, C.sub.B2 connected to ground. These capacitors C.sub.B1, C.sub.B2 not only have the function of increasing the no-load voltage U.sub.LL in accordance with the invention, but also the function of EMC suppression capacitors, resulting in cost savings. The capacitors C.sub.B1, C.sub.B2 also close the current path for the signals of a high-frequency ignition system 17, shown schematically.

(27) The improvement of the ignition characteristics of the welding current source 1 not only depends on the level of the no-load voltage U.sub.LL, but also, in the present case, on the amount of energy stored in the capacitor C.sub.B, that is to say, C.sub.B1, C.sub.B2. Therefore, the capacitor C.sub.B should comprise at least 10 nF so as to ensure sufficient ionisation of the gas in the ignition path. In the example of embodiment shown in FIG. 7, this would mean that each of the capacitors C.sub.B1, C.sub.B2 has a capacitance of at least 20 nF.

(28) The magnitude of the inductance of the current-limiting inductor L.sub.LR can be estimated approximately from the formula

(29) $L_{LR}=\frac{{\left(U_3-U_2\right)}^2}{4.Math.P_{3\max }.Math.\mathrm{fs}}$

(30) If a range from 40V to 70V is assumed for U.sub.3−U.sub.2, and a range between 250 W and 2000 W is assumed for the power P.sub.3max, there ensues for the inductance of the current-limiting inductor L.sub.LR a range between one fifth of the figure of the reciprocal of the switching frequency f.sub.S and five times the figure of the reciprocal of the switching frequency f.sub.3. For welding current sources 1 with a maximum welding current of less than 600 A, a range from half to two-and-a-half times the figure of the reciprocal of the switching frequency f.sub.S has proved to be particularly advantageous.

(31) Once the capacitor C.sub.B and the current-limiting inductor L.sub.LR have been determined, the dimensioning of the load resistor R can beneficially be executed by way of a cautious approach using at least two values of R. The circuit arrangement is preferably put into operation with a load resistor R at the lower limit of the range. A first value for R thus ensues as

(32) $R=\frac{1}{f_S.Math.C_B}.$

(33) Depending on the resulting first no-load voltage U.sub.LL, a second measurement of the no-load voltage U.sub.LL can be executed, for example, at 5 times the value of the load resistor R. By linear interpolation and, if necessary, a further iteration, the required magnitude of the load resistor R for the desired no-load voltage U.sub.LL can be determined.

(34) In the example shown in FIG. 7, the primary winding 6 has a number of turns N.sub.1=9, the first secondary winding 7 has a number of turns N.sub.2=1+1, and the second secondary winding 8 has a number of turns N.sub.3=2+2. Accordingly, the number of turns N.sub.3 of the second secondary winding 8 is twice as large as the number of turns N.sub.2 of the first secondary winding 7 of the transformer 5.

(35) The power P.sub.3max that can be transmitted via the second secondary winding 8 is preferably more than 250 W, in order not only to improve the ignition behaviour of the welding current source 1, but also the welding characteristics at low currents I. With a transmittable power P.sub.3max of 1000 W, for example, voltages U greater than 40V contribute to the stabilisation of the arc at currents I of less than 25 A. On the other hand, at a transferable power P.sub.3max of 2000 W, for example, the second secondary winding 8 contributes to the stabilisation of the arc, even at currents I of less than 50 A. This corresponds approximately to the values of the output characteristic shown in FIG. 3. This is particularly advantageous for special rod electrodes, such as cellulose electrodes.

(36) The switching frequency f.sub.S of the primary-side inverter 4 is preferably between 20 kHz and 200 kHz, which allows the use of cheaper and lighter transformers 5.

(37) A particularly beneficial ignition behaviour in compliance with currently applicable safety regulations ensues, if the maximum value of the no-load voltage U.sub.LL at the output is between 90V and 113V DC.