Abstract
An ADC (50) is disclosed. It has two VTCs (110a, 110b) for converting a first and a second input voltage, respectively, to pulses with delays corresponding to the magnitudes of these voltages. It further has a pulse-detector circuit (120) coupled to outputs (114a, 114b) of the two VTCs (110a, 110b). The pulse-detector circuit (120) is configured to make a transition from a first logic state (0) to a second logic state (F) at a first output (124a) of the pulse-detector circuit (120) in response to the start of the pulse from one of the VTCs (110a) and to make a transition from the first logic state (0) to the second logic state (F) at a second output (124b) of the pulse-detector circuit (120) in response to the start of the pulse from the other VTC. Furthermore, the pulse-detector circuit (120) is configured to reset both the first and the second output (124a, 124b) to the first logic state (0) in response to both the first output (124a) and the second output (124b) of the pulse-detector circuit (120) being set in the second logic state (1). The ADC (50) further has a first TDC (130a) coupled to the first output (124a) of the pulse-detector circuit (120) and a second TDC (130b) coupled to the second output (124b) of the pulse-detector circuit (120).
Claims
1.-17. (canceled)
18. An analog-to-digital converter (ADC) for converting samples of a differential input voltage, formed by pairs of concurrent samples of a first input voltage (Va[n]) and a second input voltage (Vb[n]), the ADC comprising: a differential input port comprising a first input terminal configured to receive the first input voltage (Va[n]) and a second input terminal configured to receive the second input voltage (Vb[n]); a first voltage-to-time converter (VTC) configured to receive the first input voltage (Va[n]); a second VTC configured to receive the second input voltage (Vb[n]); a pulse-detector circuit coupled to outputs of the first VTC and the second VTC and having a first output and a second output; a first time-to-digital converter (TDC) coupled to the first output of the pulse-detector circuit; and a second TDC coupled to the second output of the pulse-detector circuit; wherein for each pair of concurrent samples (Va[n], Vb[n]) of the first input voltage (Va[n]) and the second input voltage (Vb[n]): the first VTC is configured to generate a first pulse (P1) delayed an amount of time (td1) corresponding to the magnitude of the sample of the first input voltage (Va[n]); the second VTC is configured to generate a second pulse (P2) delayed an amount of time (td2) corresponding to the magnitude of the sample of the second input voltage (Vb[n]); the pulse-detector circuit is configured to: make a transition from a first logic state (0) to a second logic state (1) at the first output of the pulse-detector circuit in response to the start of the first pulse (P1); make a transition from the first logic state (0) to the second logic state (1) at the second output of the pulse-detector circuit in response to the start of the second pulse (P2); and in response to both the first output and the second output of the pulse-detector circuit being set in the second logic state (1), reset both the first and the second output to the first logic state (0); whereby a third pulse (P3) is generated at the first output of the pulse-detector circuit and a fourth pulse (P4) is generated at the second output of the pulse-detector circuit; the first TDC is configured to receive the third pulse (P3) and generate a first digital value (xa[n]) corresponding to the duration of the third pulse (P3); and the second TDC is configured to receive the fourth pulse (P4) and generate a second digital value (xb[n]) corresponding to the duration of the fourth pulse (P4).
19. The ADC of claim 18 wherein the first digital value (xa[n]) and the second digital value (xb[n]) form an output sample of the ADC.
20. The ADC of claim 18 wherein the ADC comprises a circuit configured to generate an output sample (x[n]) of the ADC in response to the first digital value (xa[n]) and the second digital value (xb[n]).
21. The ADC of claim 20, wherein the output sample (x[n]) is the difference between the first digital value (xa[n]) and the second digital value (xb[n]).
22. The ADC of claim 18, wherein each of the first TDC and the second TDC is a pulse-shrinking TDC.
23. The ADC of claim 18, wherein the pulse-detector circuit comprises a first flip-flop and a second flip-flop, each having a data input (D), a clock input (clk), a reset input (reset), and an output (Q), wherein the data input (D) of each of the first flip-flop and the second flip-flop is configured to receive a constant signal corresponding to the second logic state (1); the clock input (clk) of the first flip-flop is configured to receive the first pulse (P1); the clock input (clk) of the second flip-flop is configured to receive the second pulse (P2); the output (Q) of the first flip-flop is connected to the first output of the pulse-detector circuit; the output (Q) of the second flip-flop is connected to the second output of the pulse-detector circuit; and the pulse-detector circuit comprises a logic circuit having a first input connected to the output (Q) of the first flip-flop, a second input connected to the output (Q) of the second flip-flop, and an output connected to the reset inputs (reset) of the first flip-flop and the second flip-flop for resetting the first flip-flop and the second flip-flop in response to the outputs (Q) of both the first flip-flop and the second flip-flop being set in the second logic state (1).
24. The ADC of claim 23, wherein the logic circuit is further configured to reset the first flip-flop and the second flip-flop in response to a reset pulse.
25. The ADC of claim 24, comprising circuitry configured to provide said reset pulse in preparation of converting each sample.
26. The ADC of claim 18, wherein the first VTC comprises: a first capacitor configured to, for each sample of the first input voltage (Va[n]), be charged to the voltage value of that sample of the first input voltage (Va[n]) during a first phase of operation; a first current source configured to discharge or charge the first capacitor during a subsequent second phase of operation; a first comparator circuit configured to compare the voltage (Vca) across the first capacitor with a first reference voltage (Vrefa) and to generate the first pulse (P1) at an output of the first comparator circuit.
27. The ADC of claim 26, wherein the second VTC comprises: a second capacitor configured to, for each sample of the second input voltage (Vb[n]), be charged to the voltage value of that sample of the second input voltage (Vb[n]) during the first phase of operation; a second current source configured to discharge or charge the second capacitor during the second phase of operation; a second comparator circuit configured to compare the voltage (Vcb) across the second capacitor with the first or a second reference voltage (Vrefa, Vrefb) and to generate the second pulse (P2) at an output of the second comparator circuit.
28. A receiver circuit comprising the ADC of claim 18.
29. An electronic apparatus comprising the ADC of claim 18.
30. The electronic apparatus of claim 29, wherein the electronic apparatus is a communication apparatus.
31. The electronic apparatus of claim 30, wherein the communication apparatus is a wireless communication device for a cellular communications system.
32. The electronic apparatus of claim 30, wherein the communication apparatus is a base station for a cellular communications system.
33. An analog-to-digital converter (ADC) comprising: a first and a second voltage-to-time converter (VTC), each having an input terminal and an output terminal, and each comprising: a sampling capacitor having a first terminal and a second terminal, wherein the second terminal is connected to a signal ground node; a sampling switch connected between the input terminal of the VTC and the first terminal of the sampling capacitor; a charge-transfer circuit comprising a series connection of a current source and a switch connected to the first terminal of the capacitor; and a comparator circuit having a first input terminal connected to the first terminal of the capacitor, a second input terminal configured to receive a reference voltage (Vrefa, Vrefb), and an output terminal connected to the output terminal of the VTC; a pulse-detector circuit comprising: a first flip-flop and a second flip-flop, each having a data input (D), a clock input (clk), a reset input (reset), and an output (Q), wherein the output (Q) of each of the first and second flip-flop can be in a first logic state (0) or a second logic state (1) and each of the first and the second flip-flop is configured to reset its output (Q) to the first logic state (0) in response to a reset signal at its reset input (reset); and a logic circuit having a first input, a second input, and an output; wherein the data input (D) of each of the first flip-flop and the second flip-flop is configured to receive a constant signal corresponding to the second logic state (1); wherein the clock input (clk) of the first flip-flop is connected to the output terminal of the first VTC; wherein the clock input (clk) of the second flip-flop is connected to the output terminal of the second VTC; wherein the output (Q) of the first flip-flop is connected to a first output of the pulse-detector circuit; wherein the output (Q) of the second flip-flop is connected to a second output of the pulse-detector circuit; wherein the first input of the logic circuit is connected to the output (Q) of the first flip-flop, the second input of the logic circuit is connected to the output (Q) of the second flip-flop, and the output of the logic circuit is connected to the reset inputs (reset) of the first and the second flip-flop; wherein the logic circuit is configured to generate the reset signal in response to both of its first input and its second input being set in the second logic state (1); a first time-to-digital converter, TDC, connected to the output of the first flip-flop of the pulse-detector circuit; and a second TDC connected to the output of the second flip-flop of the pulse-detector circuit.
34. An integrated circuit comprising the ADC of claim 33.
35. An electronic apparatus comprising the ADC of claim 33.
36. The electronic apparatus of claim 35, wherein the electronic apparatus is a communication apparatus.
37. The electronic apparatus of claim 36, wherein the communication apparatus is a wireless communication device for a cellular communications system or a base station for a cellular communications system.
Description
BRIEF DESCRIPTION OF THE DRAWINGS
[0018] FIG. 1 illustrates a communication environment;
[0019] FIG. 2 illustrates a transceiver circuit;
[0020] FIG. 3 is a block diagram of an ADC;
[0021] FIG. 4 shows pulse waveforms;
[0022] FIG. 5 is a block diagram of an ADC;
[0023] FIGS. 6 and 7 are a circuit diagrams of pulse-detector circuits;
[0024] FIG. 8 is a circuit diagram of a VTC;
[0025] FIG. 9 shows waveforms;
[0026] FIG. 10 is a circuit diagram of a VTC; and
[0027] FIG. 11 is a circuit diagram of a pulse-shrinking TDC.
[0028] FIG. 12 illustrates an integrated circuit comprising an ADC.
DETAILED DESCRIPTION
[0029] FIG. 1 illustrates a communication environment wherein embodiments of the present invention may be employed. A wireless communication device 1, or wireless device 1 for short, of a cellular communications system is in wireless communication with a radio base station 2 of the cellular communications system. The wireless device 1 may be what is generally referred to as a user equipment (UE). The wireless device 1 is depicted in FIG. 1 as a mobile phone, but may be any kind of device with cellular communication capabilities, such as a tablet or laptop computer, machine-type communication (MTC) device, or similar. Furthermore, a cellular communications system is used as an example throughout this disclosure. However, embodiments of the present invention may be applicable in other types of systems as well, such as but not limited to WiFi systems.
[0030] The radio base station 2 and wireless device 1 are examples of what in this disclosure is generically referred to as communication apparatuses. Embodiments are described below in the context of a communication apparatus in the form of the radio base station 2 or wireless device 1. However, other types of communication apparatuses can be considered as well, such as a WiFi access point or WiFi enabled device.
[0031] FIG. 2 is a block diagram of an embodiment of a transceiver circuit 10, which can be comprised in a communication apparatus, such as the radio base station 2 or the wireless device 1. In the embodiment illustrated in FIG. 2, the transceiver circuit 10 comprises a digital signal processing (DSP) circuit 15. The DSP circuit 15 may e.g. be what is commonly referred to as a baseband processor. The DSP circuit 15 may e.g. be configured to perform various digital signal processing tasks, such as one or more of coding, decoding, modulation, demodulation, fast Fourier transform (FFT), inverse FFT (IFFT), mapping, demapping, etc.
[0032] Furthermore, in the embodiment illustrated in FIG. 2, the transceiver circuit 10 comprises a transmitter circuit 20. The transmitter circuit 20 comprises a digital-to-analog converter (DAC) 25. The DAC 25 is connected to the DSP circuit 15 and configured to receive, as an input signal of the DAC 25, a digital representation of a signal to be transmitted from the DSP circuit 15. The DAC 25 is further configured to convert the signal to be transmitted to an analog representation, which is an output signal of the DAC 25. The transmitter circuit 20 also comprises a transmitter (Tx) frontend (FE) circuit 30 connected between the DAC 25 and an antenna 35. The Tx FE circuit 30 is configured to transform the output signal from the DAC to a format suitable for transmission via the antenna 35. This may include operations such as frequency upconversion, filtering, and/or amplification. The Tx FE circuit 30 may comprise one or more mixers, filters, and/or amplifiers, such as power amplifiers (PAs), to perform such operations. The design of such Tx FE circuits is, per se, well known to a person skilled in the field of radio transceiver design, and is not discussed herein in any further detail.
[0033] Moreover, in the embodiment illustrated in FIG. 2, the transceiver circuit 10 comprises a receiver circuit 40. The receiver circuit 40 comprises a receiver (Rx) FE circuit 45 connected to the antenna 35. Furthermore, the receiver circuit 40 comprises an ADC 50. The ADC 50 is connected between the Rx FE circuit 45 and the DSP circuit 15. The Rx FE circuit 45 is configured to transform a signal received via the antenna 35 to a format suitable to be input to the ADC 50. This may include operations such as frequency downconversion, filtering, and/or amplification. The Rx FE circuit 45 may comprise one or more mixers, filters, and/or amplifiers, such as low-noise amplifiers (LNAs), to perform such operations. The design of such Rx FE circuits is, per se, well known to a person skilled in the field of radio transceiver design, and is not discussed herein in any further detail. The ADC 50 is configured to receive its (analog) input signal from the Rx FE circuit, and convert it to a digital representation to generate the digital output signal of the ADC 50. This digital output signal of the ADC 50 is input to the DSP circuit 15 for further digital signal processing. In some embodiments, the ADC 50 may be configured to receive a radio-frequency (RF) input signal, without any preceding frequency downconversion.
[0034] In embodiments of this disclosure, the ADC 50 is implemented as a time-domain ADC further described below. Furthermore, embodiments of the ADC 50 disclosed herein are configured to convert samples of a differential input voltage. Said samples of the differential input voltage are formed by pairs Va[n], Vb[n] of concurrent samples Va[n] and Vb[n] of a first input voltage Va and a second input voltage Vb. A difference between the first input voltage Va and the second input voltage Vb forms the value of differential input voltage of the ADC 50. For instance, Va may be a positive component and Vb may be a negative component, respectively, of the differential input voltage, or vice versa. Samples Va[n] and Vb[n] having the same value of the sequence index n are sampled at the same sampling time and referred to herein as concurrent samples. The sampling time for sequence index is below referred to as nT, where T is the sampling period of the ADC 50. However, this is a mere example used for simplicity. For instance, nonuniform sampling may be used, in which case the ADC has no fixed sampling period.
[0035] FIG. 3 illustrates an embodiment of the ADC 50. It has a differential input port 102. The differential input port 102 comprises a first input terminal 102a configured to receive a series of samples of a first input voltage Va[n], where n is a sequence index. The differential input port 102 further comprises a second input terminal 102b configured to receive a series of samples of a second input voltage Vb[n].
[0036] The ADC 50 comprises a first voltage-to-time converter (VTC) 110a configured to receive the first input voltage Va[n]. In FIG. 3, the first VTC 110a has an input terminal 112a connected to the first input terminal 102a of the ADC 50 for this purpose. The first VTC 110a further has an output 114a. Furthermore, the ADC 50 comprises a second VTC 110b configured to receive the second input voltage Vb[n]. In FIG. 3, the second VTC 110b has an input terminal 112b connected to the second input terminal 102b of the ADC 50 for this purpose. The second VTC 110b further has an output 114b. The functionality of the VTCs 110a and 110b is further described below.
[0037] The ADC 50 comprises a pulse-detector circuit 120 coupled to the outputs 114a and 114b of the first VTC 110a and the second VTC 110b. In FIG. 3, the pulse-detector circuit 120 has a first input 122a connected to the output 114a of the first VTC 110a and a second input 122b connected to the output 114b of the second VTC 110b for this purpose. Furthermore, the pulse-detector circuit 120 has a first output 124a and a second output 124b. The functionality of the pulse-detector circuit 120 is further described below.
[0038] Furthermore, the ADC 50 comprises a first time-to-digital converter (TDC) 130a coupled to the first output 124a of the pulse-detector circuit 120. In FIG. 3, the first TDC 130a has an input 132a connected to the first output 124a of the pulse-detector circuit 120 for this purpose. Moreover, the first TDC 130a has an output 134a.
[0039] The ADC 50 further comprises a second TDC 130b coupled to the second output 124b of the pulse-detector circuit 120. In FIG. 3, the second TDC 130b has an input 132b connected to the second output 124b of the pulse-detector circuit 120 for this purpose. Furthermore, the second TDC 130b has an output 134b.
[0040] According to some embodiments, the VTCs 110a and 110b, the pulse-detector circuit 120, and the TDCs 130a and 130b are configured to operate as follows. The operation is described for a pair of concurrent samples Va[n], Vb[n] (i.e. for a given value of n) of the first input voltage Va and the second input voltage Vb. The operation is the same for each such pair of concurrent samples Va[n], Vb[n] (i.e. for each value of n). The operation is illustrated with pulse waveforms in FIG. 4. Reference is made below to a first logic state and a second logic state. In the examples presented, the first logic state is considered to be 0, generally represented with a low voltage level (such as 0V), and the second logic state is considered to be 1, generally represented with a high voltage level (i.e. a voltage level higher than the low voltage level, e.g. equal to a supply voltage of the circuitry). However, in some embodiments, the first logic state and the second logic state could equally well be 1 and 0, respectively, with suitable modifications of the circuitry as would be readily understood by a skilled person. Furthermore, in FIG. 4, the pulses start with a rising edge and end with a falling edge. However, in some embodiments, pulses that are inverted with respect to those shown in FIG. 4 may be used, i.e. starting with a falling edge and ending with a rising edge.
[0041] In some embodiments, the first VTC 110a is configured to perform the sampling of the first input voltage Va. That is, in these embodiments, the voltage Va at the input 112a of the first VTC 110a is a time-continuous voltage. In other embodiments, the sampling may be performed by a sampling circuit (not shown in the drawings) preceding the first VTC 110a. That is, in these embodiments, the voltage Va at the input 112a of the first VTC 110a is a time discrete (e.g. piecewise constant) voltage.
[0042] Similarly, in some embodiments, the second VTC 110b is configured to perform the sampling of the second input voltage Vb. That is, in these embodiments, the voltage Vb at the input 112b of the second VTC 110b is a time-continuous voltage. In other embodiments, the sampling may be performed by a sampling circuit (not shown in the drawings) preceding the second VTC 110b. That is, in these embodiments, the voltage Vb at the input 112b of the second VTC 110b is a time discrete (e.g. piecewise constant) voltage.
[0043] The first VTC 110a is configured to generate a first pulse P1 (see FIG. 4) delayed an amount of time td1 that corresponds to the magnitude of the sample of the first input voltage Va[n]. Ideally, the delay td1=kVa[n]+m, where k and m are constants. However, such an exact ideal relation is not possible in practice due to manufacturing tolerances, noise, etc. In FIG. 4, the starting time for the pulse P1 is labeled t1.
[0044] Similarly, the second VTC 110b is configured to generate a second pulse P2 (see FIG. 4) delayed an amount of time td2 corresponding to the magnitude of the sample of the second input voltage Vb[n]. Again, ideally, the delay td2=kVb[n]+m with the same k and m as for td1. Again, such an exact ideal relation is not possible in practice due to manufacturing tolerances, noise, etc. In FIG. 4, the starting time for the pulse P2 is labeled t2. Furthermore, in FIG. 4, t2 occurs after t1. Of course, which of t1 and t2 is first depends on the values of the input voltage Va[n] and Vb[n]. As illustrated in FIG. 4, the first and second pulses P1 and P2 are ended before the next sample time instant (n+1)T. In FIG. 4, this happens at a time instant labeled t6.
[0045] In FIG. 4, the delays td1 and td2 are measured from the sampling time instant nT. However, since differential signals are considered, the signal is represented by the difference between td1 and td2, so any time instant other than nT could be used as reference as well.
[0046] In a conventional differential time-domain ADC as outlined in the background section, a TDC would use one of P1 and P2, whichever comes first, as a start signal, and the other one of P1 and P2 as a stop signal, and thereby measure the absolute difference between td1 and td2 (and use some additional logic circuitry that keeps track of the sign of the difference). If this difference is small, many known TDC implementations have difficulties to accurately measure the difference, resulting in an overall poor linearity of such a conventional differential time-domain ADC. According to embodiments of the ADC 50 disclosed herein, this problem is alleviated by means of the pulse-detector circuit 120, which guarantees a certain (nonzero) minimum duration of time for the TDCs 130a and 130b to measure, as further described below.
[0047] The pulse-detector circuit 120 is configured to make a transition from the first logic state (0) to the second logic state (1) at the first output 124a of the pulse-detector circuit 120 in response to the start of the first pulse P1, as illustrated in FIG. 4. In FIG. 4, this happens at a time instant labeled t3. Furthermore, pulse-detector circuit 120 is configured to make a transition from the first logic state (0) to the second logic state (1) at the second output 124b of the pulse-detector circuit 120 in response to the start of the second pulse P2. In FIG. 4, this happens at a time instant labeled t4. Moreover, the pulse-detector circuit is configured to reset both its first and its second output 124a, 124b to the first logic state (0) in response to both the first output 124a and the second output 124b of the pulse-detector circuit 120 being set in the second logic state (1). In FIG. 4, this happens at a time instant labeled t5.
[0048] Thereby, as illustrated in FIG. 4, a third pulse P3 is generated at the first output 124a of the pulse-detector circuit and a fourth pulse P4 is generated at the second output 124b of the pulse-detector circuit 120. The difference between the durations of the third pulse P3 and the fourth pulse P4 is a time-domain representation of the differential input voltage Va[n]Vb[n]. Ideally, said difference is proportional to Va[n]Vb[n].
[0049] The first TDC 130a is configured to receive the third pulse P3 and generate a first digital value xa[n] corresponding to the duration of the third pulse P3. The second TDC 130b configured to receive the fourth pulse P4 and generate a second digital value xb[n] corresponding to the duration of the fourth pulse P4.
[0050] Due to inherent delays in the electronic components used to implement the pulse-detector circuit, the reset of the outputs 124a and 124b of the pulse-detector circuit 120 does not occur instantaneously when both outputs 124a and 124b have been set to the second logic state (1). This means that the pulses P3 and P4 have a guaranteed minimum duration. In FIG. 4, it is the fourth pulse P4 that has this minimum duration (which is t5t4), but depending on the values of Va[n] and Vb[n], it could instead be the third pulse P3 that has the minimum duration. Thus, the durations to be converted by the TDCs 130a and 130b are not shorter than said guaranteed minimum duration. This helps alleviate linearity problems that many TDCs experience when measuring and converting short time durations, e.g. since the TDCs 130a and 130b are operated in a region where the difference between their respective outputs is a relatively linear function of the difference in pulse lengths at their respective inputs, and thus also a relatively linear function of the differential input voltage Va[n]Vb[n]. In this region, any non-linearity in the TDCs 130a and 130b, which would cause problems for shorter pulse durations, influences both TDCs 130a and 130b to the same extent, and thus cancels in the differential domain. It should be noted that the components in the pulse-detector circuit can be deliberately designed or dimensioned to provide a suitable value of said guaranteed minimum delay.
[0051] In some embodiments, the first digital value xa[n] and the second digital value xb[n] form an output sample of the ADC 50. That is, in these embodiments, the output of the ADC 50 is in the form of a vector with two elements, xa[n] and xb[n], i.e. a differential digital signal.
[0052] In other embodiments, the ADC 50 comprises a circuit 150 configured to generate an output sample x[n] of the ADC 50 in response to the first digital value xa[n] and the second digital value xb[n]. An example of this is illustrated in FIG. 5, where the output sample x[n] is the difference between the first digital value xa[n] and the second digital value xb[n]. Hence, the circuit 150 may be a subtraction circuit, or adder circuit with one positive and one negative input. However, other mathematical operations may be involved as well, including for instance gain and offset adjustments.
[0053] FIG. 6 illustrates an embodiment of the pulse-detector circuit 120. In FIG. 6, the pulse-detector circuit 120 comprises a first flip-flop 200a and a second flip-flop 200b, each having a data input D, a clock input clk, a reset input reset, and an output Q. This type of flip-flop is commonly referred to as a D flip-flop. The data input D of each of the first flip-flop 200a and the second flip-flop 200b is configured to receive a constant signal corresponding to the second logic state (1).
[0054] The clock input clk of the first flip-flop 200a is configured to receive the first pulse P1. In FIG. 6, the clock input clk of the first flip-flop 200a is connected to the first input 122a of the pulse-detector circuit 120 for this purpose.
[0055] The clock input clk of the second flip-flop 200b is configured to receive the second pulse P2. In FIG. 6, the clock input clk of the second flip-flop 200b is connected to the second input 122b of the pulse-detector circuit 120 for this purpose.
[0056] Hence, the clock inputs of the flip-flops 200a and 200b are not driven by what would normally be considered as clock signals, but by the pulses P1 and P2, respectively. The term clock input is, however, used since this is common terminology for such inputs in flip-flops.
[0057] Furthermore, in FIG. 6, the output Q of the first flip-flop 200a is connected to the first output 124a of the pulse-detector circuit 120. Moreover, in FIG. 6, the output Q of the second flip-flop 200b is connected to the second output 124b of the pulse-detector circuit 120.
[0058] In the following discussion, the flip-flops 200a and 200b are assumed to be positive edge triggered, i.e. when a rising edge (or positive edge) is presented at their respective clock inputs clk, they transfer the logic value present at their respective data inputs D to their respective outputs Q. Let us further assume that the data outputs Q of both flip-flops 200a and 200b are initially in the first logic state (0) and consider the pulse wave forms in FIG. 4 as an elucidating example. At time instant t1, a rising edge (i.e the start of pulse P1) is presented at the clock input clk of the first flip flop 200a. Since the data input D of the first flip-flop 200a is set to the second logic state (1), the output Q of the first flip-flop 200a makes a transition to the second logic state (1) in response to this rising edge, which results in the start of pulse P3. In FIG. 4, this happens at time instant t3, slightly after t1 due to a slight delay in the first flip-flop 200a. Similarly, at time instant t2, a rising edge (i.e the start of pulse P2) is presented at the clock input clk of the second flip flop 200b. Since the data input D of the second flip-flop 200b is also set to the second logic state (1), the output Q of the second flip-flop 200b makes a transition to the second logic state (1) in response to this rising edge, which results in the start of pulse P4. In FIG. 4, this happens at time instant t4, slightly after t2 due to a slight delay in the second flip-flop 200b.
[0059] In order to reset the first and second outputs 124a, 124b of the pulse-detector circuit 120 to the first logic state (0) in response to both the first output 124a and the second output 124b of the pulse-detector circuit 120 being set in the second logic state (1), embodiment of the pulse-detector circuit 120 illustrated in FIG. 6 comprises a logic circuit 210. The logic circuit 210 has a first input 212a connected to the output Q of the first flip-flop 200a and a second input 212b connected to the output Q of the second flip-flop 200b. Furthermore, the logic circuit 210 has an output 214 connected to the reset inputs reset of the first flip-flop 200a and the second flip-flop 200b for resetting the first flip-flop 200a and the second flip-flop 200b in response to the outputs Q of both the first flip-flop 200a and the second flip-flop 200b being set in the second logic state (1). In FIG. 6, it is assumed that the reset inputs reset of the flip-flops 200a and 200b are active high, i.e. that the flip-flops 200a and 200b are reset when a 1 is presented at their respective reset inputs reset. Then, the functionality of the logic circuit 210 as described above can be implemented with an AND gate, as illustrated in FIG. 6. If, on the other hand, the reset inputs reset of the flip-flops 200a and 200b are active low, i.e. the flip-flops 200a and 200b are reset when a 0 is presented at their respective reset inputs reset, the functionality of the logic circuit 210 as described above can be implemented with a NAND gate. Other types of implementations can be used in other scenarios, e.g. if the first logic state would be 1 and the second logic state would be 0, etc.
[0060] An inherent delay in the logic circuit 210 and the flip-flops 200a and 200b provides the above-mentioned guaranteed minimum duration of the pulses P3 and P4. In order to provide a desired length of said guaranteed minimum duration, different design techniques can be used in the design of the logic circuit 210, as would be appreciated by a person skilled in the art of electronic design. For instance, with reference to FIG. 6, the AND gate can be intentionally designed to be slow, e.g. using a relatively small width-over-length (W/L) ratios in the transistors in the AND gate, using current-starving techniques, and/or using a relatively high capacitive load at the output of the AND gate, etc. Alternatively or additionally, delay elements may be connected to the inputs and/or the output of the AND gate.
[0061] In some embodiments, the logic circuit 210 is further configured to reset the first flip-flop 200a and the second flip-flop 200b in response to a reset pulse. This can be useful if, for instance, the first or second flip-flop is stuck in an erroneous state, e.g. due to noise or an input signal outside the intended input range of the ADC 50, such that the standard reset mechanism based on both outputs 124a and 124b of the pulse-detector circuit being in the second logic state (1) is not triggered for one or more samples. Then, by including circuitry (not shown in the drawings) in the ADC 50 configured to provide said reset pulse in preparation of converting each sample, it can be guaranteed that both outputs 124a and 124b of the pulse-detector circuit are initially in the first logic state (0) for each sample. An example of such an embodiment of the logic circuit 210 is illustrated in FIG. 7. In FIG. 7, the logic circuit 210 comprises the AND gate shown in FIG. 6 and an OR gate. Instead of having the output of the AND gate connected to the reset inputs reset of the flip-flops 200a and 200b, the output of the AND gate is instead connected to a first input of the OR gate, and the output of the OR gate is connected to the reset inputs reset of the flip-flops 200a and 200b. A second input of the OR gate is connected to a third input 212c of the logic circuit 210. The third input 212c is configured to receive said reset pulse. With this configuration, the flip-flops 200a and 200b will be reset if said reset pulse is provided at the third input 212c or a 1 is provided at the output of the AND gate in the logic circuit 210.
[0062] According to some embodiments, the first VTC 110a comprises a first capacitor 250a configured to, for each sample of the first input voltage Va[n], be charged to the voltage value of that sample of the first input voltage Va[n] during a first phase of operation (for that sample). Furthermore, according to some embodiments, the first VTC 110a comprises a first current source 260a configured to discharge the first capacitor 250a during a subsequent second phase of operation (for that sample). Moreover, according to some embodiments, the first VTC 110a comprises a first comparator circuit 270a configured to compare the voltage Vca across the first capacitor 250a with a first reference voltage Vrefa and to generate the first pulse P1 at an output of the first comparator circuit 270a. An example of such a configuration is illustrated in FIG. 8, where the first reference voltage Vrefa is provided to the positive input of the first comparator circuit 270a. In FIG. 8, one terminal of the first capacitor 250a is connected to a signal ground node. Furthermore, in FIG. 8, the other terminal of the first capacitor 250a is connected to a negative input of the first comparator circuit 270a. Moreover, in FIG. 8, a sampling switch 280a is connected between the input terminal 112a of the first VTC 110a and the negative input terminal of the first comparator circuit 270a. A sampling control signal smp_p controls the sampling switch 280a such that, for each sample, the sampling switch 280a is closed during said first phase of operation and open during said second phase of operation. The sampling control signal smp_p can also be used as the above-mentioned reset pulse provided to the third input 212c (FIG. 7) of the logic circuit 210 in some embodiments. For the embodiment illustrated in FIG. 8, the voltage Va provided at the input terminal 112a of the first VTC 110a can be either time continuous or time discrete (e.g. piecewise constant). In the case of a time-continuous voltage Va, said sample Va[n] is the value of Va at the end of the first phase of operation.
[0063] In FIG. 8, the first current source 260a is connected in series with a switch 290a between the negative input of the first comparator circuit 270a and the signal ground node. A control signal smp_m controls the switch 290a such that, for each sample, the switch 290a is open during said first phase of operation and closed during said second phase of operation. Thereby, during the second phase of operation, the first capacitor 250a is gradually discharged and the voltage Vca across the first capacitor is thus gradually reduced. When it reaches below the first reference voltage Vrefa, a transition is made at the output of the first comparator circuit 270a, which is the start of the first pulse P1.
[0064] The operation of the first VTC as described above is illustrated with some waveforms in FIG. 9 for the sample Va[n]. The first phase of operation corresponds to the time interval during which the sampling control signal smp_p is 1. In FIG. 9, the sampling-time instant nT is defined as the end of the first phase of operation, i.e. at the falling edge of smp_p. The second phase of operation corresponds to the time interval during which the control signal smp_m is 1. In FIG. 9, the first phase of operation and the second phase of operation are non-overlapping, with some short guard time intervals between the first phase of operation and the second phase of operation, and between the second phase of operation and the first phase of operation for the next sample.
[0065] For simplicity of illustration, it is assumed in FIG. 9 that Va is constant during the first phase of operation, during which the voltage Vca across the first capacitor 250a is set to Va[n]. During the second phase of operation, the voltage Vca is decreased, ideally at a constant rate, and when it crosses Vrefa, a transition is made at the output of the first comparator circuit 270a, which is the start of the first pulse P1, as mentioned above.
[0066] At the next rising edge of smp_p, the first capacitor 150a is charged to the next sample Va[n+1] of Va. This naturally resets the output of the first comparator circuit 270a and ends the first pulse P1 in preparation for processing of the next sample Va[n+1].
[0067] According to some embodiments, the second VTC 110b has a similar or identical design as the first VTC 110a. For example, according to some embodiments, the second VTC 110b comprises a second capacitor 250b configured to, for each sample of the second input voltage Vb[n], be charged to the voltage value of that sample of the second input voltage Vb[n] during the first phase of operation. Furthermore, according to some embodiments, the second VTC 110b comprises a second current source 260b configured to discharge the second capacitor 250b during the second phase of operation. Moreover, according to some embodiments, the second VTC 110b comprises a second comparator circuit 270b configured to compare the voltage Vcb across the second capacitor 250b with the first reference voltage Vrefa or a second reference voltage Vrefb and to generate the second pulse P2 at an output of the second comparator circuit 270b. Using the same reference voltage Vrefa in both the first VTC 110a and the second VTC 110b has the benefit of a relatively simple design and no need to generate multiple reference voltages for the comparator circuits 270a and 270b. Using different reference voltages Vrefa and Vrefb in the first VTC 110a and the second VTC 110b has the benefit of providing more degrees of freedom in calibrating the two VTCs 110a and 110b, e.g. to compensate for a mismatch between the comparator circuits 270a and 270b.
[0068] An example of such a configuration of the second VTC 110b is illustrated in FIG. 10, where the first reference voltage Vrefa or the second reference voltage Vrefb is provided to the positive input of the second comparator circuit 270b. In FIG. 10, one terminal of the second capacitor 250b is connected to the signal ground node. Furthermore, in FIG. 10, the other terminal of the second capacitor 250b is connected to a negative input of the second comparator circuit 270b. Moreover, in FIG. 10, a sampling switch 280b is connected between the input terminal 112b of the second VTC 110b and the negative input terminal of the second comparator circuit 270b. The sampling control signal smp_p controls the sampling switch 280b a such that, for each sample, the sampling switch 280b is closed during said first phase of operation and open during said second phase of operation. As above, for the embodiment illustrated in FIG. 10, the voltage Vb provided at the input terminal 112b of the second VTC 110b can be either time continuous or time discrete (e.g. piecewise constant). In the case of a time-continuous voltage Vb, said sample Vb[n] is the value of Vb at the end of the first phase of operation.
[0069] In FIG. 10, the second current source 260b is connected in series with a switch 290b between the negative input of the second comparator circuit 270b and the signal ground node. The control signal smp_m controls the switch 290b such that, for each sample, the switch 290b is open during said first phase of operation and closed during said second phase of operation. Thereby, during the second phase of operation, the second capacitor 250b is gradually discharged and the voltage Vcb across the second capacitor is thus gradually reduced. When it reaches below the first reference voltage Vrefa or the second reference voltage Vrefb (i.e. the one of Vrefa and Vrefb that is actually used in the embodiment), a transition is made at the output of the second comparator circuit 270b, which is the start of the second pulse P2.
[0070] In FIGS. 8 and 10, the current sources 260a and 260b are coupled, via the switches 290a and 290b, respectively, to the same signal ground node as the capacitors 250a and 250b. However, in other embodiments, they may be coupled to different nodes than the signal ground node that the capacitors 250a and 250b are connected to. Furthermore, in some embodiments, the polarity of the current sources 260a and 260b may be reversed, so that they are configured to charge the capacitors 250a and 250b instead. Then the voltages Vca and Vcb will be gradually increased instead of being gradually reduced during the second phase of operation, and will pass through the respective reference voltages Vrefa or Vrefb from below rather than from above. The general principle of operation is the same for these embodiments with reversed polarity compared with what is illustrated in the drawings.
[0071] In the following, a particular embodiment of the ADC 50 is described. According to this embodiment, the ADC 50 comprises the first and the second VTC 110a, 110b, each having an input terminal 112a, 112b and an output terminal 114a, 114b. Each of the first VTC 110a and the second VTC 110b comprises a sampling capacitor (corresponding to the first capacitor 250a in FIG. 8 and the second capacitor 250b in FIG. 10, respectively) having a first terminal and a second terminal, wherein the second terminal is connected to a signal ground node. Furthermore, each of the first VTC 110a and the second VTC 110b comprises a sampling switch (corresponding to the sampling switch 280a in FIG. 8 and the sampling switch 280b in FIG. 10, respectively) connected between the input terminal 112a, 112b of the VTC 110a, 110b and the first terminal of the sampling capacitor 250a, 250b. Moreover, each of the first VTC 110a and the second VTC 110b comprises a charge-transfer circuit comprising a series connection of a current source (corresponding to the first current source 260a in FIG. 8 and the second current source 260b in FIG. 10, respectively) and a switch (corresponding to the switch 290a in FIG. 8 and the switch 290b in FIG. 10, respectively) connected to the first terminal of the capacitor 250a, 250b (e.g. between the first terminal of the capacitor 250a, 250b and the signal ground node as in FIGS. 8 and 10, or between the first terminal of the capacitor 250a, 250b and some other node, as discussed above). Each of the first VTC 110a and the second VTC 110b further comprises a comparator circuit (corresponding to the first comparator circuit 270a in FIG. 8 and the second comparator circuit 270b in FIG. 10, respectively) having a first input terminal connected to the first terminal of the capacitor 250a, 250b, a second input terminal configured to receive a reference voltage, and an output terminal connected to the output terminal 114a, 114b of the VTC 110a, 110b. As described above, said reference voltage may be the same (Vrefa) in both VTCs 110a and 110b, or different (Vrefa in the first VTC 110a and Vrefb in the second VTC 110b). According to the embodiment, the ADC 50 further comprises a pulse-detector circuit 120 as described above with reference to FIGS. 6 and 7. The pulse-detector circuit 120 comprises the first flip-flop 200a and the second flip-flop 200b, each having a data input D, a clock input clk, a reset input reset, and an output Q. The output Q of each of the first and second flip-flop 200a, 200b can be in the first logic state (0) or the second logic state (1). Each of the first and the second flip-flop 200a, 200b is configured to reset its output Q to the first logic state (0) in response to a reset signal at its reset input reset. Furthermore, the pulse-detector circuit 120 comprises the logic circuit 210 having the first input 212a, the second input 212b, and the output 214. The data input D of each of the first flip-flop 200a and the second flip-flop 200b is configured to receive a constant signal corresponding to the second logic state (1). The clock input clk of the first flip-flop 200a is connected to the output terminal 114a of the first VTC 110a. The clock input clk of the second flip-flop 200b is connected to the output terminal 114b of the second VTC 110b. The output Q of the first flip-flop 200a is connected to the first output 124a of the pulse-detector circuit 120. The output Q of the second flip-flop 200b is connected to the second output 124b of the pulse-detector circuit 120. The first input 212a of the logic circuit 210 is connected to the output Q of the first flip-flop 200a. The second input 212b of the logic circuit 210 is connected to the output Q of the second flip-flop 200b. The output 214 of the logic circuit 210 is connected to the reset inputs reset of the first and the second flip-flop 200a, 200b. The logic circuit 210 is configured to generate the reset signal at its output 214 in response to both of its first input 212a and its second input 212b being set in the second logic state (1). In addition, as described above with reference to FIG. 7, the logic circuit 210 may have the third input 212c configured to receive said reset pulse (which may e.g. be the same as the sampling control signal smp_p described above), and may be configured to generate the reset signal at its output 214 also in response to receiving the reset pulse at its third input 212c. Furthermore, according to the embodiment, the ADC 50 also comprises the first TDC 130a connected to the first output 124a of the pulse-detector circuit 120 and the second TDC 130b connected to the second output 124b of the pulse-detector circuit 120.
[0072] There are various options for implementing the TDCs 130a and 130b in the embodiments disclosed herein. For example, oscillator-based TDCs or Vernier-type TDCs may be used. According to some embodiments, each of the first TDC 130a and the second TDC 130b is a pulse-shrinking TDC. In a pulse-shrinking TDC, the pulse whose duration is to be measured (below referred to as the pulse to be measured) is fed into a chain of pulse-shrinking elements. Each of the pulse-shrinking elements receives an input pulse and outputs an output pulse that has duration that is reduced with a certain amount of time compared with the input pulse. This can e.g. be accomplished by designing the pulse-shrinking elements, or components therein, to have different rise and fall times. The output pulse is input to a next pulse-shrinking element in the chain as an input pulse (except for the last pulse-shrinking element in the chain). As the pulse propagates along the chain of pulse-shrinking elements, the pulse duration will become shorter and shorter, and at some point, the pulse will vanish (unless the pulse to be measured is long enough to allow the pulse to propagate through all elements in the chain). The number of pulse-shrinking elements that the pulse propagates through before vanishing is a metric of the duration of the pulse to be measured.
[0073] An example of a pulse-shrinking TDC 130 is shown in FIG. 11. The reference number 130 is used in FIG. 11 as a generic reference number representing both the first TDC 130a and the second TDC 130b (i.e. they can both be implemented in the same way). In the same way, the reference number 132 is used in FIG. 11 as a generic reference number representing both the input 132a of the first TDC 130a and the input 132b of the second TDC 130b. Similarly, the reference number 134 is used in FIG. 11 as a generic reference number representing both the output 134 of the first TDC 130a and the output 134b of the second TDC 130b.
[0074] In FIG. 11, the output 134 has N bits 134_1-134_N. Furthermore, the pulse-shrinking TDC 130 in FIG. 11 comprises N flip-flops of the type described in the context of the embodiments of the pulse-detector circuit 120 illustrated in FIGS. 6-7. Moreover, in FIG. 11, the pulse-shrinking TDC 130 comprises a chain of N pulse-shrinking elements 350_1-350_N. The output Q of each flip-flop is connected to a unique one of the N bits 134_1-134_N of the output 134 of the pulse-shrinking TDC 130. Furthermore, the clock input clk of each flip-flop is connected to the output of a unique one of pulse-shrinking elements 350_1-350_N. Moreover, the reset input reset of all flip-flops is driven by the sampling-control signal smp_p, whereby, for each sample, the outputs Q of all flip-flops are initially reset to the first logic state (0). The data inputs D of all flip-flops are set to the second logic state (1). As the pulse provided at the input 132 (from the pulse-detector circuit 120) propagates along the chain of pulse-shrinking elements 350_1-350_N, the outputs Q of a number of flip-flops, namely those whose clock inputs are connected to outputs of the pulse-shrinking elements that the pulse propagates through before vanishing, will be set to the second logic state (1). The output bits 134_1-134_N thus provide a thermometer-coded digital representation of the duration of the pulse provided at the input 132. Pulse-shrinking TDCs are known in the art of time-to-digital conversion, and the functionality is not described in any further detail herein.
[0075] An advantage of pulse-shrinking TDCs is that they have a relatively low energy consumption. For most pulse-durations, only a subset of the flip-flops will be reached by a pulse at its clock input clk and contribute to the dynamic energy consumption of the TDC. However, the inventors have identified the pulse-shrinking TDC as one of the types of TDCs that suffer from relatively poor linearity when the duration to be converted is relatively short, as mentioned above. This problem is alleviated with the use of the pulse-detector circuit 120, which, as mentioned above, provides a guaranteed minimum duration of the pulses that are input to the TDCs 130a and 130b.
[0076] The embodiments of the ADC 50 described herein are suitable for integration on an integrated circuit (IC). Thus, according to some embodiments, there is provided an IC 500 comprising the ADC 50, as schematically illustrated in FIG. 12.
[0077] The disclosure above refers to specific embodiments. However, other embodiments than the above described are possible within the scope of the invention. For example, the circuitry used in embodiments of the pulse-detector circuit 120 shown in FIGS. 6 and 7 can also be used as phase detectors in phase-locked loop (PLL). Other types of circuitry that are used in phase-detectors in PLLs may thus be used to implement the pulse-detector circuit 120 in embodiments of the ADC 50. Furthermore, the ADC 50 may be used in other types of electronic apparatuses than communication apparatuses. The different features of the embodiments may be combined in other combinations than those described.
