Abstract
A method of characterizing frequency fluctuations of a resonator comprising the steps of: a) driving the resonator, in a linear regime, by simultaneously applying two periodical driving signals having respective frequencies, the frequencies being different from each other and from a resonant frequency of the resonator, but contained within a resonance linewidth thereof; b) performing simultaneous measurements of response signal of the resonator at the frequencies of the periodical driving signal; and c) computing a value representative of a correlation between the measurements, the value being indicative of frequency fluctuations of the resonator. An apparatus for implementing such a method is provided.
Claims
1. A method of characterizing frequency fluctuations of a resonator having a resonant frequency f.sub.0, comprising the steps of: a) driving the resonator by simultaneously applying two periodical driving signals having respective frequencies f.sub.1, f.sub.2, said frequencies being different from each other, but contained within a resonance linewidth of the resonator; b) performing simultaneous measurements of response signals of said resonator at the frequencies f.sub.1, f.sub.2, of said periodical driving signals; and c) computing a value representative of a correlation between said measurements, said value being indicative of frequency fluctuations of the resonator.
2. The method of claim 1 wherein step a) comprises driving the resonator in a linear regime.
3. The method of claim 1 wherein the frequencies f.sub.1 and f.sub.2 are both different from the resonant frequency f.sub.0.
4. The method of claim 1, wherein step b) comprises measuring time-varying phases of said response signals at the frequencies f.sub.1, f.sub.2 of said periodical driving signals.
5. The method of claim 4, further comprising a step b′), carried out before step c), of performing filtering of said time-varying phases.
6. The method of claim 5 wherein said filtering is a band-pass filtering.
7. The method of claim 4, wherein: step b) comprises performing a plurality of measurements of the time-varying phases of the response signals at said frequencies f.sub.1, f.sub.2 using different integration times τ.sub.1; the method further comprises a step of performing band-pass filtering of each measured time-varying phase using a filter whose bandwidth is centered on a frequency inversely proportional to the respective integration time; and step c) comprises computing a value representative of a correlation between band-pass filtered time-varying phases of said spectral components of the resonance signal for each integration time.
8. The method of claim 7 further comprising: a step d) of using the value computed during step c) for determining a range of integration times wherein the response signal of the resonator is dominated by frequency fluctuations thereof; and a step e) of performing a feedback-loop control on the resonant frequency of the resonator within a frequency range corresponding to said range of integration times.
9. The method of claim 4, wherein step c) comprises converting the measured time-varying phases to time-varying frequency values using a frequency-phase relationship of said resonator, and computing a correlation thereof.
10. The method of claim 4, further comprising performing closed-loop control of the frequencies of the periodical driving signals using the respective measured time-varying phases as feedback signals, and wherein step c) comprises computing a correlation of said frequencies.
11. The method of claim 1 wherein said simultaneous measurements are performed by heterodyne detection.
12. The method of claim 1 wherein said simultaneous measurements are performed by homodyne detection.
13. The method of claim 1 wherein said resonator is a MEMS and/or NEMS.
14. An apparatus for characterizing frequency fluctuations of a resonator comprising: a driving signal generation, configured for simultaneously generating at least the two periodical driving signals at different frequencies f.sub.1, f.sub.2, said frequencies being different from each other, but contained within a resonance linewidth of the resonator; and applying them to the resonator; at least one sensing device, configured for performing simultaneous measurements of response signal of said resonator at the frequencies f.sub.1, f.sub.2 of said periodical driving signals; and a signal processor, configured for computing a value representative of a correlation between said measurements, said value being indicative of frequency fluctuations of the resonator.
15. The apparatus of claim 14, wherein the sensing device is configured for measuring time-varying phases of said response signals at the frequencies of said periodical driving signals.
16. The apparatus of claim 15, wherein the signal processor is configured for performing filtering of said time-varying phases.
17. The apparatus of claim 16 wherein said filtering is a band-pass filtering.
18. The apparatus of claim 15, wherein: the sensing device is configured for performing a plurality of measurements of the time-varying phases of said response signals at the frequencies of said periodical driving signals using different integration times τ.sub.1; the signal processor is configured for performing band-pass filtering of each measured time-varying phase using a filter whose bandwidth is centered on a frequency inversely proportional to the respective integration time; and for computing a value representative of a correlation between band-pass filtered time-varying phases of said spectral components of the resonance signal for each integration time.
19. The apparatus of claim 18 wherein the signal processor is further configured for determining a range of integration times wherein the response signal of the resonator is dominated by frequency fluctuations thereof, the apparatus further comprising a feedback loop controller configured for controlling the resonant frequency of the resonator within a frequency range corresponding to said range of integration times.
20. The apparatus of claim 15, wherein the signal processor is configured for converting the measured time-varying phases to time-varying frequency values using a frequency-phase relationship of said resonator, and computing a correlation thereof.
21. The apparatus of claim 15 further comprising two feedback loops configured for performing closed-loop control of the frequencies of the periodical driving signals using the respective measured time-varying phases as feedback signals, and wherein the signal processor is configured for computing a correlation of said frequencies.
Description
BRIEF DESCRIPTION OF THE DRAWINGS
[0041] A more complete understanding of the present disclosure thereof may be acquired by referring to the following description taken in conjunction with the accompanying drawings wherein:
[0042] FIG. 1 is a generic heterodyne apparatus for M-NEMS resonators frequency response measurement.
[0043] FIG. 2 is a schematic flow diagram illustrating steps of a method to distinguish additive noise from frequency fluctuations and to implement a compensation apparatus of the latter, according to a specific embodiment of the present invention.
[0044] FIG. 3 is the adaptation of FIG. 1 setup for simultaneous measurement of the two phase traces. Representations are according to a specific embodiment of the present invention.
[0045] FIG. 4 is the representation of the homodyne correlation measurement setup according to a specific embodiment of the present invention.
[0046] FIG. 5 top windows are representations of additive phase noise and frequency fluctuations effects on the magnitude spectrum of the resonance. Bottom windows show the phase response to two excitation frequencies around the resonance during phase spectrum observation of the resonance. Representations are according to a specific embodiment of the present invention.
[0047] FIG. 6 upper part shows the computation of the Allan deviation while the bottom part shows the result of the correlation technique used in the present invention to distinguish frequency fluctuations from additive phase noise.
DETAILED DESCRIPTION
[0048] While the present invention is susceptible to various modifications and alternative forms, specific example embodiments thereof have been shown in the drawings and are herein described in detail. It should be understood, however, that the description herein of specific example embodiments is not intended to limit the disclosure to the particular forms disclosed herein, but on the contrary, this disclosure is to cover all modifications and equivalents as defined by the appended claims.
[0049] FIG. 1 is a generic schematic representation of the experimental setup used for detecting the resonant motion of a NEMS in prior art “In-plane nanoelectromechanical resonators based on silicon nanowire piezoresistive detection”, E. Mile et al. Nanotechnology, 2010, vol. 21 no 16, p. 165504. The NEMS 107 is composed by a fixed-free lever beam 115 and two piezoresistive gauges 113 and 114 anchoring the lever. The actuation of the NEMS is performed with the pad 112 by means of a signal at angular frequency ω/2 coming from the drive VCO (Voltage Controlled Oscillator) 102 and PS (Power Splitter) 103′. The drive signal is set to the angular frequency ω/2 which is half of the actuation frequency in order to reduce parasitic signals. The two gauges 113 and 114 are receiving a signal that passed through the phase shifter 106 such as their respective signals have a relative phase of 180°. The phase shifter incoming signal is generated by the bias VCO 101. The bias signal's angular frequency is ω-Δω where ω is the actuation frequency and Δω an arbitrary readout angular frequency. The piezoresistive gauges behave such as they convert the stress applied to the lever into a resistance variation. They alternatively operate in tensile and compressive strain allowing an improved dynamic range. Meaning there is an enhanced rejection of the background. The heterodyne architecture is performing a 2ω down-mixing operation with notably, the mixer 105 and the frequency doubler 104 thus the term 2ω. The output of the mixer 105 is filtered with the LPF (Low Pass Filter) 108 to only keep the information at angular frequency Δω. Requirement for such architecture is explained by the need to be far away from the cut off frequency formed by the NEMS parasitic capacitance, the coaxial cable 110 capacitance 111 and the readout instrument Lock-In 109 input impedance. The 2ω down-mixing also improves cross-talk.
[0050] FIG. 2 is a flow diagram showing the major steps of an exemplary embodiment of a method in accordance with the present invention. At step a″ of the method of FIG. 1 a resonator is selected, i.e., the resonator may be electronic, mechanical, optical or any combination of the three. The method described herein is applied to a NEMS or MEMS resonator and is as follows: [0051] At step a′, the resonator parameters such as the quality factor Q and the resonant frequency f.sub.0 are measured. In another embodiment, said resonator parameters may be obtained from the datasheet, the manufacturer or any other means. [0052] Step a consists in driving the resonator, preferably but not necessarily in a linear regime, with two periodical signals having respective frequencies f.sub.1 and f.sub.2. The excitation frequencies are different from each other. These frequencies are within a certain linewidth when compared to the resonant frequency: the choice of the two excitation frequencies is defined such as they are sufficiently close to the resonance; one of them may indeed coincide with the resonant frequency f.sub.0. In one embodiment of the invention, the choice of the excitation frequencies is such as they are chosen to be contained in a frequency range that is within 1% of the resonant frequency. This guarantees a steep, and almost linear, phase spectral response of the resonator, where its measurement is the most sensitive. [0053] Step b consists in performing a plurality of measurements of the time-varying phase spectral response to said frequency excitations with simultaneous excitation of the resonator at frequencies f.sub.1 and f.sub.2. A closed-loop control of the frequencies of the periodical driving signals uses the respective time-varying phases as feedback signals. Said responses are measured and integrated for samples of duration Said excitations and responses are illustrated in FIG. 5. [0054] An optional step b′ consists in filtering the time-varying phases occurs. It allows keeping relevant information for the computation of a value representative of the data correlation. Other embodiments may not include filtering. In such case any frequency fluctuations contribution will appear on the data. Among them, frequency drift caused by temperature fluctuations which is a deterministic slow-varying process and may mislead data interpretation. [0055] Step b, possibly supplemented by the filtering step b′, results in a pair of resonant frequency traces f.sub.1 and f.sub.2, each of length N. [0056] Step c is the computation of a value representative of the data correlation; said correlation being indicative of frequency fluctuations of the resonator.
[0057] Different types of band-pass filtering can be performed on the data. In one embodiment of the invention, the filtering of the data is performed with the band-pass filter of the Allan deviation, of the form:
[00002]
where f is the center frequency of the band-pass filter, said center frequency f being inversely proportional to the integration time τ.sub.1.
[0058] Use of a low-pass or high-pass filter is also possible, albeit less preferred.
[0059] The computation of the value representative of the correlation of traces can be determined using a variety of methods. These include, by way of example (and not of a limiting nature), the Pearson's correlation coefficient and the Spearman's rank correlation coefficient.
[0060] A step c of computing a value representative of a correlation between band-pass filtered measurements of time-varying phase values of spectral component for each integration time.
[0061] In one embodiment of the invention the method to measure the correlation is the Pearson's correlation coefficient, which is defined as:
[00003]
where f.sub.1 and f.sub.2 are sample means of f.sub.1 and f.sub.2 respectively, S.sub.f1 S.sub.f2 are their standard deviations and N is representing the number of data points. If filtering step is performed, equation. 2 is applied for each integration time τ.sub.1. The apparatus used to perform the correlation is shown in FIG. 3.
[0062] A step d of determining a range of integration times wherein response signal of the resonator is dominated by frequency fluctuations. This is the first step towards enabling frequency fluctuations compensation by utilizing time-varying frequency values of the correlation to discriminate regions where frequency fluctuations are not dominating. This steps occurs if a filtering step was performed.
[0063] A step e of performing a feedback-loop control on the resonant frequency of the resonator within a frequency range corresponding to said range of integration times of step d. This further and terminates the compensation of the frequency fluctuations.
[0064] FIG. 3 illustrates one embodiment of the object of the invention with the apparatus used for the detection of two time-varying phase traces. It consists of a driving signal generator configured for the simultaneous generation of two driving signals 201 and 202 which are respectively V.sub.Drive,1 and V.sub.Drive,2 and associated to the frequencies f.sub.1 and f.sub.2. The two driving signals are then summed by the adder 203 in order to actuate the resonator. The two driving signals are used to actuate the resonator 207 on the side-gate 212. Similarly to the driving signals, two signals are generated to bias the gauges 213 and 214. V.sub.Bias,1 and V.sub.Bias,2 are respectively generated by signal generators 204 and 205. Bias signals are also summed by the adder 206 before going through the phase shifter 208. The phase shifter sets a relative phase difference of 180° between the two bias signals. f.sub.corr represents the separation of each measurement signal from the resonant frequency while Δf.sub.1 and Δf.sub.2 are the frequencies of the output signals of the resonator. These frequencies are different in order to be readout by the sensing device. Said sensing device 209 is then used for performing simultaneous measurements of the resonator at two the different frequencies. Wherein sensing device is also configured for performing a plurality of measurements of time-varying phases of both spectral components of the resonance signal using different integration time τ.sub.1. In another embodiment, the detection apparatus may contain more than one sensing device. In the present invention, the sensing device is a Lock-In Amplifier.
[0065] A signal processor 210 is for computing a value representative of a correlation between said measurements, with a value being indicative of the frequency fluctuations of the resonator. A signal processor includes, by way of example (and not of a limiting nature), a DSP (Digital Signal Processing) pertinently programmed. Said signal processor is also performing band-pass filtering of said time-varying phases. Wherein signal processor is performing band-pass filtering of the measured time-varying phase signals with a filter centered on a frequency inversely proportional to their respective integration time τ.sub.1. The signal processor in one embodiment of the present invention is configured to compute a representative value of the correlation between band-pass filtered time-varying phases for each integration time.
[0066] In another embodiment, the signal processor is configured to determine a range of integration times wherein the response of the resonator's signal is dominated by frequency fluctuations. The signal processor is further configured to convert measured time-varying phases into time-varying frequency values, and then compute said correlation.
[0067] In another embodiment the correlation measurement is performed with a homodyne apparatus. FIG. 4 is the representation of the homodyne apparatus configured for the measurement of the resonator when driven at f.sub.0+f.sub.corr and f.sub.0−f.sub.corr and reading the output at the same frequency. Then a selective filter is used for the separation of the two measurement signals. Said filter is for example a digital filter processing data either in real time or post-processing. When compared to the apparatus of FIG. 3, the homodyne setup is also composed by two generating driving signals 301 and 302 that are summed with the adder 303 and actuate the resonator 307 side-gate 304. The main difference between the two measurement systems concerns the bias 305 and 306 voltage signals of the gauges 308 and 309. Biases are DC (Direct Current) voltage signals and are directly connected to the gauges. The constant bias voltage results in the generation of the response signals at the actuation frequency (i.e. homodyne readout). The sensing device 310 remains identical to the one of FIG. 3.
[0068] FIG. 5 is a representation of the effects of additive phase noise sources on the resonances 405 and 411. Top windows are representing the spectral magnitude of the resonance 405 with superimposition of the effect of additive white noise and the effects of frequency fluctuations on the spectral magnitude of the resonance 411. The effect of said noise is the manifestation of spurs 413 on the amplitude of the resonance, while frequency fluctuations effects are shifts 414 of the resonance along an axis parallel to the frequency axis. For additive white noise, excitation frequencies mentioned in the description of FIG. 2 are represented by 401 and 402 along with their respective time-varying phase responses 403 and 404. In this exemplary embodiment, the excitation frequencies are on both sides of the resonant frequency 406 but this is not essential—they could as well be on the same side of the resonance peak. Such excitation frequencies are located in the linear region of the phase spectral response of the resonator, where its sensitivity is the highest. On the one hand, additive white noise is an uncorrelated process, thus its associated time-varying phase responses are uncorrelated. On the other hand frequency fluctuations processes are correlated and therefore their associated time-varying phase responses are correlated too. This is one of the reasons for which the interpretation of the correlation is easier if the resonator is driven in its linear regime.
[0069] FIG. 6 is a representation of both the Allan deviation 501 and the result of the filtered correlation method 502 used in the present invention. The plot is divided in three regions of noise source dominant contribution. The first region is dominated by additive phase noise source 505, while the third region is dominated by frequency fluctuations noise source 507. Between these two regions, an intermediate region 506 is the contribution of a combination of the first and third noise sources. At 1 second integration time 503, the Allan deviation shows a frequency stability of about 8.10.sup.−6 504 but it does not provide any information on the origin of the noise source contributing to the frequency stability degradation. The correlation method on the other hand does not quantify the frequency stability, but it shows the frequency stability is limited by the resonator's frequency fluctuations 507. Shorter integration time tends to show effects of rapidly-varying phenomena such as for example additive white noise. This would occur at high Fourier frequencies of the phase spectral density, thus the thermal floor or so-called white noise. In contrast, regions of longer integration time suggest phenomena occurring at lower Fourier frequencies, for example near the carrier of a phase spectral density measurement, therefore near the resonance. The Allan deviation and the correlation method of the present invention are complementary as they provide a diagnostic tool to both quantify frequency stability but also identify the incriminated noise source.
[0070] While embodiments of this disclosure have been depicted, described, and are defined by reference to example embodiments of the disclosure, such references do not imply a limitation on the disclosure, and no such limitation is to be inferred. The subject matter disclosed is capable of considerable modification, alteration, and equivalents in form and function, as will occur to those ordinarily skilled in the pertinent art and having the benefit of this disclosure. The depicted and described embodiments of this disclosure are examples only, and are not exhaustive of the scope of the disclosure. The true scope and spirit of the invention is indicated by the following claims.
