Programmable analog beamformer

Abstract

A programmable analog beamformer controls phase and amplitude of radio frequency sine signals independently on n channels. In a preferred embodiment, each of n channels achieves full cycle phase sifting by using m first order programmable filters isolated by buffer amplifiers, with maximum phase shift amount of 180/m degrees in each filter. By flipping the polarity of sine signal in differential path, the beamformer achieves additional 180 degrees phase shift. There is an amplitude control unit in each channel, that both compensates amplitude attenuations due to phase shifting of filters, and to control the amplitude of the RF signal per user choice. There is a core algorithm software that handles all the digital programming of the system, as well as error correction of the phase and amplitude. The analog beamformer can drive piezoelectric ultrasonic transducers with no unwanted harmonics, or other loads per application.

Claims

1. A programmable analog beamformer device comprising: at least two independent user programmable logic controlling input radio frequency channels generating a single frequency sine signal driven by user selected phase and amplitude inputs; each channel having components serially connected from a buffer circuit to a phase-shifter, an amplitude control unit, a power amplifier and a load to a ground: each phase-shifter having a first order filter isolated by a buffer amplifier at input and output to maintain filters at first order; each load carrying half of the actual impendence in differential power delivery and complete impedance with a single ended power delivery, and each ground being a virtual ground wherein the differential power is zero voltage of a beamformer circuit and a real ground of the beamformer circuit where a single ended power delivers piezo electric loads for ultrasound transducer output whereby a user selected phase and amplitudes independently in an array of radio frequency parallel channels can focus ultrasound beams via transducer output onto a target.

2. A programmable analog beamformer device as in claim 1 further comprising phase shifting using filter delays to an RF signal from 0 degrees to 180/m degrees, phase shifting in each by programmable coupling a digital potentiometer resistor with a programmed capacitor forming a single pole filter for a phase delay.

3. A programmable analog beamformer device as in claim 1 further comprising additional phase shifting by conducting the RF sine signal in a channel through parallel inverting amplifier and non-inverting amplifiers and downstream serially to a RF switch with at least 3 dB bandwidth with respect to the frequency of operation.

4. A programmable analog beamformer device as in claim 1 further comprising phase shifting using filter amplitude attenuation compensation responsive to digital logic responsive to a channel filter capacitor, the channel signal frequency of operation and the channel filter resistance, amplitude compensation metered and provided by an amplitude control unit.

5. A programmable analog beamformer device as in claim 1 further comprising the amplitude control unit controlling the channel signal amplitude through a current feedback amplifier and a digital potentiometer in its feedback network that changes the gain and hence the channel signal amplitude.

6. A programmable analog beamformer device as in claim 1 further comprising the amplitude control unit operatively connected to user interface error correction logic responsive to changes in the gain of amplifier errors.

7. A programmable analog beamformer device as in claim 1 further comprising loads for piezoelectric transducers in ultrasound applications with standard impedance of 50 ohm at resonance frequencies, providing electric power to the loads single ended or differentially, wherein the loads are half of the actual impedance in differential power delivery, and complete impedance in case of single ended power delivery.

8. A programmable analog beamformer device as in claim 1 further comprising the first order filter having input from a buffer amplifier at input and buffer amplifier at output, acting as isolators to maintain a single pole of the filter with a decoupling capacitor between the input and output amplifiers, the filter having bias resistors, a programmable digital potentiometer resistor R and a capacitor C before the input to the output amplifier buffer for adjusting the filter R*C time constant.

9. A programmable analog beamformer device as in claim 1 further comprising control of the base shifting amount, with digital logic based on the component and frequency values resulting in a base shift of angle equal to arc tan(2*.Math.pi.Math.*f*C*R1)−arc tan(2*.Math.pi.Math.*f*C*R2), for resistor values programmed in the buffer logic for the filter output biasing resistors R1 and R2 where f is the frequency of operation in Hertz and C is a fixed capacitor in each filter in Farad, creating a filter non-linear and negative phase shift.

10. A method for programmable analog beamformer device further comprising the steps of: providing at least two independent user programmable logic controlling input radio frequency channels generating a single frequency sine signal driven by user selected phase and amplitude inputs; providing each channel with components serially connected from a buffer circuit to a phase-shifter, an amplitude control unit, a power amplifier and a load to a ground: isolating each phase-shifter with a first order filter with input and output buffer amplifiers to maintain filters at first order; carrying each load with half of the actual impendence in differential power delivery and complete impedance with a single ended power delivery, and providing virtual grounding for ground wherein the differential power is zero voltage of a beamformer circuit and a real ground of the beamformer circuit where a single ended power delivers piezo electric loads for ultrasound transducer output whereby a user selected phase and amplitudes independently in an array of radio frequency parallel channels can focus ultrasound beams via transducer output onto a target.

11. A method for programmable analog beamformer device as in claim 10 further comprising the steps of providing phase shifting using filter delays to an RF signal from 0 degrees to 180/m degrees, phase shifting in each by programmable coupling a digital potentiometer resistor with a programmed capacitor forming a single pole filter for a phase delay.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) For a better understanding of the present invention, reference is made to the accompanying drawings, which are incorporated herein by reference and in which:

(2) FIG. 1 is the architecture of n channel programmable analog beamformer in accordance with the present invention;

(3) FIG. 2 is m phase shifters in each of n channels in accordance with the present invention;

(4) FIG. 3 is a signal sign inverter in each of n channels in accordance with the present invention;

(5) FIG. 4 is amplitude control unit in each of n channels in accordance with the present invention;

(6) FIG. 5 is power stage to the loads in each of n channels in accordance with the present invention;

(7) FIG. 6 is a first order filters in one of m phase shifters, with isolating buffer amplifiers in each of n channels in accordance with the present invention.

DETAILED DESCRIPTION

(8) Simplified block diagram of programmable analog beamformer is shown in FIG. 1. Input Radio Frequency (RF) sine signal 7 at frequency of user choice enters the system (which can come from a function generator or a builtin signal generator embedded in the beamformer). The sine signal branches out to n channels 1.sub.1, 1.sub.2 . . . 1.sub.n in buffer circuit 8 (details not shown). Branching can be done via a buffer amplifier, or directing wiring to n channels as inputs of channels are high impedance and there will be no loading effect.

(9) Each channel 1.sub.1, 1.sub.2 . . . 1.sub.n has three parts: phase shifters 11.sub.1, 11.sub.2 . . . 11.sub.n; Amplitude control units 12.sub.1, 12.sub.2 . . . 12.sub.n; and final power amplifiers 13.sub.1, 13.sub.2 . . . 13.sub.n. These three parts generate same RF sine signal of buffer 8 with different phase and amplitude at outputs to loads 3.sub.1, 3.sub.2 . . . 3.sub.n. Phases and amplitudes in each channel is set through user interface software by users.

(10) Loads 3.sub.1, 3.sub.2 . . . 3.sub.n can be piezoelectric transducers in ultrasound applications with standard impedance of 50 ohm at resonance frequencies. The electrical power to the loads can be applied single ended or differentially, as shown in FIG. 5. The loads are half of the actual impedance in differential power delivery, and complete impedance in case of single ended power delivery.

(11) The grounds 4.sub.1, 4.sub.2 . . . 4.sub.n are virtual grounds, in case of differential power delivery (zero voltage of beamformer circuit), and real grounds of beamformer circuit, in case of single ended power delivery. Micro-controller 9 receives its digital instructions from a core software algorithm written in C++ that runs on computer 10, and programs each channel for different phase and amplitude; it also handles error correction mechanism.

(12) FIG. 2 and FIG. 3 together show one of the phase shifters 11.sub.1, 11.sub.2 . . . 11.sub.n in each channel 1.sub.1, 1.sub.2 . . . 1.sub.n. As seen in FIG. 2, there are m first order filters 15.sub.1, 15.sub.2 . . . 15.sub.m. Each filter 15.sub.1, 15.sub.2 . . . 15.sub.m is isolated by buffer amplifiers at input and output. For example, filter 15.sub.1 has buffer amplifier 14.sub.1 and 14.sub.2, and filter 15.sub.m has 14.sub.m+1 and 14.sub.m. The purpose of buffer amplifiers 14.sub.1, 14.sub.2 . . . 14.sub.m are to keep the filters first order, as known in the art.

(13) FIG. 6 shows one of m filters 15.sub.1, 15.sub.2 . . . 15.sub.m in dotted line 36. Buffer amplifier 31 at input and buffer amplifier 43 at output, as mentioned above, act as isolators to keep the single pole of the filter 36. The reason 36 is a single pole filter is that filter capacitor 34 on its left hand side has output of amplifier 31 with a shunt feedback which results in a very low impedance in compare to resistor 35 (technically a ground). On the right hand side of capacitor 34, there is a decoupling capacitor 37 which is a large value for the frequency of operation and short circuit in AC analysis, as known in the art. Bias resistors 39 and 40 of amplifier buffer 43 are very large values in compare to filter resistor 35, which are all three in parallel with each other in AC analysis, so resistor 39 and 40 can be ignored. Also input impedance of buffer amplifier 43 is very large value which is again in parallel with resistor 35. These will result in capacitor 34 sees a ground on its left hand side and only resistor 35 on its right hand side, and filter time constant is R*C where R is resistor 35 and C is capacitor 34. The filter 36 pole is located at frequency 1/(2*π*R*C).

(14) In this invention, R above is a programmable digital potentiometer resistor, which by changing R, the location of the pole is changed and hence the phase shifting amount can be controlled, as known in the art. Alternatively, C can be changed, for example using a varactor, or both R and C be changed. It is also possible to swap the position of R and C in filter 36, which results in a same single pole filter (which is a low pass filter in this case, and 36 is a high pass filter), and same concept can be applied, as know in the art.

(15) Phase shifting amount, AO, in each filter 15.sub.1, 15.sub.2 . . . 15.sub.m will be based on the following equation when the resistor is programmed from R.sub.1 to R.sub.2:
ΔØ=arc tan(2*f*C*R.sub.2)−arc tan(2*π*f*C*R.sub.1),  (1)
where f is the frequency of operation in Hertz and C is the fixed capacitor in each filter in Farad. As seen in equation (1) above, by increasing R the filter create positive phase shift, and by decreasing R the filter creates negative phase shift (sine signal moves toward left). Also this phase shifting is non-linear, which can be medicated by use of enough number of filters and fine enough value changes in R (the phase shift amount can be less than phase resolution requirements).

(16) Each filter 15.sub.1 to 15.sub.m can create phase shifting of 0 degrees to 180/m degrees, where all m filters in each channel create phase shifting of 0 to 180 degrees. Value m in the current invention is 4-7 which has good experimental results for 0.1-3 MHz range circuit operation with fine phase steps of less then 5 degrees. C is chosen to be 1 nF, and R is a digital potentiometer from 40 ohm to 10 k, with 256 values to choose. For 1-2 MHz operation which is mostly the case in HIFU, m of 4, digital potentiometer of 40 ohm to 1 k with 64 values to choose have achieved the goals. Different variations for different specifications can be chosen. For higher frequency, like 10 MHz, the fixed filter capacitor C can be switched to a smaller value, like 0.2 nF, using a RF switch similar to FIG. 3 in sign inverter.

(17) Additional 180 degree phase shifting in each channel is shown in FIG. 3, where the RF sine signal in the channel goes through inverting amplifier 16 and non-inverting amplifier 17 (circuitry details not shown). By using a RF switch 18 with large enough 3 dB bandwidth in compare to the frequency of operation, inverting amplifier 16 output is selected for 180 degree phase shift or non-inverting amplifier 17 output is selected for no phase shifting. This selection can also be opposite by defining the zero phase origin of the signal. This sign inverter can be placed anywhere in the channel signal path, but in this invention it is placed before final power amplifiers to provide negative polarity signal for differential power delivery; in this case another RF switch with same select signal is used in parallel with switch 18 with the switch inputs swapped (details not shown).

(18) Phase shifting in each filter introduces amplitude loss which needs to be compensated in order to keep the amplitude flat for the RF signal. Amplitude loss in each filter is calculated in the core software algorithm, by following equation:
{hacek over (A)}=(R*C*2*π*f)/√(1+(R*C*2*π*f).sup.2),  (2)
where, C is the capacitor in filter, f is the frequency of operation, and R is the resistor in the filter. The value {hacek over (A)} is the attenuation amount which is a real number between 0 and 1, with 1 means no attenuation and 0 means complete signal attenuation (complete signal loss). Equation (2) above come from high pass filter nature of the filters where they have a zero at DC, and a single pole at 1/RC. If low pass filter circuit is used instead in filters 15.sub.1 to 15.sub.m the numerator of equation (1) will be constant 1, as known in the art.

(19) Amplitude losses in all m filters of each channel are calculated by core software algorithm, based on equation (2), and are compensated accordingly by amplitude control unit in FIG. 4. Amplifier buffers 19 and 22 at input and output of the voltage divider resistors 20 and 21 isolate the divider similar to filters explained above in this disclosure. Resistor 20 is a fixed resistor and resistor 21 is a digital potentiometer programmed by core algorithm software runs on computer 10. Another amplitude control unit is current feedback amplifier 23 and a digital potentiometer 24 in its feedback network that changes the gain and hence the channel signal amplitude. The amplitude control unit is also used to program the amplitude of RF sine signal in each channel per user amplitude choice through the user interface (details not shown). There are minor phase shifts due to low pass filter nature of the voltage divider. Also, changes in the gain of amplifier 23 slightly changes the 3 dB bandwidth of the amplifier and consequently changes the phase of RF signal in the channel. These errors are captured in error correction mechanism in core software algorithm. In this invention only the first method of voltage divider is used. Amplitude control unit can be placed anywhere in the channel signal path, but in this invention it is placed after phase shifter filters to keep the signal swings low in filters due to the current rating limitation of digital potentiometer.

(20) Due to analog nature of this invention, there can be errors in phase and amplitude in each channel after programming which can come from numerous sources including but not limited to layout parasitics, electronic components variations, and so on. There is an error correction mechanism (not shown here) that observes electrical signals at each output of final power amplifiers 13.sub.1, 13.sub.2 . . . 13.sub.n at least once for any new load or frequency. Amplitude and phase errors are recorded in a deviation list. In this invention standard oscilloscope 10 communication is used to read the amplitude and phase errors, and results are sent to core software algorithm to create the deviation list. The deviation list values are added to the user desired phase and amplitudes to get correct operation of the beamformer. Also all digital potentiometers are read back after programming via micro-controller 9 to make sure programming was correct (details not shown).

(21) After phase shifting and amplitude control in each channel, the electrical power is boosted by power amplifiers 13.sub.1, 13.sub.2 . . . 13.sub.n in differential or signal ended fashion as shown in FIG. 5. Differential power delivery is when final power amplifiers 25 and 26 receive the inverted and non-inverted sine signal, respectively, from the differential path before them (outputs of sign inverter shifter) and deliver power differentially to their load; this is what is implemented in this invention. It is also possible for both final power amplifiers 25 and 26 to receive same signal and one of them invert the signal for differential power delivery purposes.

(22) Differential power delivery doubles the voltage swings to the load and hence make the electrical power 4 times more, where RF power is V.sup.2/(2*|Z|), with |Z| is the magnitude of load impedance. |Z| is a number near 50 ohm for standard piezoelectric loads in ultrasound applications, and V is the output peak voltage value at the load. In single ended fashion, amplifier 28 delivers the power to load 29 which is connected to circuit ground 30.

(23) There are whole set of digital circuitry especially in the buffer circuit 8 (details not shown) that they control the analog beamformer, such as programming digital potentiometers and controlling RF switches. The digital commands to the beamformer hardware come from the micro-controller 9 which communicates with core software algorithm runs on computer 10.

(24) This invention has been implemented on printed circuit boards (PCBs) using already designed electronic active and passive components. Same architecture can be implemented on chips for miniaturization to achieve most importantly higher channel counts, as in some medical applications up to 2000 channels are needed. High signal-to-noise ratio sine signals on PCBs at outputs of channels have been achieved with some circuit techniques, such as electromagnetic interference rejection techniques, circuit board layout techniques, and noise suppression techniques via low pass filtering.

(25) Electrical power delivered to loads 13.sub.1, 13.sub.2 . . . 13.sub.n convert to mechanical power in ultrasound applications when loads are, for example, piezoelectric transducers. The mechanical acoustic power from each channel of the multi-channel system with different phase and amplitude (energy) per user choice will penetrate the object, where in case of medicine, is a human body. The penetrated waves will interfere with each other to form a focal point of energy where the acoustic energy is focused on that part of body. In case of HIFU, the focused energy will increase the temperature in small portion of the target tissue, and consequently cancer cells can be ablated non-invasively. This beamformer has many other applications in other medical fields, not only radiation oncology, as well as in non-medical fields. The invention can be used as an electromagnetic analog beamformer also.