Method for Noncontact, Radiation Thermometric Temperature Measurement

Abstract

In a method for noncontact, radiation thermometric temperature measurement, a short-circuit photocurrent that is proportional to a received radiant power is produced in a photodiode radiation detector that is operating photovoltaically without bias voltage. The photocurrent is processed in a current to voltage converter. Subsequently, a temperature signal corresponding to the radiant power is generated. A corrective current, dependent on a temperature of the photodiode radiation detector, is added to the short-circuit photocurrent to compensate a fault current, wherein the fault current is based on an input bias current and an input offset voltage of the current to voltage converter across a temperature-dependent shunt resistance of the photodiode radiation detector. A device with a corrective current source controlled by a microcontroller is provided that can be used to perform the method.

Claims

1. A method for noncontact, radiation thermometric temperature measurement, the method comprising: producing in a photodiode radiation detector, operating photovoltaically without bias voltage, a short-circuit photocurrent that is proportional to a received radiant power; processing the short-circuit photocurrent in a current to voltage converter; generating subsequently a temperature signal corresponding to the radiant power; adding a corrective current, dependent on a temperature of the photodiode radiation detector, to the short-circuit photocurrent for compensation of a fault current, wherein the fault current is comprised of an input bias current and an input offset voltage of the current to voltage converter across a temperature-dependent shunt resistance of the photodiode radiation detector.

2. The method according to claim 1, further comprising determining the temperature of the photodiode radiation detector by a temperature sensor arranged at the photodiode radiation detector.

3. The method according to claim 1, further comprising adjusting the corrective current as a function of the temperature of the photodiode radiation detector by a current source controlled by a microcontroller and controlling the current source based on a temperature-dependent equation stored in a memory unit correlated with the microcontroller.

4. The method according to claim 3, further comprising determining the corrective current based on the following equation: $I_K=C\left(K_1+K_2\left({10}^{\frac{T-T_0}{T_{\mathrm{Diff}}}}-1\right)\right)$ wherein T means the temperature of the photodiode radiation detector, T.sub.0 refers to a reference temperature of the photodiode radiation detector, T.sub.Diff refers to a temperature difference at which the shunt resistance of the photodiode radiation detector changes by one order of magnitude, K.sub.1 and K.sub.2 are calibration constants, and C is a device-specific constant.

5. The method according to claim 4, further comprising determining the calibration constants K.sub.1 and K.sub.2 by: a) adjusting the calibration constants K.sub.1 and K.sub.2 so that no correction takes place; b) bringing the photodiode radiation detector to the reference temperature T.sub.0; c) measuring a known temperature of a black body with the photodiode radiation detector; d) subsequently, adjusting the calibration constant K.sub.1 until the generated temperature signal corresponds to the known temperature of the black body; e) bringing the photodiode radiation detector to a temperature different from the reference temperature; f) subsequently, measuring the temperature of a black body with the same known temperature as in step c) with the photodiode radiation detector; g) subsequently, adjusting K.sub.2 until the generated temperature signal corresponds to the known temperature of the black body.

6. A device for noncontact, radiation thermometric temperature measurement for performing the method according to claim 1, the device comprising: a photodiode radiation detector, operating photovoltaically without bias voltage, configured to produce a short-circuit photocurrent proportional to a radiation intensity detected by the photodiode radiation detector; a current to voltage converter configured to process the short-circuit photocurrent; an output device for outputting a temperature signal corresponding to the radiation intensity; a digitally controllable corrective current source additively connected to an input of the current to voltage converter; a microcontroller unit operatively connected to the corrective current source to control the corrective current source, wherein the microcontroller unit comprises a microcontroller and a memory unit correlated with the microcontroller.

7. The device according to claim 6, wherein the photodiode radiation detector has a shunt resistance of less than 1 MΩ.

8. The device according to claim 6, wherein the corrective current source is a digital analog converter with a downstream voltage to current converter.

9. The device according to claim 8, wherein the photodiode radiation detector comprises a photodiode with a cathode and an anode, wherein the corrective current source is arranged downstream of the digital analog converter and comprises a differential amplifier having an output voltage reference potential that relates to the cathode of the photodiode, wherein the corrective current source comprises an output connected by at least one resistor to the anode of the photodiode.

10. The device according to claim 6, further comprising a temperature sensor arranged on the photodiode, wherein the temperature sensor is configured to produce a control signal supplied to the microcontroller unit.

Description

BRIEF DESCRIPTION OF THE DRAWING

[0024] FIG. 1 illustrates the photocurrent of a photodiode radiation detector as a function of the temperature of a black body as an object to be measured.

[0025] FIG. 2 illustrates the relative shunt resistance of the photodiode radiation detector as a function of its temperature.

[0026] FIG. 3 illustrates the temperature compensation current I.sub.K as a function of the temperature of the photodiode radiation detector.

[0027] FIG. 4 is a circuit diagram of an implementation of the invention.

[0028] FIG. 5 illustrates the measuring deviation when measuring the temperature of a black body as a function of the temperature of the radiator when using the method according to the invention.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0029] Individual technical features of the embodiment described in the following can be combined also with afore described embodiments as well as with features of one of the independent claims and possibly further claims to other configurations in accordance with the invention.

[0030] Inasmuch as applicable, elements that are functionally the same are identified with same reference characters.

[0031] FIG. 1 shows the photocurrent of a photodiode radiation detector as a function of the temperature of a black body as an object to be measured. It can be clearly seen that the photocurrent changes in a temperature range of 50° to 700° across six orders of magnitude. In this context, the relative change of the photocurrent based on temperature change is more pronounced for low temperatures than for high temperatures. Below a temperature of the black body of 120° C., the current drops from a range of approximately 10 nA to a few 100 pA at 50° C.

[0032] In FIG. 2, the change of the shunt resistance of the photodiode as a function of the temperature of the photodiode radiation detector is illustrated relative to the resistance at reference temperature of 25° C. In the illustrated temperature range of −10° C. to 60° C., the shunt resistance changes by two orders of magnitude. This change follows the equation as shown below:

[00002]$R_s=R_0\left({10}^{\frac{T_0-T}{T_{\mathrm{Diff}}}}\right)$

wherein R.sub.s means the shunt resistance as a function of the photodiode radiation detector temperature, R.sub.0 means the shunt resistance at a reference temperature T.sub.0, and T.sub.Diff means the temperature difference that causes a change of the resistance by one decade. T.sub.Diff and the temperature difference T.sub.0−T are to be put into the equation with the same units, here with the employed unit K.

[0033] It has been found that a relevant temperature-dependent fault current flows across the varying shunt resistance due to the input offset voltage of an operational amplifier.

[0034] In FIG. 3, the temperature compensation current I.sub.K is illustrated as a function of the temperature of the photodiode radiation detector. This curve represents the temperature-dependent corrective current. The corrective current is determined to be 0 at a reference temperature of 25′C because here no correction of the temperature effect is performed.

[0035] FIG. 4 shows a circuit diagram of a pyrometer with photodiode radiation detector and a current to voltage converter circuit with exponentiating transfer characteristic for dynamics compression.

[0036] The preamplifier is constructed of the operational amplifiers V1, V2, and V4 and compresses the great dynamic range of the photocurrent Iph of the photodiode D1. The current which is applied by D1 flows through the transistor T1 which is switched as a diode and generates thereat a voltage drop between collector/base and emitter which corresponds to the logarithm of the photocurrent Iph. This voltage appears as U2 at the output of the chopper operational amplifier V1 which is switched as an impedance converter. Since the anode of diode D1 is connected with the non-inverted input and the cathode with the inverted input of the amplifier, only the input offset voltage of the operational amplifier V1 is applied to the photodiode and only a small fault current can thus flow across the shunt resistance of the photodiode.

[0037] At the output of the amplifier V2 which, like V1, is switched also as an impedance converter, the voltage U3 appears which is logarithmized from the reference current Iref across the transistor T2. At the outputs of V1 and V2, between the voltages U2 and U3, a voltage divider is provided which is formed of the resistors R7 and R8 and whose divided voltage U4 is applied to the base of the transistor T3.

[00003]$U_4=U_2.Math.\frac{R_8}{R_7+R_8}+U_3.Math.\frac{R_7}{R_7+R_8}$

[0038] In this way, the transistor T3 is acting for the output voltage U5 as an e-function generator with divided voltage U4 as input parameter. In this way, the circuit functions for the input current Iph as an exponentiating device whose exponent can be determined by means of a voltage divider based on the resistors R7 and R8 that are preferably designed with narrow tolerances. The exponents n and m are for Iph: n=R8/(R7+R8) and for Iref: m=R7/(R7+R8). It is not problematic that, depending on the selection of the resistors, the exponents n and m can be different because Iref as constant current can be selected freely so as to match the desired operating point adjustment. In order for the base current of T3 not to impermissibly change the output voltage of the voltage divider, the two resistors R7 and R8 are selected to be so low ohmic, preferably in a range of preferably 100Ω to 500Ω, that in this way an additional impedance converter upstream of the base connector of T3 is not required. Despite of this, the current which is flowing through this voltage divider still remains relatively low because the voltage difference U2−U3, due to the currents being logarithmized, is only in the range of maximally 100 mV-200 mV.

[0039] The circuitry principle which is illustrated in FIG. 4 has several advantages relative to alternative solutions in which the logarithmic transistor is integrated in common base configuration in the negative feedback branch of an operational amplifier.

[0040] The transistor T1 generates no voltage amplification which can impair the stability of the operational amplifier when the voltage-amplifying transistor in common base configuration is integrated in the negative feedback branch of the amplifier. Depending on the collector current, a current-dependent oscillating tendency of the circuitry is thereby generated that can be eliminated by measures that reduce the amplifier bandwidth; however, this undesirably prolongs the signal adjustment time for small photocurrents.

[0041] The circuitry expenditure for the exponentiating circuit for signal dynamics compression is very minimal with three impedance converters and three transistors.

[0042] The circuitry requires only a unipolar supply voltage wherein approximately 3 V-5 V are sufficient for operation. Accordingly, the power loss, inherent heating of the circuitry, and the complexity of the voltage supply can be significantly reduced.

[0043] The transistors T1 to T3 are all connected with their emitters to a common ground so that transistors on a monolithic transistor array can be used without thereby producing, due to different voltage potentials between the transistors, substrate leakage currents which for small measured currents then would generate disruptive fault currents. The monolithic configuration provides in addition a good synchronization of the transistor characteristic line for fluctuating environmental temperature which is desirable for a low-drift operation of the circuit.

[0044] The compensation device is comprised of a digital analog converter (D/A converter) controllable by a microcontroller; its output voltage U0 can be changed from ground potential to Uref and generates in the downstream differential amplifier V3 the output voltage U1 which does not relate to the ground potential of the D/A converter but to the cathode potential of the photodiode D1 (voltage U2). Because the required input compensation current at V1 depending on the situation of the component tolerance, can have a positive as well as a negative sign, the resistor circuit of V3 is designed such that the output voltage U1 relative to U2 can be changed bipolar. Accordingly, for the D/A converter output voltage U0=0, the voltage difference U1−U2 reaches the negative maximum value; U1−U2=0 for U0=Uref/2; and U1−U2 reaches the positive maximum value for U0=Uref. The voltage difference U1−U2 then produces across the high ohmic resistor R5C the desired corrective current with positive or negative sign.

[0045] In order to keep disturbing effects of V3 due to voltage drift and noise at a minimum for the photocurrent measurement, the resistor R5C in relation to the shunt resistance Rsh of the photodiode is high ohmic, preferably with R5C≧1,000×Rsh. Since the chopper operational amplifier used for V1 requires only small corrective currents ≦1 nA, fulfilling this condition by high ohmic resistors of e.g. ≧100 MΩ is possible. When, however, high ohmic resistors are not available in the desired quality and configuration, the output voltage of V3, of course, can also be applied through a voltage divider R5A and R5B to the resistor R5C. Under this condition, R5C should always be greater than Rsh, preferably R5C≧5 Rsh, in order not to unnecessarily reduce the effective parallel resistance at the measuring input because otherwise the signal/noise ratio of the input stage decreases.

[0046] FIG. 5 shows the measuring imprecision in the form of the deviation of the measured temperature from the known temperature of a black body for measurement with the method according to the invention and a device designed for this method. Here, the measuring deviations across a temperature range of 75° C. to 200° C. are in a range of <±0.2 K. This demonstrates the high measuring precision that can be achieved with the method according to the invention for noncontact, radiation thermometric temperature measurement.

[0047] The specification incorporates by reference the entire disclosure of German priority document 10 2016 005 321.6 having a filing date of May 2, 2016.

[0048] While specific embodiments of the invention have been shown and described in detail to illustrate the inventive principles, it will be understood that the invention may be em bodied otherwise without departing from such principles.