Wireless power transfer through embedded geometric configurations

Abstract

A wireless power transmission system includes a planar source conductor configured to generate a first periodically fluctuating electromagnetic near field in response to an alternating current received from the power source. A planar resonant source element is coplanar with the planar source conductor and has a first resonant frequency. The planar resonant source element has a Q factor that is at a maximum at the first resonant frequency. A planar resonant load element resonates at the first resonant frequency. A planar load conductor is electromagnetically coupled to and coplanar with the planar resonant load element and generates a current in response to the second periodically fluctuating electromagnetic near field from the planar resonant load element.

Claims

1. A device for transmitting power wirelessly, comprising: (a) a source unit, including: (i) an alternating current power source; (ii) a non-spiral source conductor element electrically coupled to the alternating current power source; and (iii) a resonant source element, that is coplanar with the source conductor element and that surrounds the source conductor element and that is physically decoupled from the source conductive element, the conductive resonant element having a resonant frequency and having a maximum Q factor at the resonant frequency, the resonant source element configured to resonate in response to the alternating current being applied to the source conductor element, the resonant source element comprising a first source conductive spiral, the first source conductive spiral having a helix radius and a cross sectional wire radius in which a ratio of the helix radius to the cross sectional wire radius is substantially 9.52 so as to achieve the maximum Q factor at the resonant frequency; and (b) a load unit, including: (i) a resonant load element that is spaced apart from and that is physically decoupled from the resonant source element, the resonant load element resonant at the resonant frequency and having a maximum Q factor at the resonant frequency, the resonant load element configured to resonate in response to resonance in the resonant source element, the resonant load element comprising a first load conductive spiral, the first load conductive spiral having a helix radius and a cross sectional wire radius in which a ratio of the helix radius to the cross sectional wire radius is substantially 9.52 so as to achieve the maximum Q factor at the resonant frequency; (ii) a non-spiral load conductor element, that is coplanar with the resonant load element and that is disposed within the resonant load element and that is physically decoupled from the resonant load element; and (iii) a load that is electrically coupled to the load conductor element, wherein the load conductor element is configured to apply electrical power to the load in response to resonance in the resonant load element.

2. The device of claim 1, wherein the resonant source element comprises a second source conductive spiral that is bifilar with the first source conductive spiral so that the resonant source element comprises two coplanar source conductive bifilar spirals.

3. The device of claim 2, further comprising a planar substrate and wherein the bifilar spirals include a conductive material that has been printed on the planar substrate.

4. The device of claim 1, wherein the resonant load element comprises a second load conductive spiral that is bifilar with the first load conductive spiral so that the resonant load element comprises two coplanar load conductive bifilar spirals.

5. A method of transmitting power from a source to a load, comprising: (a) generating an alternating current at the source and causing the alternating current to flow through a source conductor element that is non-spiral; (b) inductively coupling a periodic electromagnetic field resulting from the alternating current flowing through the source conductor element to a first resonant source element that is coplanar with the source conductor element and that surrounds the source conductor element, and wherein the first resonant source element comprises a first source conductive spiral, wherein the resonant source element has a resonant frequency at a frequency at which the resonant source element has a maximum Q factor, the first source conductive spiral having a helix radius and a cross sectional wire radius in which a ratio of the helix radius to the cross sectional wire radius is substantially 9.52 so as to achieve the maximum Q factor at the resonant frequency; (c) inductively coupling the resonant source element to a first resonant load element, wherein the first resonant load element comprises a first source conductive spiral wherein the resonant load element has a resonant frequency that is the same as the resonant frequency of the resonant source element, which is a frequency at which the resonant load element has a maximum Q factor, the first load conductive spiral having a helix radius and a cross sectional wire radius in which a ratio of the helix radius to the cross sectional wire radius is substantially 9.52 so as to achieve the maximum Q factor at the resonant frequency; (d) inductively coupling a load conductor element to the resonant load element that is non-spiral, wherein the load conductor element is coplanar with the first conductive spiral resonant load element, thereby inducing a current in the load conductor element; and (e) applying the current induced in the load conductor element to the load.

6. The method of claim 5, wherein the resonant source element comprises a second source conductive spiral that is bifilar with the first source conductive spiral so that the resonant source element comprises two coplanar source conductive bifilar spirals.

7. The method of claim 6, wherein the resonant load element comprises a second load conductive spiral that is bifilar with the first load conductive spiral so that the resonant load element comprises two coplanar load conductive bifilar spirals.

Description

BRIEF DESCRIPTION OF THE FIGURES OF THE DRAWINGS

(1) FIG. 1 is a schematic diagram of one embodiment of a wireless power transfer system.

(2) FIG. 2A is a schematic diagram of a model SCMR power transfer system in air.

(3) FIG. 2B is a graph demonstrating the relationship between Q.sub.max and the electrical length of the helix.

(4) FIG. 2C is a graph demonstrating the efficiency of SCMR systems with different r/r.sub.c ratios.

(5) FIG. 3 is a schematic diagram of an embodiment of a wireless power transfer system employing spiral resonant elements.

(6) FIG. 4 is a schematic diagram of an embodiment of a wireless power transfer system employing bifilar spiral resonant elements.

(7) FIG. 5 is a schematic diagram of an embodiment of a wireless power transfer system employing three-dimensional elements.

(8) FIGS. 6A-6C are schematic diagrams showing an embodiment of a wireless power transfer system employing three-dimensional elements formed by folding a flat sheet on which conductors are printed.

(9) FIGS. 7A-7B are schematic drawings of an embodiment in which each element employs three orthogonal loops.

(10) FIG. 8A is a schematic diagram of a wireless power transfer system employing multiple resonator elements.

(11) FIG. 8B is a graph relating efficiency to frequency in the embodiment shown in FIG. 7A.

(12) FIG. 9A is a schematic diagram of a wireless power transfer system employing multiple resonator elements and multiple source/load elements.

(13) FIG. 9B is a graph relating efficiency to frequency in the embodiment shown in FIG. 8A.

(14) FIGS. 10A-10C are photographs of one experimental embodiment.

(15) FIG. 11 is a photograph of a second experimental embodiment.

DETAILED DESCRIPTION OF THE INVENTION

(16) A preferred embodiment of the invention is now described in detail. Referring to the drawings, like numbers indicate like parts throughout the views. Unless otherwise specifically indicated in the disclosure that follows, the drawings are not necessarily drawn to scale. As used in the description herein and throughout the claims, the following terms take the meanings explicitly associated herein, unless the context clearly dictates otherwise: the meaning of “a,” “an,” and “the” includes plural reference, the meaning of “in” includes “in” and “on.” Also, as used herein “Q factor” means the quality factor associated with a resonant circuit.

(17) As shown in FIG. 1, one embodiment of a wireless power transmission system 100 includes a source unit 110 (transmitter unit, or TX) and a load unit 120 (receiver unit, or RX). The source unit 110 includes a planar source conductor 112 that generates a first periodically fluctuating electromagnetic near field in response to an alternating current received from the power source 114. A planar resonant source element 116 that is coplanar with the planar source conductor 112. The planar resonant source element 116 has a Q factor that is at a maximum at its resonant frequency. In one embodiment, the planar resonant source element 116 includes an inductive loop having a first end and a different second end with a capacitor 118 that couples the first end to the second end. The planar resonant source element 116 resonates with a first oscillating current at the first resonant frequency in response to excitation from the periodically fluctuating electromagnetic near field generated by the planar source conductor 112. The load unit 120 includes a planar resonant load element 126 that is spaced apart from the planar resonant source element 116 and that is preferably aligned therewith. The planar resonant load element 126 is configured to resonate at the first resonant frequency with a second oscillating current in response to excitation from the planar resonant source element 116. The planar resonant load element 126 generates a second periodically fluctuating electromagnetic near field when resonating with the second oscillating current. In one embodiment, the planar resonant load element 126 includes an inductive loop having a first end and a different second end and a capacitor 128 that couples the first end to the second end. A planar load conductor 122 is electromagnetically coupled to and coplanar with the planar resonant load element 126 and generates a current in response to the second periodically fluctuating electromagnetic near field, which is applied to a load 124.

(18) The elements are typically made from conductive wires (such as copper) or conductive ink. In one embodiment, they can be formed by depositing a conductive material (such as a metal) on a substrate (such as a crystalline substrate) and then forming the elements through an etching process, or through using conventional lithographic techniques typically employed in circuit applications.

(19) In one embodiment, the invention employs a wireless powering system based on a strongly coupled magnetic resonance (SCMR) method, which is discussed theoretically in FIGS. 2A-2C. The SCMR method is a non-radiative wireless mid-range power transfer method, which in one embodiment is effective for transferring power across a distance of between 10 cm to 300 cm. SCMR can provide wireless power transfer efficiencies that are significantly higher than the efficiencies of conventional inductive coupling methods. To achieve high efficiency, the transmitting and receiving elements (typically loops or coils) are designed so that they resonate at the desired operational frequency that coincides with the frequency of where the elements exhibit maximum Q-factor.

(20) SCMR systems use resonant transmitters and receivers that are strongly coupled. Strongly coupled systems are able to transfer energy efficiently, because resonant objects exchange energy efficiently versus non-resonant objects that only interact weakly. A standard SCMR system consists of four elements (typically four loops or two loops and two coils) as shown in FIG. 2A.

(21) The source element is connected to the power source, and it is inductively coupled to the TX element. The TX element exhibits a resonant frequency that coincides with the frequency, where its Q-factor is naturally at a maximum. Similarly, the RX exhibits a resonant frequency that coincides with the frequency where its Q-factor is naturally at a maximum. Furthermore, the load element is terminated to a load. The analysis that follows assumes that the entire system operates in air. Also, SCMR requires that the TX and RX elements are resonant at the same frequency in order to achieve efficient wireless power transfer.

(22) The analysis that follows employs TX and RX elements that have an arbitrary number of helical loops. However, in the simple embodiment shown above, only a single loop is used. The TX and RX elements can be equivalently represented by a series RLC circuit. Helices are often preferred as TX and RX SCMR elements because they exhibit both distributed inductance and capacitance and therefore, they can be designed to self-tune to a desired resonant frequency, without the need of external capacitors. Also, external capacitors have losses, which in practice can reduce the Q-factor of the TX and RX elements and in turn decrease the efficiency of SCMR systems. Based on the equivalent RLC circuit of an SCMR system, its resonant frequency, f.sub.r, can be calculated, by following equation:

(23) $\begin{array}{cc}{f}_r=\frac{1}{2\pi \sqrt{\mathrm{LC}}}& \left(1\right)\end{array}$

(24) The resonant frequency, f.sub.r is also the operational frequency for the SCMR wireless powering system. The Q-factor of a resonant RLC circuit is given by:

(25) $\begin{array}{cc}Q=\frac{{\omega }_{r}L}{R}=\frac{2\pi f_{r}L}{R}& \left(2\right)\end{array}$
Therefore, the Q-factor of a resonant helix (i.e., self resonant) can be written as:

(26) $\begin{array}{cc}Q=\frac{2\pi f_r{L}_{\mathrm{helix}}}{R_{\mathrm{ohm}}+R_{\mathrm{rad}}}& \left(3\right)\end{array}$
where L, R.sub.rad, and R.sub.ohm are the self-inductance, radiation resistance and ohmic resistance of the helix, which is for a short helix or solenoid (2r>h) are given by:

(27) $\begin{array}{cc}{L}_{\mathrm{helix}}={\mu }_{o}r{N}^2\left[\ln \left(\frac{8r}{r_c}\right)-2\right]& \left(4\right)\\ R_{\mathrm{rad}}=\left(\pi /6\right){\eta }_o{N^2\left(2\pi f_{r}r/c\right)}^4& \left(5\right)\\ R_{\mathrm{ohm}\left(\mathrm{helix}\right)}=\left(\sqrt{{\mu }_o\rho \pi f_r}\right)Nr/r_c& \left(6\right)\end{array}$
where μ is the permeability of free space, ρ is the helix's material resistivity, r is the radius of the helix, r.sub.c is the cross sectional wire radius, N is the number of turns (the simple single turn embodiment above uses N=1), f is the frequency, η.sub.o is the impedance of free space and c is the speed of light, h is the height of the helix. It should also be noted that equations (3)-(6) are valid only when r<λ/6π.

(28) SCMR requires that both RX and TX helices also exhibit maximum Q-factor at their resonant frequency f.sub.r, in order to achieve maximum wireless power efficiency. This can also be seen by the equation for describing the efficiency of an SCMR system derived in at it operation frequency f.sub.r as follows:

(29) $\begin{array}{cc}\eta \left(f_r\right)=\frac{k_{\left(\mathrm{TX}\_\mathrm{RX}\right)}^2\left(f_r\right)Q_{\mathrm{TX}}\left(f_r\right)Q_{\mathrm{RX}}\left(f_r\right)}{1+k_{\left(\mathrm{TX}\_\mathrm{RX}\right)}^2\left(f_r\right)Q_{\mathrm{TX}}\left(f_r\right)Q_{\mathrm{RX}}\left(f_r\right)}& \left(7\right)\end{array}$
where K.sub.TX.sub._.sub.RX is the mutual coupling between the RX and TX helices and where Q.sub.TX and Q.sub.RX are the Q-factors of the RX and TX helices, respectively. If the TX and RX helices are identical, then their Q-factors are equal i.e., Q.sub.TX=Q.sub.RX=Q; therefore equation (7) can be written as:

(30) $\begin{array}{cc}\eta \left(f_r\right)=\frac{k_{\left(\mathrm{TX}\_\mathrm{RX}\right)}^2\left(f\right)Q_{\mathrm{TX}}^2\left(f_r\right)}{1+k_{\left(\mathrm{TX}\_\mathrm{RX}\right)}^2\left(f_r\right)Q_{\mathrm{TX}}^2\left(f_r\right)}& \left(8\right)\end{array}$

(31) Equation (8) shows that in order to maximize the efficiency of an SMCR system, the operation frequency f.sub.r must be equal to the frequency f.sub.max, where the Q-factor is maximum. In what follows, the maximum Q-factor of a resonant helix is derived. The Q-factor of a resonant helix can be expressed in terms of its geometrical parameters using (3)-(6) as:

(32) $\begin{array}{cc}Q\left(f_r,r,r_c,N\right)=\frac{2\pi f_r{\mu }_{0}r{N}^2\left[\ln \left(\frac{8r}{r_c}\right)-2\right]}{{\left(\frac{{\mu }_0\rho \pi f_r{r}^{2}N}{r_c^2}\right)}^{\frac{1}{2}}+20{\pi }^2{N^2\left(\frac{2\pi f_{r}r}{c}\right)}^4}& \left(9\right)\end{array}$

(33) The maximum Q-factor, Q.sub.max, and the frequency, f.sub.max, where Q.sub.max occurs, can be derived from (9) using calculus as:

(34) $\begin{array}{cc}{f}_{\max }\left(r,r_c,N\right)=\frac{c^{8/7}{\mu }^{1/7}{\rho }^{1/7}}{4.Math.{15}^{2/7}{N}^{2/7}{r}_c^{2/7}{\pi }^{11/7}{r}^{6/7}}& \left(10\right)\\ Q_{\max }\left(r,r_c,N\right)=\frac{2\pi f_{\max }{\mu }_{0}r{N}^2\left[\ln \left(\frac{8r}{r_c}\right)-2\right]}{{\left(\frac{{\mu }_0\rho \pi r^2{f}_{\max }N}{r_c^2}\right)}^{\frac{1}{2}}+20{\pi }^2{N^2\left(\frac{2\pi f_{\max }r}{c}\right)}^4}& \left(11\right)\end{array}$
Based on the above discussion, an SCMR system requires that
f.sub.r=f.sub.max  (12)
which can be written based on (10) as:

(35) $\begin{array}{cc}{f}_r\left(r,r_c,N\right)=\frac{c^{8/7}{\mu }^{1/7}{\rho }^{1/7}}{4.Math.{15}^{2/7}{N}^{2/7}{r}_c^{2/7}{\pi }^{11/7}{r}^{6/7}}& \left(13\right)\end{array}$
Therefore, (13) shows that the geometrical parameters of a helix can be appropriately chosen so that the helix has maximum Q-factor at a chosen frequency, f.sub.r. For example, if the parameters f.sub.r, r.sub.c, N and ρ are specified by a designer, (13) can be solved for the radius of the maximum Q-factor, r.sub.max, as follows:

(36) 0$\begin{array}{cc}{r}_{\max }={\left[\frac{c^{8/7}{\mu }^{1/7}{\rho }^{1/7}}{4.Math.{15}^{2/7}{r}_c^{2/7}{N}^{2/7}{\pi }^{11/7}{f}_r}\right]}^{7/6}& \left(14\right)\end{array}$

(37) Next, the helices are analyzed using (10), (11) and (14) to study the behavior of the maximum Q-factor, Q.sub.max, versus the electrical length of the helix (C.sub.dev/λ.sub.Q.sub.max) at f.sub.max, which can be written as:

(38) $\begin{array}{cc}\frac{C_{\mathrm{dev}}}{{\lambda }_{\max }}=\frac{2\pi r_{\max }}{{\lambda }_{\max }}=\frac{2\pi r_{\max }{f}_{\max }}{c}& \left(15\right)\end{array}$
where L.sub.dev is the length of the helix (device), λ.sub.max is the wavelength corresponding to f.sub.max given by (10). Specifically, optimum SCMR loops with N=1 are designed in the frequency range 100 KHz≦5≦GHz for four values of the cross-sectional radius, r.sub.c=0.01, 0.1, 1.0 and 10 mm. The material of the helices is assumed copper and for each pair of f.sub.max and r.sub.c, the optimum r is calculated by (14). Then Q.sub.max from (11) is plotted in FIG. 2B versus the electrical length of the helices (C.sub.dev/λ.sub.Q.sub.max), which is calculated by (15). Specifically, FIG. 2B illustrates that for each pair of f.sub.max and r.sub.c there is an r.sub.max that provides the global maximum for the Q-factor, Q.sub.Gmax.

(39) In what follows the global maximum Q-factor of the helix, Q.sub.Gmax, is formulated. First, the local maximum Q-factor, Q.sub.Lmax, is derived by substituting (10) into (11):

(40) $\begin{array}{cc}{Q}_{\mathrm{Lmax}}=\frac{2.Math.3^{6/7}{r}_c^{6/7}{c}^{8/7}{\mu }^{8/7}{N}^{6/7}{\rho }^{1/7}\left[\ln \left(\frac{8r}{r_c}\right)-2\right]}{5^{1/7}{\pi }^{2/7}{r}^{3/7}\left[c^{4/7}{\mu }^{4/7}{\rho }^{4/7}+6r_c^{1/7}{N}^{1/7}{r}^{3/7}\sqrt{\frac{c^{8/7}{\mu }^{8/7}{\rho }^{8/7}}{r_c^{2/7}{N}^{2/7}{r}^{6/7}}}\right]}& \left(16\right)\end{array}$
Using again calculus, we can find out that the global maximum for the Q-factor occurs when:

(41) $\begin{array}{cc}\frac{r_{\left(\mathrm{Gmax}\right)}}{r_c}=\frac{e^{13/3}}{8}\approx 9.52& \left(17\right)\end{array}$
This result shows that the ratio between the helix radius, r, and the cross-sectional radius, r.sub.c, must be approximately 9.52 in order to achieve the maximum Q-factor. This ratio is also independent of frequency and material.

(42) Also, by substituting (17) into (16) we can write the global maximum for the Q-factor as:

(43) $\begin{array}{cc}{Q}_{\mathrm{Gmax}}=\frac{28.Math.2^{2/7}{r}_c^{3/7}{c}^{8/7}{\mu }^{8/7}{N}^{6/7}{\rho }^{1/7}}{{15}^{1/7}{e}^{13/7}{\pi }^{2/7}\left[c^{4/7}{\mu }^{4/7}{\rho }^{4/7}+6r_c^{4/7}{N}^{1/7}\sqrt{\frac{c^{8/7}{\mu }^{8/7}{\rho }^{8/7}}{r_c^{8/7}{N}^{2/7}}}\right]}& \left(18\right)\end{array}$

(44) Therefore, if a helix is designed to operate at the global maximum Q-factor it will yield the maximum possible wireless efficiency for the corresponding SCMR system. In order to verify the global maximum design of (17), we assume that an arbitrary ratio of r/r.sub.c=t, and solve (13) and (17) to obtain the r and r.sub.c given the number of turns, N, and the desired frequency of operation, f.sub.o:

(45) $\begin{array}{cc}{r}_{c_{\max }}=\frac{c{\mu }^{1/8}{\rho }^{1/8}}{2.Math.2^{3/4}.Math.{15}^{1/4}{N}^{1/4}{f}_o^{7/8}{\pi }^{11/8}{t}^{3/4}}& \left(19\right)\\ r_{\max }=\frac{c{\mu }^{1/8}{\rho }^{1/8}{t}^{1/4}}{2.Math.2^{3/4}.Math.{15}^{1/4}{N}^{1/4}{f}_o^{7/8}{\pi }^{11/8}}& \left(20\right)\end{array}$

(46) Based on (19) and (20), SCMR systems were designed and simulated in Ansoft HFSS for different ratios r/r.sub.c (2≦t≦50) and assuming the number of turns, N=5, distances, l.sub.1=l.sub.3=2 cm, l.sub.2=10 cm (see FIG. 2A), and operational frequency, f.sub.o=46.5 MHz. The efficiency of these designs is compared in FIG. 2C. The results clearly illustrate that the maximum efficiency is achieved for a ratio of t=9.52 that matches our derived global maximum condition of (17).

(47) The following are guidelines for designing helical TX and RX elements of SCMR wireless powering systems. An SCMR system based on helices will not be optimal unless the spacing, s, is picked so that the helices exhibit the appropriate capacitance in order to resonate at the desired operating frequency of the system. The spacing, s of an SCMR helix is an important parameter that should be picked to ensure optimal wireless power transfer efficiency. The capacitance formula for closely wound helix is as follows:

(48) $\begin{array}{cc}{C}_t=\frac{2{\pi }^{2}r{\mathrm{.Math.}}_0}{\ln \left[s/2r_c+\sqrt{{\left(s/2r_c\right)}^2-1}\right]}& \left(21\right)\end{array}$
where r is the radius of the helix, r.sub.c is the cross sectional wire radius, ε.sub.0 is the permittivity of free space, s is the spacing between adjacent turns of the helix, C.sub.t is the total distributed capacitance of the helix, and t is the thickness of the insulation coating.

(49) The capacitance formula of (21) is valid when s/2r.sub.c≦2 and t<<s−2r.sub.c. In order to resonate the helix at a desired frequency f, the spacing between two adjacent turns, s, can be adjusted to provide the required capacitance calculated from (1) as:

(50) $\begin{array}{cc}{C}_t=\frac{1}{4{\pi }^2{f}^2{L}_{\mathrm{helix}}}& \left(22\right)\end{array}$

(51) Then equation (21) can be solved for the spacing, s, as follows:

(52) $\begin{array}{cc}s=\frac{\left[e^{\left(\frac{4{\pi }^4{r}^2{\mathrm{.Math.}}_0^2}{C_t^2}\right)}+1\right]r_c}{e^{\left(\frac{2{\pi }^{2}r{\mathrm{.Math.}}_0}{C_t}\right)}}& \left(23\right)\end{array}$
Equation (23) is valid when s/2r.sub.c≦2 and t<<s−2r.sub.c. Therefore, the spacing, s, can be adjusted using (23) independently from the other geometrical parameters to achieve the necessary capacitance and without affecting the frequency where a short helix or solenoid (2r>h) exhibits maximum Q-factor since (13) shows that the f.sub.max does not depend on s.

(53) As shown in FIG. 3, the planar resonant source element 110 and the planar resonant load element 120 could each be a conductive spiral 302, which could be in the form of a conductive material that has been printed on a planar substrate. In such an embodiment, the spirals 302 have an inherent capacitance and the design of the spiral is chosen so that each spiral resonator 302 resonates at the frequency where the loop naturally exhibits its maximum Q-factor. Given the complexity of the capacitance associated with the spirals 302, their design would typically be accomplished through simulation. Similarly, as shown in FIG. 4, the planar resonant source element 110 and the planar resonant load element 120 could each include two coplanar conductive bifilar spirals 416 and 418. Because such spirals are self-resonant, they would not exhibit the same sort of capacitor loss associated with embodiments in which a capacitor is added to a conductive loop.

(54) One embodiment, as shown in FIG. 5, maintains efficiency even when the source unit 510 and the load unit 530 are not in alignment through the use of a three dimensional symmetric source unit 510 and load unit 530. In this embodiment, the source element 512 includes a first loop 514 and an electrically contiguous second loop 516 that is orthogonal to the first loop 514. Similarly, the first resonator unit 520 includes a first loop 522 and an orthogonal second loop 524. The source unit 512 is disposed inside the first resonator unit 520. The receiver unit 530 is configured similarly, having a load element 532 with a first loop 534 and a second orthogonal loop 536, and having a second resonator element 540 with a first loop 542 and an orthogonal second loop 544. More complex structures may be employed and as the spherical symmetry of the resonators increases, the effect of misalignment also decreases. A photograph of an experimental embodiment of a resonator element 1010 according to this embodiment is shown in FIG. 10A, a source element 1020 is shown in FIG. 10B and an assembled source unit 1000 is shown in FIG. 10C.

(55) One approach to making such a three-dimensional structure is shown in FIGS. 6A-6C. In this embodiment a conductive ink 612 (such as a metallic ink) is printed on a non-conductive substrate 614 (such as a plastic or a paper) to form the conductive elements of the source element 610, as shown in FIG. 6A. Similarly, as shown in FIG. 6B, a conductive ink 622 is printed on a non-conductive substrate 624 to form the conductive elements of the first resonator element 620. These shapes are then folded into cubes to form the source unit 600. A similar process can be employed to form the load unit (not shown). Also, conductive ink can be printed directly onto a three dimensional object (such as the interior of the casing of a cellular telephone, etc.) to form the load unit and the first resonator unit.

(56) As shown in FIGS. 7A-7B, the inefficiency resulting from a load unit 730 being misaligned with a source unit 710 can be reduced by increasing the spherical symmetry of each unit. One way in which this can be accomplished, as shown in FIG. 7A, is to use conductive elements 700 (i.e., source, first resonating, second resonating and load) that include a first loop 702, an electrically contiguous second loop 704 and an electrically contiguous third loop 706. In this embodiment, each loop is substantially planar and is substantially orthogonal to the other two loops. As shown in FIG. 7B, one embodiment employs a source unit 710 with a three orthogonal loop source element 712 disposed inside of a first three orthogonal loop resonator element 720, and a load unit 730 with a three orthogonal loop load element 732 disposed inside of a second three orthogonal loop resonator element 734. A photograph of one experimental embodiment of a source unit 1110 and a load unit 1120 employing elements with three orthogonal loops is shown in FIG. 11. As will be appreciated by those of skill in the art, three dimensional structures of greater complexity can increase the spherical symmetry of the elements, thereby reducing inefficiency caused by misalignment of the units.

(57) In other embodiments, multiple source and resonator elements can be employed to tune the system to more than one different frequency. Such embodiments can facilitate, for example, the transfer of both power and data from the source to the load. This ability may be useful in such situations as when it is desirable to charge a cell phone (or other type of digital device, such as a tablet) which updating some of the data stored on the device. For example, one embodiment, as shown in FIGS. 8A-8B, includes a source unit 810 with a source element 812 and two separate resonator elements: a first source resonator element 814 and a second source resonator element 816. Similarly, the load unit 820 includes a load element 822 and two resonator elements: a first load resonator element 824 that has substantially the same resonant frequency as the first source resonator element 814, and a second load resonator element 826 that has substantially the same resonant frequency as the second source resonator element 816. Use of multiple resonator elements allows the system to be tuned to multiple specific frequencies. For example, efficiency as a function of frequency is shown in FIG. 8B for an embodiment in which the distance between the units was 7 cm, the radii of the source loop 812 and the load loop 822 were 1.5 cm, the radii of the a first source resonator element 814 and the first load resonator element 824 were 2.2 cm, the radii of the second source resonator element 816 and the second load resonator element 826 and the cross-sectional radius of the wires used in each element was 2.2 mm. As can be seen, efficiency peaks at two distinct frequencies with this embodiment.

(58) In another embodiment, as shown in FIGS. 9A and 9B, the source unit 910 can include two different source elements 914 and 918 and three different source resonator elements 912, 916 and 920. Similarly, the load unit 930 includes two load elements 934 and 938 and three different load resonator elements 932, 936 and 940. As shown in FIG. 9B, an experimental embodiment using this configuration resulted in a more complex efficiency versus frequency graph. This embodiment allows for control over the bandwidth of the system, which facilitates transfer of signals (such as digital signals) during a power transfer event. This embodiment employed the following parameters: distance=10 cm; first source/load loops radius=4.7 cm; second source/load loop2 radius=8.5 cm; first TX & RX resonator loops radius=2.2 cm; second TX & RX resonator loops radius=6.5 cm; third TX & RX resonator loops radius=11.5 cm; and wire cross-sectional radius=2.2 mm. Many other combinations of source/load elements and resonator elements are possible.

(59) The above described embodiments, while including the preferred embodiment and the best mode of the invention known to the inventor at the time of filing, are given as illustrative examples only. It will be readily appreciated that many deviations may be made from the specific embodiments disclosed in this specification without departing from the spirit and scope of the invention. Accordingly, the scope of the invention is to be determined by the claims below rather than being limited to the specifically described embodiments above.