Control module with an estimator of an input electric quantity for a switching converter and method for controlling a switching converter

Abstract

Described herein is a module for controlling a switching converter, which includes at least one inductor element and one switch element and generates an output electric quantity starting from an input electric quantity. The control module generates a command signal for controlling the switching of the switch element and includes an estimator stage, which generates an estimation signal proportional to the input electric quantity, on the basis of the command signal and of an input signal indicating a time interval in which the inductor element is demagnetized. The control module generates the command signal on the basis of the estimation signal.

Claims

1. A control module for controlling a switching converter, which includes an inductor element and a switch element and is configured to generate an output electric quantity starting from an input electric quantity, said control module comprising: a switch control circuit configured to generate a command signal for controlling switching of the switch element; and an estimator stage configured to generate an estimation signal that estimates said input electric quantity; based at least in part on a length of time during which both the command signal indicates that the switch element is switched to a non-conductive state and a first control signal indicates said inductor element is magnetized; said switch control circuit being configured to generate the command signal based on the estimation signal.

2. The control module according to claim 1, comprising a reference stage, which includes the estimator stage and is configured to receive a second control signal that is a function of the output electric quantity and of an electrical reference quantity, said reference stage being configured to generate a comparison signal that depends upon the second control signal and is said input electric quantity; said switch control circuit being further configured to generate the command signal based on the comparison signal.

3. The control module according to claim 2, further comprising a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current, wherein the switch control circuit is configured to receive the comparison signal, the first control signal, and a second input signal that depends upon a current present in the switch element, said switch control circuit being configured to generate an internal signal, indicating a comparison between the second input signal and the comparison signal, and generate the command signal as a function of the internal signal and of the first control signal.

4. The control module according to claim 1, wherein the estimator stage comprises: a current generator coupled between a first reference potential terminal and an output node of the estimator stage; a first switch coupled between the current generator and the output node; a capacitor coupled between the output node and a second reference potential terminal; and a series circuit including a resistor and a second switch, said series circuit being coupled between the output node and the second reference potential terminal; wherein said first and second switches are configured to be controlled based on the first control signal and the command signal.

5. The control module according to claim 4, comprising a reference stage, which includes the estimator stage and is configured to receive a second control signal that is a function of the output electric quantity and of an electrical reference quantity, said reference stage being configured to generate a comparison signal that depends upon the second control signal and is proportional to said input electric quantity; said switch control circuit being further configured to generate the command signal based on the comparison signal, wherein the current generator is of a variable type and is configured to generate a current that depends upon the second control signal.

6. The control module according to claim 4, comprising a logic circuit configured to produce a third control signal based on the first control signal and the command signal, the third control signal indicating a time interval in which said inductor element demagnetizes; the logic circuit being configured to use the third control signal to close the first switch during the time interval in which said inductor element demagnetizes; and wherein the switch control circuit is configured to use the command signal to close the second switch during a time interval in which said switch element is in conduction, and open the second switch while the switch element is not in conduction.

7. The control module according to claim 4, comprising: a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current; and a logic circuit configured to produce a third control signal based on the first control signal and the command signal, the third control signal indicating a time interval in which said inductor element demagnetizes; the logic circuit being configured to use the third control signal to close the first switch during a time interval in which said inductor element demagnetizes, and open the first switch otherwise; and wherein the logic circuit is configured to use the command signal to close the second switch is closed during a time interval in which said inductor element is at least partially magnetized, and open the second switch during the time interval in which said inductor element is demagnetized.

8. The control module according to claim 4, comprising: a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current, the logic circuit detector being configured to use the first control signal to close the first switch during a time interval in which said inductor element is at least partially magnetized, and open the first switch during the time interval in which said inductor element is demagnetized; and wherein the switch control circuit is configured to use the command signal to close the second switch during a time interval in which said switch element is in conduction, and open the second switch otherwise.

9. A switching converter configured to generate an output electric quantity starting from an input electric quantity, the switching converter comprising: an inductor element; a switch element coupled to the inductor element; a control module that includes: a switch control circuit configured to generate a command signal for controlling switching of the switch element; and an estimator stage configured to generate an estimation signal that estimates said input electric quantity based at least in part on a length of time during which both the command signal indicates that the switch element is switched to a non-conductive state and a first control signal indicates said inductor element is magnetized; said switch control circuit being configured to generate the command signal based on the estimation signal.

10. The switching converter according to claim 9, wherein the control module includes a reference stage, which includes the estimator stage and is configured to receive a second control signal that is a function of the output electric quantity and of an electrical reference quantity, said reference stage being configured to generate a comparison signal that depends upon the second control signal and is proportional to said input electric quantity; said switch control circuit being further configured to generate the command signal based on the comparison signal.

11. The switching converter according to claim 9, wherein the control module includes a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current, wherein the switch control circuit is configured to receive the comparison signal, the first control signal, and a second input signal that depends upon a current present in the switch element, said switch control circuit being configured to generate an internal signal, indicating a comparison between the second input signal and the comparison signal, and generate the command signal as a function of the internal signal and of the first control signal.

12. The switching converter according to claim 9, wherein the estimator stage comprises: a current generator coupled between a first reference potential terminal and an output node of the estimator stage; a first switch coupled between the current generator and the output node; a capacitor coupled between the output node and a second reference potential terminal; and a series circuit including a resistor and a second switch, said series circuit being coupled between the output node and the second reference potential terminal; wherein said first and second switches are configured to be controlled based on the first control signal and the command signal.

13. The switching converter according to claim 12, wherein the control module includes a logic circuit configured to produce a third control signal based on the first control signal and the command signal, the third control signal indicating a time interval in which said inductor element demagnetizes; the logic circuit being configured to use the third control signal to close the first switch during the time interval in which said inductor element demagnetizes; and wherein the switch control circuit is configured to use the command signal to close the second switch during a time interval in which said switch element is in conduction, and open the second switch while the switch element is not in conduction.

14. The switching converter according to claim 12, wherein the control module includes: a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current; and a logic circuit configured to produce a third control signal based on the first control signal and the command signal, the third control signal indicating a time interval in which said inductor element demagnetizes; the logic circuit being configured to use the third control signal to close the first switch during a time interval in which said inductor element demagnetizes, and open the first switch otherwise; and wherein the zero crossing is configured to use the command signal to close the second switch is closed during a time interval in which said inductor element is at least partially magnetized, and open the second switch during the time interval in which said inductor element is demagnetized.

15. A lighting system comprising: a light source a switching converter coupled to the light source and configured to generate an output electric quantity starting from an input electric quantity, the switching converter including: an inductor element; a switch element coupled to the inductor element; a control module that includes: a switch control circuit configured to generate a command signal for controlling switching of the switch element; and an estimator stage configured to generate an estimation signal that estimates said input electric quantity based at least in part on a length of time during which both the command signal indicates that the switch element is switched to a non-conductive state and a first control signal indicates said inductor element is magnetized; said switch control circuit being configured to generate the command signal based on the estimation signal.

16. The lighting system according to claim 15, wherein the control module includes a reference stage, which includes the estimator stage and is configured to receive a second control signal that is a function of the output electric quantity and of an electrical reference quantity, said reference stage being configured to generate a comparison signal that depends upon the second control signal and is proportional to said input electric quantity; said switch control circuit being further configured to generate the command signal based on the comparison signal.

17. The lighting system according to claim 16, wherein the control module includes a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current, wherein the switch control circuit is configured to receive the comparison signal, the first control signal, and a second input signal that depends upon a current present in the switch element, said switch control circuit being configured to generate an internal signal, indicating a comparison between the second input signal and the comparison signal, and generate the command signal as a function of the internal signal and of the first control signal.

18. The lighting system according to claim 15, wherein the estimator stage comprises: a current generator coupled between a first reference potential terminal and an output node of the estimator stage; a first switch coupled between the current generator and the output node; a capacitor coupled between the output node and a second reference potential terminal; a series circuit including a resistor and a second switch, said series circuit being coupled between the output node and the second reference potential terminal; and a logic circuit configured to control said first and second switches based on the first control signal and of the command signal.

19. The lighting system according to claim 18, wherein the control module includes: a zero-crossing detector configured to receive a first input signal, indicative of a current through the inductor element, and produce the first control signal based on a comparison of the first input signal with a threshold indicative of zero current or close to zero current, the logic circuit being configured to use the first control signal to close the first switch during a time interval in which said inductor element is at least partially magnetized, and open the first switch during the time interval in which said inductor element is demagnetized; and wherein the switch control circuit is configured to use the command signal to close the second switch during a time interval in which said switch element is in conduction, and open the second switch otherwise.

20. A method for controlling a switching converter, which includes a inductor element and a switch element and is configured to generate an output electric quantity starting from an input electric quantity, said method comprising: generating a command signal that controls switching of the switch element; and generating an estimation signal, that estimates said input electric quantity, based at least in part on a length of time during which both the command signal indicates that the switch element is switched to a non-conductive state and a first control signal indicates said inductor element is magnetized; wherein generating the command signal comprises generating the command signal based on the estimation signal.

21. The control method according to claim 20, further comprising: generating a comparison signal, proportional to said input electric quantity, based on a second control signal, which is a function of the output electric quantity, and based on an electrical reference quantity; and generating the command signal based on the comparison signal.

22. The control method according to claim 21, further comprising: producing the first control signal based on a comparison of a first input signal, indicative of a current through the inductor element, with a threshold indicative of zero current or close to zero current; generating an internal signal indicating a comparison between the comparison signal and a second input signal that depends upon a current present in the switch element; and generating the command signal based on the internal signal and the first control signal.

23. The control method according to claim 20, wherein said generating an estimation signal comprises alternating steps of charge and steps of discharge of a capacitor, based on the first control signal and the command signal.

24. The control method according to claim 23, comprising: generating a comparison signal, proportional to said input electric quantity, based on a second control signal, which is a function of the output electric quantity, and based on an electrical reference quantity; and generating the command signal based on the comparison signal wherein said charging steps comprise injecting into the capacitor a current that varies as a function of the second control signal.

25. A method, comprising: generating an estimation signal that estimates an input electric quantity of a switching converter, which includes an inductor element and a switch element and is configured to generate an output electric quantity starting from the input electric quantity, said generating including generating said estimation signal based at least in part on a length of time during which both a command signal, which controls the switching of the switch element, indicates that the switch element is switched to a non-conductive state and a signal indicate said inductor element is demagnetized.

26. The method of claim 25, further comprising: generating a comparison signal, proportional to the input electric quantity based on a first control signal, which is a function of an output electric quantity of the switching converter and based on an electrical reference quantity; and generating the command signal based on the comparison signal.

27. The method of claim 25, wherein generating the estimation signal includes: alternating charging and discharging a capacitor based on the signal and the command signal.

28. A control module for controlling a switching converter, which includes an inductor element and a switch element and is configured to generate an output electric quantity starting from an input electric quantity, said control module comprising: a switch control circuit configured to generate a command signal for controlling switching of the switch element; and an estimator stage configured to generate an estimation signal proportional to said input electric quantity, based on the command signal and a first control signal indicating a time interval in which said inductor element is demagnetized; said switch control circuit being configured to generate the command signal based on the estimation signal, wherein the estimator stage includes: a current generator coupled between a first reference potential terminal and an output node of the estimator stage; a first switch coupled between the current generator and the output node; a capacitor coupled between the output node and a second reference potential terminal; and a series circuit including a resistor and a second switch, said series circuit being coupled between the output node and the second reference potential terminal; wherein said first and second switches are configured to be controlled based on the first control signal and the command signal.

29. The control module according to claim 28, comprising a reference stage, which includes the estimator stage and is configured to receive a second control signal that is a function of the output electric quantity and of an electrical reference quantity, said reference stage being configured to generate a comparison signal that depends upon the second control signal and is proportional to said input electric quantity; said switch control circuit being further configured to generate the command signal based on the comparison signal.

30. The control module according to claim 28, comprising a reference stage, which includes the estimator stage and is configured to receive a second control signal that is a function of the output electric quantity and of an electrical reference quantity, said reference stage being configured to generate a comparison signal that depends upon the second control signal and is proportional to said input electric quantity; said switch control circuit being further configured to generate the command signal based on the comparison signal, wherein the current generator is of a variable type and is configured to generate a current that depends upon the second control signal.

Description

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

(1) For a better understanding of the present disclosure, preferred embodiments thereof are now described, purely by way of non-limiting example and with reference to the attached drawings, wherein:

(2) FIGS. 1 and 5 show circuit diagrams of switching converters of a known type;

(3) FIGS. 2 and 3 show time plots of signals generated within the switching converter illustrated in FIG. 1;

(4) FIG. 4 shows a block diagram regarding the switching converter illustrated in FIG. 1;

(5) FIGS. 6a and 6b show time plots of signals generated within the switching converter illustrated in FIG. 5;

(6) FIGS. 7 and 10 show circuit diagrams of switching converters including the present control module;

(7) FIG. 8 shows a circuit diagram of a portion of the switching converter illustrated in FIG. 7;

(8) FIG. 9 shows time plots of signals generated within the switching converter illustrated in FIG. 7;

(9) FIGS. 11a-11d show principle circuit diagrams of converters equivalent to the converter illustrated in FIG. 10;

(10) FIGS. 12 and 13 show principle circuit diagrams of switching converters that are equivalent to one another;

(11) FIG. 14 shows a circuit diagram of a further switching converter including the present control module;

(12) FIGS. 15a and 15b show time plots of signals generated within the switching converter illustrated in FIG. 14; and

(13) FIG. 16 shows a block diagram of a lighting system.

DETAILED DESCRIPTION

(14) The present Applicant has noted how, given a switching converter, it is possible to generate a signal proportional to the input voltage V.sub.in(θ), without resorting to a resistive divider, but rather implementing an estimator circuit, which receives at input signals generated in use by the switching converter. This being said, in what follows the present control module is described with reference to a boost converter, even though it may be used also in the case of converters of a different type. In particular, the present control module is described with reference to the boost converter 60 illustrated in FIG. 7, which in turn is described with reference to the differences with respect to the boost converter 50 illustrated in FIG. 5. Components of the boost converter 60 already present in the boost converter 50 are designated by the same reference numbers, except where otherwise specified.

(15) In detail, the control module of the boost converter 60, designated by 65, includes an estimator circuit 67 and is without the first input terminal MOLT. Further, the boost converter 60 is without the resistive divider 16.

(16) In greater detail, the estimator circuit 67 comprises a current generator 68 and a first switch 70, a second switch 72, and a third switch 74, as well as a respective resistor 76 and a respective capacitor 78, referred to hereinafter as “estimation resistor 76” and the “estimation capacitor 78”, respectively.

(17) In particular, the current generator 68 is arranged between a first internal node N.sub.1 and a second internal node N.sub.2 and is configured to inject a constant current I into the second internal node N.sub.2.

(18) The first switch 70 is connected between the second internal node N.sub.2 and a third internal node N.sub.3.

(19) The estimation capacitor 78 is connected between the third internal node N.sub.3 and ground. The estimation resistor 76 is connected to the third internal node N.sub.3 and to the second switch 72, which is further connected to ground. In other words, the second switch 72 and the estimation resistor 76 form a sort of series circuit arranged in parallel to the estimation capacitor 78. In addition, the third internal node N.sub.3 is connected to the second input of the multiplier 24.

(20) The third switch 74 is connected between the second internal node N.sub.2 and ground.

(21) The first, second, and third switches 70, 72, 74 are controlled by a first command signal, a second command signal, and a third command signal, respectively. Further, the third command signal is equal to the logic negation of the first command signal. Consequently, it is possible to designate the first, second, and third command signals by A, B and Ā, respectively.

(22) In detail, when A=‘1’, the current generator 68 is electrically connected to the third internal node N.sub.3. Instead, when A=‘0’, the current generator 68 is connected to ground. Furthermore, when B=‘1’, the estimation capacitor 78 is connected in parallel to the estimation resistor 76. Instead, when B=‘0’, the estimation resistor 76 is floating.

(23) It is thus possible to designate by T.sub.A the period in which the estimation capacitor 78 is being charged, i.e., when A=‘1’ and B=‘0’. Likewise, it is possible to designate by T.sub.B the period in which the estimation capacitor 78 is discharging, i.e., when A=‘0’ and B=‘1’. Once again, it is possible to designate by T.sub.AB the period in which the estimation capacitor 78 is floating, i.e., when A=‘0’ and B=‘0’. In addition, assuming a switching period T(θ)=T.sub.A (θ)+T.sub.B (θ)+T.sub.AB(θ)<<R*C<<1/f.sub.line, where f.sub.line is the frequency of the supply line, and R and C are, respectively, the resistance of the estimation resistor 76 and the capacitance of the estimation capacitor 78, it is possible to ignore the ripple on the estimation capacitor 78, and further it may be assumed that the voltage on the estimation capacitor 78 follows the waveform of the line voltage. This being said, by applying the charge balance on the estimation capacitor 78, we obtain:

(24) $\begin{array}{cc}{\mathrm{IT}}_A\left(\theta \right)=\frac{V_e\left(\theta \right)}{R}{T}_B\left(\theta \right)& \left(1\right)\end{array}$
where R is the resistance of the estimation resistor 76. Consequently, the voltage V.sub.e(θ) on the estimation capacitor 78 itself is

(25) $\begin{array}{cc}{V}_e\left(\theta \right)=\mathrm{RI}\frac{T_A\left(\theta \right)}{T_B\left(\theta \right)}& \left(2\right)\end{array}$

(26) This being said, the calculation of the balance of the magnetic flux on the primary winding L.sub.1 yields:
V.sub.in(θ)T.sub.ON(θ)=[(V.sub.out+V.sub.F)−V.sub.in(θ)]T.sub.FW(θ)  (3)
where T.sub.FW(θ) is the period in which demagnetization of the core of the primary winding L.sub.1 occurs, whereas T.sub.ON(θ) is the period in which the transistor M is in conduction, and hence the period in which magnetization of the core of the primary winding L.sub.1 takes place.

(27) From Eq. (3) we have:

(28) $\begin{array}{cc}\frac{V_{in}\left(\theta \right)}{V_{\mathrm{out}}+V_F}=\frac{T_{\mathrm{FW}}\left(\theta \right)}{T_{\mathrm{ON}}\left(\theta \right)+T_{\mathrm{FW}}\left(\theta \right)}=\frac{T_{\mathrm{FW}}\left(\theta \right)}{T\left(\theta \right)-T_R}& \left(4\right)\end{array}$
where V.sub.out+V.sub.F is, to a first approximation, constant, and V.sub.F is the voltage drop on the output diode D.

(29) Once again with reference to Eq. (2), by imposing T.sub.A=T.sub.FW and T.sub.B=T−T.sub.R, we obtain:

(30) $\begin{array}{cc}{V}_e\left(\theta \right)=\mathrm{RI}\frac{T_{\mathrm{FW}}\left(\theta \right)}{T\left(\theta \right)-T_R}& \left(5\right)\end{array}$
i.e., the voltage V.sub.c(θ) has the same plot, but for a scale factor, as the input voltage V.sub.in(θ). In fact, from Eqs. (4) and (5) we obtain:

(31) $\begin{array}{cc}{V}_e\left(0\right)=\mathrm{RI}\frac{V_{in}\left(\theta \right)}{\left(V_{\mathrm{out}}+V_F\right)}={\mathrm{KV}}_{in}\left(0\right)& \left(6\right)\end{array}$
which demonstrates the direct proportionality present between the voltage V.sub.e(θ) on the estimation capacitor 78, and hence at input to the multiplier 24, and the input voltage V.sub.in(θ). The voltage V.sub.e(θ) and the voltage V.sub.in(θ) hence have a same phase, and consequently a same time plot.

(32) For the boost converter of FIG. 7, the control module 65 includes a logic circuit 79 that provides the control signals A, Ā, and B based on the signals sZCD and sGD such that A=sFW and B=sZCD, where sFW is a signal that is equal to ‘1’ when there occurs demagnetization of the primary winding L1, and is equal to ‘0’ during the magnetization of the primary winding L1 or when the signal sZCd is equal to ‘1’, whereas the signal sZCD is equal to the logic negation of the signal sZCD, which is equal to ‘1’ when the primary winding L1 is completely demagnetized, i.e., when the current I(t,θ) in the primary winding L1 is zero, and is equal to ‘0’ otherwise.

(33) In greater detail, the signal sZCD may be generated, for example, by the zero-current detection circuit 36. In this case, the zero-current detection circuit 36 is provided not only with the aforementioned output connected to the first logic gate 34, but also with a further output, on which it supplies the signal sZCD. In addition, the zero-current detection circuit 36 continues to provide, on the output connected to the logic gate 34, a signal such that on the set input of the flip-flop 28 the aforementioned signal sS is present.

(34) As regards the signal sFW, it is generated, as illustrated in FIG. 8, on the basis of the signal sZCD and of the signal sGD, which, as has been said, is equal to ‘1’ when the transistor M is in conduction and is equal to ‘0’ when the transistor M is inhibited. In particular, even though not illustrated in FIG. 7, the logic circuit 79 of the control module 65 comprises a second logic gate 80 of a negated OR type, which receives at input the signals sGD and sZCD and generates the signal sFW, and logic inverters configured to generate the signals sZCD and sFW, starting, respectively, from the signals sZCD and sFW. The electrical connections that involve the second logic gate 80 are not shown, as neither, on the other hand, are the logic inverters connected to the second switch 72 and the third switch 74 and designed to generate the signals sZCD and sFW. Examples of the signals sFW, sGD and sZCD are represented in FIG. 9.

(35) As illustrated in FIG. 10, and as mentioned previously, the estimator circuit 67 may be used also in the case of a flyback converter, here designated by 90. In this case, the estimator circuit 67 is again included in the control module, designated by 95. Further, we have A=sFW and B=sGD, for the reasons described in what follows. In FIG. 10, the connections between the estimator circuit 67 and the zero-current detection circuit 36 and the output Q of the flip-flop 28, as well as the second logic gate 80 and inverter for producing Ā, are not represented.

(36) In detail, the balance of the magnetic flux on the primary winding, designated by L.sub.p, yields:
V.sub.in(θ)T.sub.ON(θ)=n(V.sub.out+V.sub.F)T.sub.FW(θ)  (7)
whence we obtain:

(37) $\begin{array}{cc}{V}_{in}\left(\theta \right)=n\left(V_{\mathrm{out}}+V_F\right)\frac{T_{\mathrm{FW}}\left(\theta \right)}{T_{\mathrm{ON}}\left(\theta \right)}& \left(8\right)\end{array}$

(38) Recalling Eq. (2), from Eq. (8) it emerges how, by imposing T.sub.A=T.sub.FW and T.sub.B=T.sub.ON, and hence A=sFW and B=sGD, we obtain:

(39) $\begin{array}{cc}{V}_e\left(\theta \right)=\mathrm{RI}\frac{V_{in}\left(\theta \right)}{n\left(V_{\mathrm{out}}+V_F\right)}=K_1{V}_{\mathrm{in}}\left(\theta \right)& \left(9\right)\end{array}$

(40) Furthermore, the estimator circuit 67 may be used also in the case of converters of topologies equivalent to the flyback topology, i.e., converters having the same conversion ratio V.sub.out/V.sub.in as the one that characterizes flyback converters. In this case, the first, second, and third command signals A, B and Ā are the same as what has been described with reference to FIG. 10.

(41) Examples of topologies equivalent to the flyback topology are illustrated synthetically in FIGS. 11a-11d. In FIGS. 11a-11d, components that have already been illustrated previously are designated by the same reference numbers. Further, FIGS. 11a-11d are described briefly, with reference just to the differences with respect to what has been described with reference to FIG. 10. In addition, the primary winding is referred to as “first inductor L.sub.1”. Again, the circuit diagrams illustrated in FIGS. 11a-11d are principle circuit diagrams, and hence they are not complete, but rather are limited to showing some components and some electrical connections of the corresponding converters in order to highlight the type of the converters themselves, which substantially depends upon the arrangement of the reactive elements and of the transistor M.

(42) In particular, FIG. 11a shows a buck-boost converter 111a, where the anode of the output diode D is connected to the second terminal of the first inductor L.sub.1, whereas the output capacitor C.sub.out is connected to the first terminal of the first inductor L.sub.1 and to the cathode of the output diode. Furthermore, designated in FIG. 11a is by 100 is a gate-driving stage, which includes the control module 95. For the reason explained previously, the gate-driving stage 100 is illustrated as being without inputs, even though in actual fact it possesses the aforementioned inputs ZCD and CS, as well as the feedback terminal FB, connected in a per se known manner.

(43) FIG. 11b shows a Cuk converter 111b, which further comprises an additional capacitor C.sub.1, which is connected to the second terminal of the first inductor L.sub.1 and to the anode of the output diode D, the cathode of which is connected to the source terminal of the transistor M. In addition, the second inductor L.sub.2 is present, which is connected between the anode of the output diode D and a first terminal of the output capacitor C.sub.out, the second terminal of which is connected to the source terminal of the transistor M.

(44) FIG. 11c shows a SEPIC converter 111c, in which the positions of the output diode D and of the second inductor L.sub.2 are reversed as compared to the Cuk converter 111b. Consequently, the anode of the output diode D and a first terminal of the second inductor L.sub.2 are connected to the terminal of the additional capacitor C.sub.1 not connected to the first inductor L.sub.1. The second terminal of the second inductor L.sub.2 is connected to the source terminal of the transistor M. The output capacitor C.sub.out is arranged between the cathode of the output diode D and the source terminal of the transistor M.

(45) FIG. 11d shows a Zeta converter 111d, also known as “inverted SEPIC”, where the drain and source terminals of the transistor M are connected, respectively, to a first terminal of the input capacitor C.sub.in and to a first terminal of the first inductor L.sub.1, the second terminal of which is connected to the second terminal of the input capacitor C.sub.in. The additional capacitor C.sub.1 is arranged between the first terminal of the first inductor L.sub.1 and the cathode of the output diode D, the anode of which is connected to the second terminal of the first inductor L.sub.1. A first terminal of the second inductor L.sub.2 is connected to the cathode of the diode D. The output capacitor C.sub.out is arranged between the second terminal of the second inductor L.sub.2 and the anode of the output diode D.

(46) As illustrated in FIG. 12, the estimator circuit 67 may be used also in the case of a buck converter 120. In particular, FIG. 12 shows a principle diagram of the buck converter 120, in a way similar to the representation of FIGS. 11a-11d, i.e., without including all the components and the corresponding connections.

(47) In detail, the drain and source terminals of the transistor M are connected, respectively, to a first terminal of the input capacitor C.sub.in and to the cathode of the output diode D, the anode of which is connected to the second terminal of the input capacitor C.sub.in. A first terminal of the first inductor L.sub.1 is connected to the cathode of the output diode D, whereas a second terminal of the first inductor L.sub.1 is connected to a first terminal of the output capacitor C.sub.out, the second terminal of which is connected to the anode of the output diode.

(48) In this case, the estimator circuit 67 is still included in the gate-driving stage 100. Further, we have A=sZCD and B=sGD, for the reasons given below.

(49) In detail, the balance of the magnetic flux on the first inductor L.sub.1 yields:
[V.sub.in(θ)−V.sub.out]T.sub.ON(θ)=(V.sub.out+V.sub.F)T.sub.FW(θ)  (10)
whence, noting that V.sub.F<<V.sub.out, we obtain, to a first approximation,

(50) $\begin{array}{cc}\frac{V_{in}\left(\theta \right)}{V_{\mathrm{OUT}}}=\frac{T_{\mathrm{ON}}\left(\theta \right)+T_{\mathrm{FW}}\left(\theta \right)}{T_{\mathrm{ON}}\left(\theta \right)}=\frac{T\left(\theta \right)-T_R}{T_{\mathrm{ON}}\left(\theta \right)}& \left(11\right)\end{array}$

(51) Recalling Eq. (2), from Eq. (11) it is highlighted how, by imposing T.sub.A=T−T.sub.R and T.sub.B=T.sub.ON, and hence A=sZCD and B=sGD, we obtain:

(52) $\begin{array}{cc}{V}_e\left(\theta \right)=\mathrm{RI}\frac{V_{\mathrm{in}}\left(\theta \right)}{V_{\mathrm{out}}}=K_2{V}_{\mathrm{in}}\left(\theta \right)& \left(12\right)\end{array}$

(53) The estimator circuit 67 may be used also in the case of converters of topologies equivalent to the buck topology. In this case, the first, second, and third command signals A, B and Ā are the same as what has been described with reference to FIG. 12.

(54) An example of a topology equivalent to the buck topology is illustrated synthetically in FIG. 13.

(55) In particular, FIG. 13 shows a reverse-buck converter 130, where the cathode of the output diode D and a first terminal of the output capacitor C.sub.out are connected to a first input terminal C.sub.in. The anode of the output diode D and the second terminal of the output capacitor C.sub.out are connected, respectively, to a first terminal and a second terminal of the first inductor L.sub.1. The drain and source terminals of the transistor M are connected, respectively, to the first terminal of the first inductor L.sub.1 and to the second terminal of the input capacitor C.sub.in.

(56) FIG. 14 shows a further embodiment, described in what follows as regards the differences from the embodiment illustrated in FIG. 7.

(57) In detail, the boost converter, designated by 160 is without the multiplier 24. Furthermore, the current generator, designated by 168, of the estimator circuit, designated by 167, is of a variable type.

(58) In greater detail, the current generator 168 receives at input the control voltage V.sub.c generated by the error amplifier 58. Furthermore, in a per se known manner, the current generated by the current generator 168 is directly proportional to the control voltage V.sub.c. In other words, designating by I.sub.CH the current generated by the current generator 168, we have I.sub.CH=G.sub.M.Math.V.sub.c, with G.sub.M constant and equal to the transconductance of the current generator 168.

(59) The third internal node N.sub.3 of the estimator circuit 167 is directly connected to the negative input terminal of the comparator 26.

(60) This being said, and recalling that Eqs. (3) and (4) still apply, the charge balance on the estimation capacitor 78 yields:

(61) 0$\begin{array}{cc}{I}_{\mathrm{CH}}\left(\theta \right)T_{\mathrm{FW}}\left(\theta \right)=\frac{{\mathrm{Vcs}}_{\mathrm{REF}}\left(\theta \right)}{R}\left[T\left(\theta \right)-T_R\right]& \left(13\right)\end{array}$
where V.sub.e is set equal to Vcs.sub.REF.

(62) It follows that:

(63) $\begin{array}{cc}{\mathrm{Vcs}}_{\mathrm{REF}}\left(\theta \right)={\mathrm{RG}}_M{V}_C\frac{T_{\mathrm{FW}}\left(\theta \right)}{T\left(\theta \right)-T_R}& \left(14\right)\end{array}$

(64) Applying Eq. (4) and expressing V.sub.in(θ) as V.sub.in,pk.Math.sin (θ), where V.sub.in,pk is the input peak voltage, we finally obtain:

(65) $\begin{array}{cc}{\mathrm{Vcs}}_{\mathrm{REF}}\left(\theta \right)=V_c\frac{G_{M}R}{V_{\mathrm{out}}+V_F}{V}_{in,\mathrm{pk}}\sin \theta .& \left(15\right)\end{array}$

(66) Considering the boost converter 60 of a known type illustrated in FIG. 5, and designating by Vcs.sub.REF′ the voltage present on the output of the multiplier 24, we have
Vcs.sub.REF′(θ)=K.sub.MV.sub.cMULT(θ)=K.sub.MK.sub.PV.sub.cV.sub.in,pk sin θ  (16)
where K.sub.P=R.sub.2/(R.sub.1+R.sub.2), and K.sub.M is the gain of the multiplier 24. Consequently, considering Eqs. (15) and (16), it may be noted how Vcs.sub.REF=Vcs.sub.REF′, if K.sub.M.Math.K.sub.p=(G.sub.M.Math.R)/(V.sub.out+V.sub.F). Examples of signals generated within the boost converter 160 are illustrated in FIGS. 15a and 15b.

(67) In practice, by adopting a current generator variable in a way directly proportional to the control voltage V.sub.c, the voltage V.sub.e(θ) that is obtained on the estimation capacitor 78 may be equated to the voltage Vcs.sub.REF generated traditionally by the multiplier 24, which commonly generates a reference signal that is directly proportional to the control voltage V.sub.c and has the same profile as the voltage present on the input capacitor C.sub.in. It is hence possible to remove the multiplier 24, thus simplifying the control module and reducing the area thereof. Furthermore, even though FIG. 14 refers purely by way of example to a boost converter, the current generator 168 of a variable type may be used in converters of any type, such as, for example, flyback converters or buck converters and/or equivalent converters. In this way, it is possible to remove the multiplier also in these converters.

(68) Irrespective of the presence or otherwise of the multiplier, any one of the switching converters previously described (hence, including the estimator circuit) may be used for supplying, for example, one or more solid-state lighting devices. For instance, FIG. 16 shows a lighting system 200, which, without any loss of generality, is connected to an a.c. voltage generator 202. The lighting system 200 comprises the bridge rectifier 2 and a switching converter 204 according to any one of the embodiments previously described. Furthermore, the lighting system 200 comprises a load 206 formed, for example, by a LED or an array of LEDs.

(69) From what has been described and illustrated previously, the advantages that the present solution affords emerge clearly.

(70) In particular, the present control module enables generation of the voltage Vcs.sub.REF(θ) in such a way that it has the form of a rectified sinusoid and an amplitude that depends upon the control voltage V.sub.c, without any need to couple a resistive divider to the input capacitor C.sub.in, and hence eliminating the losses associated to the aforesaid resistive divider.

(71) Furthermore, the present control module may be applied also in the case where at input to the converter a d.c. voltage is present, instead of an a.c. voltage, as also in the case where the converter is configured to regulate an output current instead of an output voltage. In the latter case, the feedback circuit generates a signal proportional to the output current, instead of to the output voltage, in a per se known manner.

(72) In addition, in the case where the current generator of the estimator circuit is variable and directly proportional to the control voltage V.sub.c, the control module is without the traditional multiplier.

(73) In conclusion, it is clear that modifications and variations may be made to what has been described and illustrated herein, without thereby departing from the scope of the present disclosure.

(74) For instance, the third switch 74 may be connected not only to the second internal node N.sub.2, but also to the first internal node N.sub.1, instead of to ground. Furthermore, the positions within the series circuit of the estimation resistor 76 and of the second switch 72 may be reversed.

(75) Furthermore, the present control module may be included also in a switching converter controlled in the so-called “voltage mode”, or else also in a switching converter controlled in average-current mode.

(76) Finally, the present estimator circuit may be used also outside a control module of a switching converter, i.e., independently of subsequent use of the voltage V.sub.e within a control loop of a switching converter.

(77) The various embodiments described above can be combined to provide further embodiments. These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure.