Device and method for filtering the resonance peak in a power supply circuit of at least one loudspeaker

Abstract

The present invention relates to an acoustic signal supply circuit of at least one loudspeaker (HP) incorporating a filtering device of the resonance peak of said at least one loudspeaker (HP) occurring at a given frequency, characterized in that the filtering device of the resonance peak of said at least one loudspeaker (HP) is incorporated either into the first instrumentation ground circuit or in the feedback loop, this filtering device being purely electrical in the form an impedance incorporated in the first instrumentation ground circuit or in the feedback loop, the impedance parameters being predetermined as a function of the resonance peak to be filtered of said at least one loudspeaker (HP).

Claims

1. An acoustic signal supply circuit of at least one loudspeaker incorporating a filtering device of the resonance peak of said at least one loudspeaker occurring at a given frequency of the supply current of said at least one loudspeaker, said circuit comprising at least one non-inverting converter arranged upstream of said at least one loudspeaker having a positive supply terminal connected to the input power circuit and a negative supply terminal, said circuit comprising also, at the output of said at least one loudspeaker, a first instrumentation ground circuit bypassing a feedback loop connecting a point in the circuit downstream of the loudspeaker to the negative supply terminal of the non-inverting converter, the filtering device of the resonance peak of said at least one loudspeaker being purely electrical in the form of an impedance incorporated either in the first instrumentation ground circuit or in the feedback loop, the impedance parameters being predetermined as a function of the resonance peak to be filtered of said at least one loudspeaker, characterized in that: when the impedance is incorporated into the first instrumentation ground circuit, this impedance is in the form of a dead resistor coupled with a first parallel impedance called RLC comprising at least one first resistor, at least one first inductor and least one first capacitor arranged in parallel to each other, the first parallel RLC impedance being arranged in series with the dead resistor in said first instrumentation ground circuit, and when the impedance is incorporated into the feedback loop, this impedance as the second impedance comprises a second resistor coupled in parallel with at least one second inductor and at least one second capacitor, said at least one second inductor and said at least one second capacitor being arranged in series.

2. The circuit according to the preceding claim, wherein, when the impedance is incorporated into the first instrumentation ground circuit, the values of said at least one first capacitor, said at least one first resistor and said at least one first inductor are calculated according to the parameters of the at least one loudspeaker, namely the f.sub.m:M.sub.m ratio representative of mitigation and the k.sub.m:M.sub.m ratio representative of the square of the resonance angular frequency of the at least one loudspeaker according to the following equations:
f.sub.m/M.sub.m=1/R.sub.b.C.sub.b
k.sub.m/M.sub.m=1/L.sub.b.C.sub.b

3. The circuit according to the preceding claim, wherein the dead resistor value is calculated according to the following equation: $R_0=\frac{1}{C_b}.Math.\frac{1}{\sqrt{2.Math.k_m/M_m}-f_m/M_m}$

4. The circuit according to claim 1, wherein the first parallel RLC impedance comprises n first inductors and n first capacitors, n being greater than or equal to one.

5. The circuit according to the preceding claim, wherein, when the impedance is incorporated into the feedback loop, a second instrumentation ground circuit is arranged between the second impedance and the converter, the first and second instrumentation ground circuits respectively incorporating a first dead resistor for the first instrumentation ground circuit and a second dead resistor for the second instrumentation ground circuit.

6. The circuit according to the preceding claim, wherein a calibration of the device is performed based on the value of the first dead resistor, the value of the second resistor of the second impedance being determined based on the value of the first dead resistor by being 10 to 100 times greater than the value of the first dead resistor, the value of the second dead resistor being 2 to 30 times greater than the value of the first dead resistor.

7. The circuit according to claim 1, wherein, when the value of said at least one second inductor exceeds a value of 50 mH, said at least one second inductor is in the form of artificial chokes provided with gyrating means.

8. A method of current control of an acoustic signal supply circuit of at least one loudspeaker incorporating a filtering device of the resonance peak of said at least one loudspeaker according to claim 1, in which method a correction step of the resonance peak by the filtering device is carried out, said correction step being carried out downstream of said at least one loudspeaker.

9. A control method according to the preceding claim, wherein the overall resonance factor of the loudspeaker and of the filtering device is set to a Butterworth filter.

10. A control method according to the preceding claim, wherein, said at least one loudspeaker having a diaphragm, a reduction in the sound level in the highest frequencies in the direction of the perpendicular axis of the diaphragm of said at least one loudspeaker is carried out simultaneously to the filtering of the resonance peak.

11. A control method according to the preceding claim, wherein the reduction of the sound level takes place upstream of said at least one loudspeaker.

12. A control method according to claim 8, wherein temperature variations of said at least one loudspeaker are taken into account by the filtering device by variation corresponding to the parameters of the impedance of said device.

Description

[0067] Other advantages and features of the invention will appear upon reading the detailed description of implementations and embodiments, in no way limiting, and the following accompanying drawings:

[0068] FIG. 1 illustrates a schematic representation of an acoustic signal supply circuit of at least one loudspeaker, said circuit being provided with a filtering device of the resonance peak according to a first embodiment of the present invention,

[0069] FIG. 2 illustrates a schematic representation of a circuit including a non-inverting converter, this converter which may be part of an acoustic signal supply circuit of at least one loudspeaker according to the present invention,

[0070] FIG. 3 illustrates the first embodiment of the filtering device of the acoustic signal supply circuit shown in FIG. 1, this device being shown in this enlarged figure relative to FIG. 1,

[0071] FIG. 4 shows the curves of acceleration modules during a current control respectively with or without filter of the resonance peak as well as during a voltage control of a loudspeaker, the filtering being performed with a filtering device according to the first embodiment of the invention,

[0072] FIGS. 5 and 5a respectively show curves of impedance modules and degrees of angle depending on the frequencies, the filtering being performed with a filtering device according to the first embodiment of the invention,

[0073] FIG. 6 illustrates a schematic representation of a acoustic signal supply circuit of at least one loudspeaker, said circuit being provided with a filtering device of the resonance peak according to a second embodiment of the present invention,

[0074] FIG. 7 illustrates the second embodiment of the filtering device of the acoustic signal supply circuit shown in FIG. 6, the filtering device being shown in this figure expanded compared to FIG. 6,

[0075] FIG. 8 shows the curves of acceleration modules during a current control respectively with or without filter of the resonance peak as well as during a voltage control of a loudspeaker, the filtering being performed with a filtering device according to the second embodiment of the invention,

[0076] FIG. 9 shows the curves of impedance modules respectively compared of the filtering device according to the second embodiment of the invention and loudspeaker.

[0077] According to the present invention, an ideal current control solution would be to find a filtering method to filter the two effects, namely the resonance peak and the loudspeaker directivity effect without altering the current control index also known as CDI. According to the present invention, it is possible to filter only the resonance peak retaining optimally the current control index.

[0078] Applying a filter to a current control of a loudspeaker rules out any filtering structure disposed in parallel with the loudspeaker due to the finite impedance character, even of low value in terms of source according to Thevenin. This may adversely affect the index CDI so crippling a useful part of the spectrum.

[0079] As the correction of the resonance peak of the clearly comes from the intrinsic behavior of the transducer acting as loudspeaker, the present invention provides a passive solution for the downstream correction of the at least one loudspeaker.

[0080] Thus with particular reference to FIGS. 1, 3, 6 and 7, the present invention relates to a current control method for an acoustic signal supply circuit of at least one loudspeaker HP incorporating a filtering device of the resonance peak of said at least one loudspeaker, in which method a correction step of the resonance peak is performed by the filtering device, said correction step being performed downstream of said at least one loudspeaker HP.

[0081] Said at least one HP loudspeaker having a diaphragm, it is advantageously performed, simultaneously to the filtering of the resonance peak, a reduction in the level of sound in the highest frequencies in the direction of the perpendicular axis of the diaphragm of said at least one loudspeaker HP. Thus, the two main disadvantages of current control are treated simultaneously.

[0082] Advantageously, the reduction in sound level is performed upstream of said at least one loudspeaker HP. Indeed, this correction of the sound level comes from a physical acoustic phenomenon related to the loudspeaker HP environment and falls within the handling of a systematic error which may consist in an upstream correction of the loudspeaker, also known as “feedforward correction”. This has the advantage of not altering the properties of the current voltage conditioning.

[0083] Advantageously, the total resonance factor of the loudspeaker HP and the filtering device assumes the value of a Butterworth filter, which will be detailed later.

[0084] In the most general form of the present invention, it relates to an acoustic signal supply circuit of at least one loudspeaker HP incorporating a filtering device of the resonance peak of said at least one loudspeaker HP occurring at a given frequency, said circuit comprising at least one non inverting converter A.sub.0 arranged upstream of said at least one loudspeaker HP having a positive supply terminal connected to the circuit input and to a negative supply terminal.

[0085] The circuit also includes at the output of the at least one loudspeaker HP a first instrumentation ground circuit bypassing a feedback loop connecting a circuit point downstream of said at least one loudspeaker HP to the negative supply terminal of the non inverting converter A.sub.0, the resonance peak occurring at a given frequency of the supply current of the at least one loudspeaker HP.

[0086] According to an essential feature of the invention, a filtering device of the resonance peak of said at least one loudspeaker HP is incorporated either into the first instrumentation ground circuit GND.a or into the feedback loop, this filtering device being purely electrical in the form of an impedance, namely Z.sub.b shown in FIGS. 1 and 3, or Z.sub.2 shown in FIGS. 6 and 7. This impedance Z.sub.b or Z.sub.2 is incorporated into the first instrumentation ground circuit GND.a, as shown in FIGS. 1 and 3, or into the feedback loop, as shown in FIGS. 6 and 7. The parameters of the impedance Z.sub.b or Z.sub.2 are predetermined as a function of the resonance peak to be filtered of said at least one loudspeaker HP.

[0087] Thus, the filtering device is located downstream of the loudspeaker HP in both preferred embodiments that will be detailed below.

[0088] In the first preferred embodiment shown in FIGS. 1 and 3, the filtering device is incorporated in the first instrumentation ground circuit GND.a.

[0089] As shown in FIG. 2 which illustrates a non-inverting converter with a resistor R.sub.A provided between the output of the non inverting converter A.sub.0 and the feedback loop to the pole of converter A.sub.0 and a resistor R.sub.B in an instrumentation ground circuit bypassing the feedback loop, when the converter input voltage is V.sub.IN and its output voltage is V.sub.0, we can write:

[00018]$V_o=\left(1+\frac{R_A}{R_B}\right).Math.V_{\mathrm{in}}=\left(R_A+R_B\right).Math.I.Math..Math.\mathrm{as}.Math..Math.I=\frac{V_{\mathrm{in}}}{R_B}$

[0090] By comparing this equation to the equation (12)

[00019]$\begin{array}{cc}{\left[\frac{X}{E}\right]}_p={\left[\frac{X}{I}\right]}_p.Math.{\left[\frac{I}{E}\right]}_p=\frac{B_l}{M_m}.Math.\frac{1}{P_1}.Math.\frac{1}{Z_{\mathrm{HP}}}& \left(12\right)\end{array}$

[0091] and by generalizing the resistor R.sub.B to an impedance denoted Z.sub.b, in terms of transfer functions, it then appears:

[00020]$\begin{array}{cc}{\left[\frac{I}{E}\right]}_p=\frac{I}{V_{\mathrm{in}}}=\frac{1}{Z_b}& \left(13\right)\end{array}$

with, in addition:

[00021]$\begin{array}{cc}{\left[\frac{X}{I}\right]}_p=\frac{B_l}{M_m}.Math.\frac{1}{P_1}.Math.\mathrm{où}.Math..Math.P_1=\left(p^2+\frac{2}{\tau }.Math.p+{\omega }_0^2\right)& \left(6\right)\end{array}$

as was stated in the equation (6, the voltage control is subject to:

[00022]$\begin{array}{cc}{\left[\frac{X}{E}\right]}_p={\left[\frac{X}{I}\right]}_p.Math.{\left[\frac{I}{E}\right]}_p=\frac{B_l}{M_m}.Math.\frac{1}{P_1}.Math.\frac{1}{Z_{\mathrm{HP}}}& \left(12\right)\end{array}$

[0092] For a control signal denoted E, imposed at the input of the current voltage converter A.sub.0, the current control falls within a behavior such as:

[00023]$\begin{array}{cc}{\left[\frac{X}{E}\right]}_p={\left[\frac{X}{I}\right]}_p.Math.{\left[\frac{I}{E}\right]}_p=\frac{B_l}{M_m}.Math.\frac{1}{P_1}.Math.\frac{1}{Z_b}& \left(14\right)\end{array}$

[0093] A purely electrical correction can therefore be conceived by identifying at Z.sub.HP of equation (12) the constituent parameters of Z.sub.b in equation (14) so that the resonance peak is completely filtered in the same manner as the correction introduced by equation (12).

[0094] As shown in FIG. 3, the impedance Z.sub.b of the filtering device is in the form of an dead resistor R.sub.0 associated with a first parallel impedance called RLC Z.sub.bb. This first impedance comprises at least one first resistor R.sub.b, at least one first inductor L.sub.b and at least one first capacitor C.sub.b arranged in parallel to each other, the first parallel RLC impedance Z.sub.bb being arranged in series with the dead resistor R.sub.0 in said first instrumentation ground circuit.

[0095] FIG. 3 illustrates the elements leading to the development of a resonance compensation factor using a parallel RLC filter arranged in series with a dead resistor R.sub.0.

[0096] It may be suitable to build a behavior relationship identifiable to equation (7a):

[00024]$\begin{array}{cc}{Z}_{\mathrm{HP}}\approx R_e+\frac{B_l^2}{M_m}.Math.\frac{p}{P_1}=\frac{R_e.Math.M_m.Math.P_1+B_l^2.Math.p}{M_m.Math.P_1}& \left(7.Math.a\right)\end{array}$

[0097] It is indeed possible to equalize respectively the polynomials P.sub.1 and the denominator associated with the impedance Z.sub.bb in terms of the two following criteria C.sub.1 and C.sub.2:

[00025]$\begin{array}{cc}{C}_1.Math.\text{:}.Math..Math.{\omega }_0^2=\frac{k_m}{M_m}=\frac{1}{L_b.Math.C_b}.Math..Math.\mathrm{and}.Math..Math.C_2.Math.\text{:}.Math..Math.\frac{f_m}{M_m}=\frac{1}{R_b.Math.C_b}& \left(15.Math..Math.\mathrm{and}.Math..Math.15.Math.a\right)\end{array}$

[0098] F.sub.m, k.sub.m and M.sub.m having been defined in equation (5) and being predetermined parameters of the loudspeaker.

[0099] Thus, the values of said at least one first capacitor C.sub.b, of said at least one first resistor R.sub.b, and of said at least one first inductor L.sub.b are calculated according to the loudspeaker parameters, namely the f.sub.m:M.sub.m ratio representative of the mitigation and the k.sub.m:M.sub.m ratio representative of the square of the resonance angular frequency of the loudspeaker HP according to the following equations:

f.sub.m/M.sub.m=1/R.sub.b.C.sub.b   (16)

k.sub.m/M.sub.m=1/L.sub.b.C.sub.b   (16a)

[0100] Considering now the connection in series of the parallel network RLC to the resistor R.sub.0, a grouping identical to that which leads to equation (7b) allows writing the transfer function in displacement:

[00026]$\begin{array}{cc}{\left[\frac{X}{E}\right]}_p=\frac{B_l}{M_m.Math.R_0}.Math.\frac{1}{\left[p^2+\left(\frac{f_m}{M_m}+\frac{1}{R_0.Math.C_b}\right).Math.p+{\omega }_0^2\right]}=\frac{B_l}{M_m.Math.R_0}.Math.\frac{1}{W_1}& \left(17\right)\end{array}$

[0101] This association has a composite resonance factor denoted Q.sub.HP+Zb between the mechanical factor in the equation (9) denoted Q.sub.m and a purely electrical factor Q.sub.Zb such that:

[00027]$\begin{array}{cc}\frac{1}{Q_{\mathrm{HP}+\mathrm{Zb}}}=\frac{1}{Q_m}+\frac{1}{Q_{\mathrm{Zb}}}.Math.\mathrm{avec}.Math..Math.Q_{\mathrm{HP}+\mathrm{Zb}}=\frac{\sqrt{k_m/M_m}}{f_m/M_m+1/R_0.Math.C_b}& \left(18\right)\end{array}$
soit si f.sub.m>0, alors: Q.sub.Zb=R.sub.0.Math.C.sub.b.Math.√{square root over (k.sub.m/M.sub.m)}  (19)

[0102] [so if . . . then:]

[0103] After choosing the inductor values L.sub.b and of capacitor C.sub.b with regard to the criteria C.sub.1 and C.sub.2, an idealized behavior may be considered in choosing to arrange the resistor R.sub.0 such that the overall resonance factor takes an optimum value, that of a Butterworth filter corresponding to Q.sub.HP+Zb+1√{square root over (2)}.

[0104] Thus:

[00028]$\begin{array}{cc}{R}_0=\frac{1}{C_b}.Math.\frac{1}{\sqrt{2.Math.k_m/M_m}-f_m/M_m}& \left(20\right)\end{array}$

[0105] In the first preferred embodiment shown in FIGS. 1 and 3, the filtering device is therefore incorporated into the first instrumentation ground circuit GND.a. As previously mentioned, depending on the input voltage V.sub.in of the circuit, the current I flowing in the first instrumentation ground circuit GND.a through the impedance Z.sub.b is defined by I=V.sub.in/Z.sub.b. The current I.sub.f running through the converter feedback loop may be equal to 0.

[0106] A non-limiting and purely illustrative example for a loudspeaker will now be given. The selected loudspeaker is Morel EM 428 with the following parameters: L.sub.e=0.36 mH, R.sub.e=5.4 Ω, BI=5.4 T.m, M.sub.m=6.55 g, k.sub.m=1136 N/m, f.sub.m=0.86 kg/s, Resonance F.sub.0=66.29 Hz

[0107] The resonance, mechanical, electrical and combined factors, measured and theoretical, are respectively, the theoretical factor being that between parenthesis, Qm 3.03 (3.17), Qe 0.48 (0.505), Qm+e 0.41 (0.436).

[0108] As defined above, the application of the criterion C.sub.1 on the resonance frequency is to be considered depending on the availability and price of the respective two components: the one or more inductor(s), advantageously chokes wirewound on air, and the one or more bipolar capacitor(s).

[0109] To first order, the choice of a choke of L.sub.b=12 mH requires a capacity of C.sub.b=480 μF to cover the resonant frequency of 66.3 Hz. However, such a choice may be discussed, in terms of price and components-specific defects, in the light of possible combinations, including with, possibly, a choke n times weaker associated with n capacitors arranged in parallel.

[0110] The application of the criterion C.sub.2 on the equivalence of relaxation times leads to a resistor value of R.sub.b=15.8 Ω. Finally, equation (20) leads to the selection of a series resistor of R.sub.0=4.55 Ω. Without prejudice to their ability to dissipate heat resulting from an operational system with the transducer, such components are readily available commercially.

[0111] To simplify notation, the reasoning focused on the displacement function, but it is the acceleration that should be the focus as to the acoustic result.

[0112] It is important to maintain the system stability, whatever the operating regime of the combination converter and said at least one loudspeaker. The impedance relationship representative of the phase shift between the signals does not present a phase difference close to 180°, which could cause the oscillation of the amplifier.

[0113] FIG. 4 illustrates the curves of the acceleration modules during a current control with or without filter of the resonance peak as well as during a voltage control of a loudspeaker, filtering being performed with a filtering device according to the first embodiment of the invention. The intermediate curve is the curve with current control and filtering with a filtering device according to the first embodiment and shows the absence of a resonance peak unlike the upper curve with current control without filtering. In addition, the intermediate curve has a substantially constant acceleration module range, wider than that of the lower curve which is the curve with voltage control.

[0114] In FIGS. 5, 5a for the loudspeaker taken as an example, the curves of impedance modules Z.sub.HP, Z.sub.bb and Z.sub.b, and their degrees of angle depending on the frequency, were traced respectively in a comparative manner. The curve for module Z.sub.HP is the one with rectangles, that for Z.sub.bb with lozenges, and that for Zb with circles. The curves shown in FIG. 5a thus show that the phase shift angle remains within a range of perfectly permissible values, preferably from −40° to +40°, on the frequency domain considered.

[0115] The second embodiment of the filtering device provides for its incorporation into the feedback loop of inverter A.sub.0. This is shown in FIGS. 6 and 7.

[0116] The filtering device is incorporated into the feedback loop and is in the form of a second impedance Z.sub.2 comprising a second resistor R.sub.2 coupled in parallel with at least one second inductor L.sub.2 and at least one second capacitor C.sub.2, said at least one second inductor L.sub.2 and said at least one second capacitor C.sub.2 are arranged in series.

[0117] Compared to the first embodiment which required the passage of a strong current in the filter R.sub.0, R.sub.b, L.sub.b, C.sub.b as shown in FIG. 1, the second embodiment avoids this disadvantage by choosing to have the filtering device in the feedback loop of the converter A.sub.0.

[0118] In this second embodiment, shown in FIGS. 6 and 7, wherein a second instrumentation ground circuit is arranged between the second impedance Z.sub.2 and the converter A.sub.0, the first and second instrumentation ground circuits respectively incorporate a first dead resistor R.sub.B for the first instrumentation ground circuit GND.a and a second dead resistor R.sub.3 to the second instrumentation ground circuit.

[0119] V.sub.in being the input voltage of the circuit, V.sub.0 the output voltage of converter A.sub.0 and V.sub.f the voltage in the bypass loop, we can define a factor β such that:

V.sub.f=βV.sub.0 et V.sub.0=1/βV.sub.in

[0120] The current I.sub.f returning to the negative pole of the converter after bypassing the second instrumentation ground circuit may be equal to 0.

[0121] Another conventional calculation shows that the transconductance is then defined by:

[00029]$\begin{array}{cc}\frac{I}{V_{\mathrm{in}}}=\frac{1}{R_B}.Math.\left(1+\left(Z_2+R_B\right)/R_3\right)& \left(21\right)\end{array}$

[0122] Referring to all equations mentioned above, it is possible to calculate the second impedance Z.sub.2, the mechanical resonance factor of the impedance Z.sub.2 and the initial angular speed ω.sub.0:

[00030]$Z_2-\frac{p^2.Math.R_2+R_2/L_2.Math.C_2}{p^2+\frac{R_2}{L_2}.Math.p+1/L_2.Math.C_2}$${\omega }_0=\frac{1}{\sqrt{L_2.Math.C_2}}$$Q_{Z.Math..Math.2}=\frac{1}{R_2}.Math.\sqrt{\frac{L_2}{C_2}}$

[0123] A calibration may be performed first with the choice of R.sub.B, then R.sub.2 much higher than R.sub.B, since R.sub.2 determines the generic behavior outside the resonance. The quality coefficient of this filter structure, however, requires an inductor value much higher than that defined for the above solution.

[0124] For example, without being exhaustive, the calibration of the device can be achieved based on the value of the first dead resistor R.sub.B, the value of the second resistor R.sub.2 of the second impedance Z.sub.2 being determined based on the value of the first dead resistor R.sub.B by being 10 to 100 times greater than the value of the first dead resistor R.sub.B, the value of the second dead resistor R.sub.3 being 2 to 30 times greater than the value of the first dead resistor R.sub.B.

[0125] Furthermore, depending on the structure of the converter A.sub.0, the R.sub.B value must not advantageously be too high to maintain stability. For the loudspeaker used as an example, with a unit value of R.sub.B, the behaviors of the filtering device and the transconductance are illustrated in FIG. 8. At the resonance frequency, the phase paths emphasize the phase rotation presented by Z.sub.2 and the symmetry observed by transconductance compared with the transducer.

[0126] The component values shown in the figure are adjusted to respond to an electric quality factor of the order of 0.45 for a correct association with the resonance of the transducer.

[0127] For a control voltage denoted E soliciting the entry of the conditioner and considering the last elements presented assigned their respective values, the acceleration behavior resulting from the arrangement is expressed:

[00031]$\begin{array}{cc}{\left[\frac{A}{E}\right]}_p=p^2.Math.{\left[\frac{X}{E}\right]}_p=p^2.Math.{\left[\frac{X}{I}\right]}_p.Math.{\left[\frac{I}{E}\right]}_p=p^2.Math.\frac{B_l}{M_m}.Math.\frac{1}{P_1}.Math.\frac{1}{R_B}.Math.\left(1+\left(Z_2+R_B\right)/R_3\right)& \left(22\right)\end{array}$

[0128] In this notation, E has a unit value. A gain or attenuation can allow an adjustment, either by adding a stage upstream of the conditioner, or by changing the value of the resistor R.sub.B subject not to alter the stability depending on the amplifier used.

[0129] In the figures, it should be noted that the loudspeaker taken as an example has a moderate mechanical resonance factor value of Q.sub.m≈3, value low enough to allow the proposed adjustment. The calculation shows that for a Q.sub.m value greater than 8, for example a loudspeaker Pioneer® TS-123, the mode of application of this correction are more demanding. In fact, for the chosen example, this solution requires an inductor of 80 mH, value can be crippling for a winding on air. An artificial choke assembly, in particular with gyrating devices, may nevertheless be envisaged, an inductor variation being thus easily obtained.

[0130] Another non-limiting example can also be given for a loudspeaker having the following characteristics: B.sub.l=2.675 Tm, M.sub.m=3.67 g, f.sub.m=0.539 N/m, k.sub.m=5650 N/m, resonance frequency=197 Hz, R.sub.e=3.655Ω, L=0.12 mH.

[0131] In this case, according to the first embodiment of the filtering device, the relationship concerning criteria C.sub.1 and C.sub.2 leads for example to L.sub.b=2.2 mH, C.sub.b=295 μF, R.sub.b=23Ω. Similarly, the Butterworth equivalence results in: R.sub.0=2.2Ω.

[0132] These values are not binding and even allow considering standard components. Finally, the overall result is idealized if the nominal current transconductance voltage conditioner is set at 2 A/V.

[0133] Advantageously, the chokes serving as inductors in the various embodiments described can be made of metals or light alloys other than copper, the main criterion being a good electrical conductivity.

[0134] FIG. 8 illustrates the current control curves with or without filtering of the resonance peak, as well as the voltage control curve of a loudspeaker, filtering being performed with a filtering device according to the second embodiment of the invention. The intermediate curve is the curve with current control and filtering with a filtering device according to the second embodiment and shows the absence of a resonance peak unlike the upper curve with current control without filtering. In addition, the intermediate curve has a substantially constant acceleration module range wider than that of the lower curve which is the curve with voltage control.

[0135] In FIG. 9, the respectively compared impedance module curves of the filtering device Z.sub.2 and said at least one loudspeaker Z.sub.HP were traced for the loudspeaker used as an example. These impedance modules curves increase and decrease in opposition. The curve with rectangles illustrates the impedance curve Z.sub.2 of the filtering device while the curve with the circles illustrates the impedance curve Z.sub.HP of the loudspeaker.

[0136] In what has been described above, at least one non-inverting converter was used in the circuit to simplify the calculations. This is not limitative and the present invention can however also be applied to a circuit comprising several non inverting converters as well as one or several inverting converters.

[0137] The market for audio reproduction, especially high-end reproduction, is directly concerned by filtering devices according to the present invention. The big brands, such as Bose®, Bang & Olufsen®, Harman Kardon®, B&W®, etc . . . should certainly be interested in the commercial distribution of such filtering devices.