Antenna for implant and associated apparatus and methods

Abstract

An antenna for a medical implant device is described. The antenna has a magnetic field radiator portion and an electric field radiator portion coupled to the magnetic field radiator portion. The magnetic and electric field radiators are arranged to result in generation, by the antenna, of at least one of a transverse electric leaky wave and a transverse magnetic leaky wave in lossy body tissue of a human or animal body such that the lossy body tissue acts as a waveguide for the transverse electric leaky wave or transverse magnetic leaky wave, whereby to optimize at least one of the efficiency of the antenna and the far field gain of the antenna.

Claims

1. An antenna for transmitting signals from a transmitter to a corresponding receiver even when there is signal attenuating human body tissue of a user positioned in a signal path between the transmitter and the corresponding receiver, the antenna comprising: a magnetic field radiator portion and an electric field radiator portion; wherein the magnetic field radiator portion and electric field radiator portion are arranged relative to one another such that, when said signal attenuating human body tissue of the user is located between the transmitter and the corresponding receiver, the antenna generates at least one of a transverse electric leaky wave or a transverse magnetic leaky wave in body tissue of the user such that the body tissue, in which the at least one transverse electric leaky wave or a transverse magnetic leaky wave is generated, acts as a waveguide for said at least one of a transverse electric leaky wave or a transverse magnetic leaky wave, and such that at least one of the radiation efficiency of the antenna or the far field gain of the antenna is maximized.

2. An antenna according to claim 1, wherein the antenna is configured to operate with a transmitter of a device, when the device is located in a pocket of body tissue of said user, in which the signal attenuating human body tissue is located in a first direction relative to the transmitter, and the body tissue in which the at least one transverse electric leaky wave or a transverse magnetic leaky wave is generated extends in a second direction, relative to the transmitter, the second direction being different than the first direction.

3. An antenna according to claim 1, wherein the magnetic field radiator portion and electric field radiator portion are arranged for the generation of at least one surface wave, the at least one surface wave propagating on body tissue comprising a skin layer, whereby the skin layer acts as a waveguide for propagating signals from said transmitter to said receiver.

4. An antenna according to claim 3, wherein the magnetic field radiator portion and electric field radiator portion are arranged for the generation of at least one surface wave mode, the at least one surface wave minimizing propagation loss between the transmitter and the corresponding receiver.

5. The antenna as recited in claim 1, wherein the magnetic field radiator portion comprises a loop, formed by a first conducting member having a first end and a second end, and having a feed pin at the first end and a ground point at the second end, and wherein the magnetic field radiator portion is configured to generate a magnetic field in the frequency of operation as current flows from the feed pin to the ground point.

6. The antenna as recited in claim 1, wherein the antenna is configured to generate a transverse electric leaky wave in a frequency of operation in said human body tissue.

7. The antenna as recited in claim 1, wherein the magnetic field radiator portion comprises a loop and wherein said electric field radiator is connected to said magnetic field radiator at a location, the location being predetermined to be a location resulting in said generation of at least one of a transverse electric leaky wave or a transverse magnetic leaky wave.

8. The antenna as recited in claim 1, wherein the magnetic field radiator portion comprises a loop and wherein said electric field radiator is connected to said magnetic field radiator at a specific location on a perimeter of the loop to minimize current reflected back into the loop and disrupting magnetic current flowing through the loop, to reduce a phase difference between electric and magnetic fields generated in the body tissue.

9. The antenna as recited in claim 1, wherein the magnetic field radiator portion comprises a loop and wherein the magnetic loop comprises a capacitive gap configured to reduce a resonant frequency of the loop.

10. The compound field antenna as recited in claim 9, wherein the antenna is configured to operate with a radio chipset, wherein a conducting member extends proximate the capacitive gap, and wherein the conducting member has a length configured such that an input impedance of said loop matches an impedance of the radio chipset.

11. The antenna as recited in claim 1 configured such that, in operation when there is signal attenuating human body tissue of a user positioned in the signal path between the transmitter and the corresponding receiver, the body tissue in which the at least one transverse electric leaky wave or transverse magnetic leaky wave is generated enhances the far field radiation efficiency of the antenna.

12. The antenna as recited in claim 1 configured such that, in operation when there is signal attenuating human body tissue of a user positioned in the signal path between the transmitter and the corresponding receiver, the body tissue in which the at least one transverse electric leaky wave or transverse magnetic leaky wave is generated enlarges an effective aperture of the antenna.

13. A device comprising a transmitter provided with an antenna as claimed in claim 1.

14. A device comprising a transmitter for transmitting signals to a corresponding receiver even when there is signal attenuating human body tissue positioned in a signal path between the transmitter and the corresponding receiver, wherein the transmitter comprising an antenna having a magnetic field radiator portion and an electric field radiator portion, wherein the magnetic field radiator portion and electric field radiator portion are arranged relative to one another such that, when said signal attenuating human body tissue of the user is located between the transmitter and the corresponding receiver, the antenna generates at least one of a transverse electric leaky wave or a transverse magnetic leaky wave in body tissue of the user such that the body tissue, in which the at least one transverse electric leaky wave or a transverse magnetic leaky wave is generated, acts as a waveguide for said at least one of a transverse electric leaky wave or a transverse magnetic leaky wave, and such that at least one of the radiation efficiency of the antenna or the far field gain of the antenna is maximized.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) Embodiments of the invention will now be described by way of example only with reference to the attached figures in which:

(2) FIG. 1 illustrates, in simplified overview, a medical implant system;

(3) FIG. 2 illustrates, in simplified form, key electromagnetic wave propagation mechanisms in a human body;

(4) FIG. 3 is a plot of the typical dielectric constant and the conductivity for skin, fat, and muscle;

(5) FIG. 4 illustrates constant planes of amplitude and phase of a propagating wave in body tissue and air;

(6) FIG. 5 is a dispersion diagram for real and imaginary parts of a complex wave number in a cylindrical body model for fat and skin tissue;

(7) FIG. 6 shows Transverse Electric (TE) and Transverse Magnetic (TM) dispersion diagrams for different thicknesses of fat;

(8) FIGS. 7(a) and 7(b) illustrate an implant device suitable for the medical implant system of FIG. 1;

(9) FIGS. 8(a) and 8(b) illustrate another implant device suitable for the medical implant system of FIG. 1;

(10) FIG. 9 is a flow diagram showing a procedure that may be used to design an antenna such as the antennas of the implants shown in FIGS. 7 and 8;

(11) FIGS. 10(a) and 10(b) show measured antenna gain as a function of angle in the elevation plane for a compound antenna;

(12) FIG. 11 shows the real and imaginary parts of an input impedance of an antenna as a function of frequency; and

(13) FIG. 12 shows the simulated and measured return loss of an antenna as a function of frequency.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT(S)

(14) FIG. 1 shows, in simplified overview, a medical implant system generally at 100. The implant system comprises an implant device 102 comprising a transmitter for transmitting radio frequency signals, and an associated fixed or mobile (e.g. tablet) based programmer 104-1, 104-2 comprising a receiver for receiving signals transmitted from the implant device to allow the programmer 104 to monitor the implant device 102 when the implant device 102 is implanted in a patient and operational.

(15) The implant device 102 comprises an implant can which encloses the analog circuitry, radio circuitry and battery of the device. The radio circuitry of the implant device 102 is configured to transmit signals in one of the industrial, scientific and medical (ISM) portions of the radio spectrum or any other suitable part of the spectrum. In the example shown in FIG. 1 the transmitter is configured to transmit signals in a 2.4 GHz to 2.5 GHz ISM frequency band centered on 2.45 GHz (suitable for Bluetooth communication). It will be appreciated, however, that in other examples the transmitter may be configured for transmission in one or more other frequency bands depending, for example, on local preferences or regulatory requirements (e.g. in the Medical Implant Communication Service (MICS) band between 402 and 405 MHz). To facilitate transmission in the appropriate radio band, the transmitter of the implant device 102 is provided with an appropriately configured antenna 106 via which to transmit signals for receipt by the receiver of the programmer 104.

(16) In FIG. 1, the implant device is shown implanted inside the body 112 of a patient (shown in simplified transverse cross-section) lying prone on an operating table 122 or the like. In this example, the implant is located in the buttocks area (typically just below the belt line) of a patient with a large waist (e.g. ≣60 inch). The patient's body 112 comprises many different types of tissue including the vertebrae 114 (i.e. the backbone region) surrounded by predominantly muscle region 116, which in turn is surrounded by a fat layer 118 and an external skin layer 120.

(17) In this example, the programmer 104 is located at some distance (typically 2 m to 3 m) from the patient, effectively in the far field of (ten wavelengths or greater from) the transmitter of the implant device 102. By way of example, at 2.45 GHz, the free space wavelength is approximately 12.5 cm (compared to a guided wavelength in the region of 1.7 cm in muscle). Accordingly, when the implant is a pacemaker, or the like, implanted approximately 8 cm beneath the skin, the pacemaker will be around 25 wavelengths away from the receiver in the programmer 104 (which is firmly in the far field).

(18) As explained in more detail later, there are significant differences between the dielectric constant and conductivity (or loss tangent) of fat (e.g. in terms of water content) in comparison to skin and muscle for the frequency bands of interest (e.g. both the 2.4 GHz to 2.5 GHz Bluetooth band or the 402 to 405 MHz MICS band). This leads to two key propagation mechanisms as shown in FIG. 2 which illustrates, in simplified form, key electromagnetic wave propagation mechanisms in a human body for radio frequency signals from an implant device such as that illustrated in FIG. 1.

(19) As seen in FIG. 2, in a first propagation mechanism 212 comprises guided and leaky wave propagation in the fat layer 118 and a second propagation mechanism 214 comprises surface wave propagation on the skin layer. In more detail, electromagnetic waves propagating from the fat layer to the skin layer or from the fat layer to the muscle layer propagate from a rarer (lower dielectric constant) to a denser (higher dielectric constant) medium resulting in a portion of the energy being reflected back into (and then around) the fat layer and a portion of energy transmitted into the skin or muscle layer. In known implant technology, this can significantly inhibit (or even prevent) electromagnetic waves from leaving the body as shown at 216.

(20) Beneficially, the antenna 106 of this example is an implantable compound antenna has a combination of a loop and a monopole configured such that the gain of the antenna is increased by approximately 4 dB compared to a standard loop antenna. The implantable antenna 106 is designed using a particularly advantageous method in which the antenna design is specifically tailored to enhance the far field gain of the transmitter, using a model of the patient's body, to take advantage of the electromagnetic nature of the human body and, in particular, to make use of, rather than suppress, the effects of the propagation mechanisms exhibited in the body. Specifically, the implantable compound antenna 106 is designed using a new method that has the potential to allow the successful use of compound antennas, that are immersed in dispersive lossy media such as human body tissue, for effective communication with external transceivers that are located at a significantly greater distance, compared to that of known implant communication technologies, from the transmitter (in the far-field). In operation, the antenna 106 excites electric and magnetic fields in lossy dielectric media (like muscle and fat) such that they achieve enhanced radiation efficiency compared to standard dipole antennas or loop antennas.

(21) In effect, the method results in an antenna design that makes use of the body tissue, as a lossy dielectric waveguide, to excite leaky wave modes in the fat and skin layer, such that the energy radiated by the implant antenna is not trapped inside the body but instead couples efficiently to the air medium surrounding the body, so as to achieve enhanced radiation efficiency.

(22) This contrasts with more conventional implant antenna design in which the feed point impedance of the antenna is merely tuned to match the antenna's impedance to the impedance of the transceiver by either measuring power locally at the implant or by using closed loop control with the external device. Whilst this may involve measuring the loading of the body tissue on the implant antenna it does not use the body tissue to enhance radiation efficiency.

(23) The antenna 106 of FIG. 1, the method of designing it, and the other examples described herein, have the potential to provide one or more of the following advantages: a smaller antenna size; a maximized communication distance between the implant and the external device for implant depths from 2 cm to 12 cm; reduced sensitivity to changes in body morphology after implantation (patient adding or losing weight), lowered specific absorption rate (permits the implant radio to transmit at higher radio power without exceeding the regulatory limits for tissue heating) and an improved battery life of the implant or smaller batteries for fixed implant life span.

(24) It will be appreciated that, whilst the techniques described herein are particularly beneficial for the design of compound antennas, the methodology can be beneficially extended such that any antenna type can be designed to excite leaky wave modes in the different body tissue thereby effectively increasing the aperture of the antenna to enhance the antenna's radiation efficiency.

(25) Radio Propagation in the Human Body

(26) Whilst those skilled in the art will be familiar with radio propagation mechanisms in the human body a brief summary of the science of such mechanisms, and how it is beneficially applied in the technology disclosed herein, will now be provided, by way of example only, to aid understanding.

(27) Radio wave propagation in the human body is quite involved as the body is a dispersive lossy dielectric medium with a complex dielectric constant ϵ.sub.r given by the so called 4 pole Cole-Cole model:

(28) $\begin{array}{cc}{ϵ}_r\left(\omega \right)={ϵ}_{\infty }+\underset{n=1}{\overset{4}{.Math.}}\frac{{\mathrm{\Delta ϵ}}_n}{1+\left(j{\mathrm{\omega \tau }}_n\right)\left(1-{\alpha }_n\right)}+\frac{{\sigma }_i}{j{\mathrm{\omega ϵ}}_0}& \left(1\right)\end{array}$
where ω is the angular frequency for the radio wave, ϵ.sub.∞ is the permittivity at high frequencies (in the THz frequency range with ωτ.sub.n>>1), ϵ.sub.0 is the permittivity of free space, σ.sub.i is the ionic conductivity for each dispersion region, τ.sub.n is the relaxation time, α.sub.n is a distribution parameter which causes a broadening of the dispersion (0 for water, >0 for most tissues, negligible for body fluids), and Δϵ.sub.n the difference between the low frequency permittivity ϵ.sub.s (ωτ.sub.n<<1) and the high frequency permittivity.

(29) The values of these variables for different types of tissue are given in [17].

(30) FIG. 3 shows a plot of the dielectric constant (in Farads/meter) and the conductivity (in Siemens/meter) for skin, fat, and muscle. FIG. 3 illustrates the significant difference in the dielectric constant and conductivity (or loss tangent) of fat in comparison to skin and muscle, both in the 402 and 405 MHz MICS band and the 2.4 to 2.5 GHz Bluetooth band. Specifically, both skin and muscle can be seen to exhibit similar dielectric constants, for these frequency bands, generally in the 50 F/m to 70 F/m range. Contrastingly fat can be seen to have a much lower dielectric constant in the 5 to 10 F/m region. This leads to the two key propagation mechanisms described above with reference to FIG. 2, namely: (1) guided and leaky wave propagation in the fat layer, and (2) surface wave propagation on the skin layer.

(31) As explained above, therefore, electromagnetic waves propagating from the fat layer to the skin layer or from the fat layer to the muscle layer always propagate from a rarer (lower dielectric constant) to a denser (higher dielectric constant) medium resulting in a portion of the energy being reflected into the fat layer and a portion of energy transmitted into the skin or muscle layer. The ratio of the reflected and the transmitted portions depends on the electrical properties of the tissue, the polarization of the electromagnetic wave, and the angle of incidence of the wave at either interface. The energy transmitted into the muscle layer is lost as heat but the energy transmitted in the skin layer is important, in the examples disclosed herein, because it maximizes the coupling of energy out of the body.

(32) The wave number, k, in units of m.sup.−1, of the electromagnetic wave is complex and is given as:
{right arrow over (k)}={right arrow over (β)}−j{right arrow over (α)}  (2)
where {right arrow over (β)} is the propagation constant (in radians/meter) that provides information about the direction of propagation of the amplitude of the electromagnetic wave, and {right arrow over (α)} is the attenuation constant (in Neper/meter) that provides information about the direction of propagation of the amplitude wavefronts of the electromagnetic wave. Based on the amplitude and phase of these quantities there can exist different types of complex waves in different body tissue.

(33) Referring to FIG. 4, this illustrates the constant planes of amplitude and phase of the propagating wave in a dispersive medium. The angle ϕ=arctan(β.sub.0x/β.sub.z) is the transmitted wave angle and ψ=arctan(α.sub.0x/α.sub.z) is the angle of the normal vector to the amplitude planes from the normal to the interface plane (z-axis in FIG. 3).

(34) A dispersion diagram relates the values of α and β, as per (2), as functions of frequency. The method to obtain the dispersion diagram is by solving for the zeros of the related characteristic equation whilst varying the frequency f. The characteristic equation for axially symmetric wave modes is given by:

(35) $\begin{array}{cc}\frac{{\mathrm{.Math.}}_{r1}}{k_1}\frac{J_1\left(k_1\rho \right)}{J_0\left(k_1\rho \right)}-\frac{{\mathrm{.Math.}}_{r2}}{k_2}\frac{H_1^{\left(2\right)}\left(k_2\rho \right)}{H_0^{\left(2\right)}\left(k_2\rho \right)}=0\mathrm{for}{\mathrm{TM}}_{0n}\frac{{\mu }_{r1}}{k_1}\frac{J_1\left(k_1\rho \right)}{J_0\left(k_1\rho \right)}-\frac{{\mu }_{r2}}{k_2}\frac{H_1^{\left(2\right)}\left(k_2\rho \right)}{H_0^{\left(2\right)}\left(k_2\rho \right)}=0\mathrm{for}{\mathrm{TE}}_{0n}& \left(3\right)\end{array}$
where μ is the magnetic permeability, ε is the electric permittivity, H.sub.n.sup.(2) is the Hankel function of the second kind of order n, J.sub.n is the Bessel function of the first kind of order n, ρ is the radius of the dielectric medium and k.sub.1 and k.sub.2 are given by:
.sub.1=.sub.0√{square root over (ϵ.sub.1μ.sub.1−.sup.2)}  (4)
.sub.2=.sub.0√{square root over (ϵ.sub.2μ.sub.2−.sup.2)}  (5)

(36) The Davidenko method is used to calculate the roots of (3) as given in [25].

(37) FIG. 5 shows a dispersion diagram for the real and imaginary parts of a complex wave number in a cylindrical body model with fat and skin tissue. The plot assumes a fat layer of radius 5 cm. The skin layer is assumed to be infinite homogenous lossy medium. This plot provides information regarding the type of wave modes that can be excited in different body tissue. The plot shows the following interesting features: 1. The {right arrow over (β)} in the fat layer is always less than |{right arrow over (k)}| in the skin layer indicating that the phase velocity of the wave propagating in the fat layer is faster than the speed of light. This permits the existence of leaky wave modes that can contribute to radiation. 2. There are specific frequencies for the transverse electric (TE) and transverse magnetic (TM) modes where the real and imaginary parts of the complex wavenumber are equal i.e. {right arrow over (α)}={right arrow over (β)} known as the ‘resonance gain condition’. This condition permits maximum radiation to leak out of a lossy dielectric waveguide when there is a radiating current element J({right arrow over (ρ)}) oriented along the axis of the cylinder. The specific frequencies where the {right arrow over (α)}={right arrow over (β)} condition holds are different for TE and TM polarization suggesting that using compound antennas would enable leaky modes to be generated in different frequency bands.

(38) Body morphology varies from one patient to the next and following implantation as a patient gains or loses weight and muscle mass. This results in differences in the electromagnetic environment surrounding the implant antenna. This confirms that the implant antenna designs proposed need to be robust enough to maintain performance for different body types.

(39) FIG. 6 shows the transverse electric (TE) and transverse magnetic (TM) dispersion diagram for 8 cm of fat and 20 cm of fat respectively (ρ.sub.fat=4 cm and ρ.sub.fat=10 cm respectively). FIG. 6 shows that the number of resonant frequencies where {right arrow over (α)}={right arrow over (β)} move depending on the thickness of the fat layer and whether the antenna is TE or TM. This suggests that having a compound field antenna that generates at least one TM and one TE mode wave orthogonal to each other implies that one of the two polarizations will excite leaky wave modes with maximum radiation efficiency present in the frequency band of interest.

(40) Based on the above observations of wave dispersion in body tissue a novel method has been developed for designing compound field antennas that exhibit such orthogonality of the TE and the TM polarization and in which leaky wave modes are generated, when the antennas are in operation, so as to enhance radiation efficiency.

(41) Considering first a implantable compound antenna comprising a dipole (TM) and a loop (TE) antenna inside a dispersive medium like the body tissue in which: (a) the loop antenna has an area A (in mm.sup.2) and a current I.sub.M (in mA) flowing through it; and (b) the dipole antenna has a length L and current I.sub.E; the phase of the electric and magnetic fields of a dipole and a loop inside the body tissue is given by:

(42) $\mathrm{Magnetic}\mathrm{Loop}$${\varphi }_E=\mathrm{kr}-\arctan \left(\mathrm{kr}\right)+\frac{\pi }{2}$${\varphi }_H=\mathrm{kr}-\arctan \left(\frac{\mathrm{kr}}{1-{\left(\mathrm{kr}\right)}^2}\right)$$\mathrm{Electric}\mathrm{Dipole}$${\varphi }_E=\mathrm{kr}-\arctan \left(\frac{\mathrm{kr}}{1-{\left(\mathrm{kr}\right)}^2}\right)+\frac{\pi }{2}$${\varphi }_H=\mathrm{kr}-\arctan \left(\mathrm{kr}\right)$
where the wave number k is given by equation (2) and r is the radial distance from the antenna. The phase difference between the two fields Δ is given by:

(43) $\begin{array}{cc}\Delta =\frac{\pi }{2}+{\left(-1\right)}^p\arctan \left({\left(\mathrm{kr}\right)}^3\right)& \left(6\right)\end{array}$
where p=1 for electric dipole (TM polarization) and 2 for magnetic dipole (TE polarization). The impedance of the dipole and loop normalized to the free space impedance of 120πΩ is given as:

(44) $\begin{array}{cc}{\hat{Z}}_E=\frac{{\left(\mathrm{kr}\right)}^2}{1+{\left(\mathrm{kr}\right)}^2}+j\frac{-1}{\left(\mathrm{kr}\right)\left(1+{\left(\mathrm{kr}\right)}^2\right)}& \left(7\right)\\ {\hat{Z}}_H=\frac{{\left(\mathrm{kr}\right)}^4}{{\left(\mathrm{kr}\right)}^4-{\left(\mathrm{kr}\right)}^2+1}+j\frac{\mathrm{kr}}{{\left(\mathrm{kr}\right)}^4-{\left(\mathrm{kr}\right)}^2+1}& \left(8\right)\end{array}$

(45) These equations suggest that very close to the antenna a small electric dipole appears like a high impedance capacitive open circuit and a small magnetic loop looks like a low impedance short circuit. The loop antenna has a magnetic dipole moment {right arrow over (p.sub.m)} and the dipole antenna has an electric dipole moment {right arrow over (p.sub.e)} which are related as follows:

(46) $\begin{array}{cc}\stackrel{.fwdarw.}{p_m}=A{\exp }^{\mathrm{jB}}Z\stackrel{.fwdarw.}{p_e}\mathrm{where}Z=\sqrt{\frac{\mu }{{ϵ}_r\left(\omega \right)}}& \left(9\right)\end{array}$
is the impedance of the dispersive medium, A is the amplitude ratio and B is the phase difference between the electric and magnetic dipole moments. B is related to the complex wave number of the medium as follows:

(47) $\begin{array}{cc}\cos \left(B\right)=\frac{{\left(\mathrm{kr}\right)}^3}{\sqrt{1+{\left(\mathrm{kr}\right)}^6}}& \left(10\right)\end{array}$

(48) The radiation efficiency of an antenna embedded in a dispersive medium is given by:

(49) $\begin{array}{cc}{\eta }_{\mathrm{eff}}=\{\begin{array}{cc}-\frac{\mathrm{Re}\left[Z\right]}{{\mathrm{Re}\left[j.Math.k\right]}^2{a}^2{\mathrm{ZH}}_l\left(\mathrm{ka}\right){\left({\left(H_l\right)}^{\prime }\left(\mathrm{ka}\right)+\frac{H_l\left(\mathrm{ka}\right)}{\mathrm{ka}}\right)}^{*}}& \mathrm{magnetic}\mathrm{antenna}\\ -\frac{\mathrm{Re}\left[Z\right]}{{\mathrm{Re}\left[j.Math.k\right]}^2{a}^2{\mathrm{ZH}}_l^{*}\left(\mathrm{ka}\right)\left(H_l^{\prime }\left(\mathrm{ka}\right)+\frac{H_l\left(\mathrm{ka}\right)}{\mathrm{ka}}\right]}& \mathrm{electric}\mathrm{antenna}\end{array}& \left(11\right)\end{array}$
where a is the radius of the smallest sphere that completely encloses the implant antenna, H.sub.l(ka) is the spherical Hankel function of the second kind with an order l. Physically l stands for the number of modes that the electric or magnetic current on the antenna will support and contribute to radiation.
Implant having Single Band Antenna

(50) FIGS. 7(a) and 7(b) illustrate the implant device 102 of FIG. 1 in more detail.

(51) As seen in FIG. 7, the implant device 102 comprises a housing (or ‘can’) having a header portion 102(a) and a body portion 102(b).

(52) The header portion 702(a) is fabricated from a suitable material such as, for example, a combination of tecothane and medical grade epoxy which has a relative dielectric constant of 3 to 5 F/m. The header portion comprises the antenna 106 while the body portion 102(b) houses, amongst other things, the antenna 106, a matching circuit 701, a transmitter 702, a controller 704 and one or more implant functions 706.

(53) The matching circuit 701 is electrically tunable to allow appropriate matching of the impedance of the antenna 106 to the transmitter 702 output. The transmitter 702 operates under the control of the controller 704 which also controls general operation of the IMD's implant functions 706 (e.g. its operation as a pacemaker, defibrillator, neurostimulator or the like).

(54) The antenna 106 is shown in more detail in FIG. 7(b). The antenna 106 can be manufactured using any suitable materials using any appropriate manufacturing technology. For example, the antenna 106 may be formed by laser cutting or stamping and may be fabricated from suitable metal, such as titanium, copper or platinum iridium, commonly used in medical implants. In a particularly advantageous example, the antenna is formed by laser direct structuring of an appropriate antenna material in a dielectric member in the header portion 102(a).

(55) The antenna 106 comprises a single dipole-loop pair where the dipole moments are fed in phase quadrature. The antenna 106 is configured for single band operation in the Bluetooth band (in the 2400 to 2480 MHz band) and comprises a compound antenna having a magnetic field radiator portion 710 and an electric field radiator portion 708.

(56) The magnetic field radiator portion 710 is the transverse electric (TE) component of the compound antenna 106 and is formed by a conducting member forming a ‘loop’ that is generally curved in shape. The loop 710 has two ends 710(a) and 710(b) arranged to form a generally open base portion that may be connected via appropriate circuitry to the transmitter 702. One end of the loop is a feed point 710(b) that is typically connected to a feed pin, or the like, whilst the other end provides a ground connection that is connected to a ground point 710(a). The magnetic field radiator portion 710 has a shape configured to generate a transverse electric leaky wave in a frequency of interest, in the lossy body tissue of the implant patient, as the current flows from the feed pin to the ground point. The ground connection may be provided via a pin connected to a ground potential on a circuit board inside the housing, or the entire housing could form the ground plane.

(57) The electric field radiator 708 is the transverse magnetic (TM) component of the compound antenna 106 and is formed by a curved or meandered conducting member that is located externally to the loop 710. The electric field radiator 708 comprises a stub portion 708′ coupled to the loop portion 710 at a connection point (x.sub.i, y.sub.i) on the perimeter of the loop that is specifically designed to cause the magnetic current flowing through the loop 710 and incident on the electric field radiator 708 to be reflected such that the reflection current amplitude is minimized and does not disrupt the magnetic current flowing in the loop 710 thereby resulting in a minimization of the phase difference between the electric and the magnetic fields generated in the lossy body tissue. Thus, the electric field radiator portion 708 is configured to generate a transverse electric and a transverse magnetic leaky wave, in a frequency of interest, in the lossy body tissue of the implant patient, thereby increasing the efficiency of the antenna structure.

(58) The antenna 106 is thus configured, in accordance with the principles described herein, to provide orthogonality of the of the TE and the TM polarization, to excite leaky wave modes in the fat and skin layer in operation, and to maximise far field gain of the transmitter. Specifically, the antenna 106 has a geometry in which connection point (x.sub.i, y.sub.i) of the stub along the perimeter of the loop portion 710 and the length of the stub L.sub.i is configured to tune the antenna to provide the desired orthogonality, leaky wave modes and far field gain.

(59) Implant having Dual Band Antenna

(60) FIGS. 8(a) and 8(b) illustrate another implant device 802 similar to that shown in FIG. 1.

(61) As seen in FIG. 8, like the device 102 of FIG. 7, the implant device 802 comprises a housing (or ‘can’) having a header portion 802(a) and a body portion 802(b).

(62) The header portion 802(a) is fabricated from a suitable material such as, for example, a combination of tecothane and medical grade epoxy which has a relative dielectric constant of 3 to 5 F/m. The header portion 802(a) comprises an antenna 806 while the body portion 802(b) houses, amongst other things, the antenna 806, a matching circuit 801, a transmitter 802, a controller 804 and one or more implant functions 808.

(63) The matching circuit 801 is electrically tunable to allow appropriate matching of the impedance of the antenna 806 to the transmitter 802 output. The transmitter 802 operates under the control of the controller 804 which also controls general operation of the implant device's implant functions 808 (e.g. its operation as a pacemaker, defibrillator, neurostimulator or the like).

(64) The antenna 806 is shown in more detail in FIG. 8(b). The antenna 806 can be manufactured using any suitable materials using any appropriate manufacturing technology. For example, the antenna 806 may be formed by laser cutting or stamping and may be fabricated from a suitable metals, such as titanium, copper or platinum iridium, commonly used in medical implants. In a particularly advantageous example, the antenna is formed by laser direct structuring of an appropriate antenna material in a dielectric member in the header portion 802(a).

(65) The antenna 806 comprises a dipole-loop pair with a gap capacitance which lowers the resonant frequency of the loop and thereby allows the size of the antenna 806 to be reduced. The antenna 806 is configured, by virtue of the gap capacitance, for dual band operation in the MICS Band (402 to 405 MHz) or the Bluetooth band (in the 2400 to 2480 MHz band) and comprises a compound antenna having a magnetic field radiator portion 810 and an electric field radiator portion 814.

(66) The magnetic field radiator portion 810 is the transverse electric (TE) component of the compound antenna 806 and is formed by a conducting member forming a ‘loop’ that is generally rectangular in shape. The loop 810 has two distinct parts, 810(a) and 810(b), arranged with an appropriately dimensioned gap 812 therebetween to provide a desired value for the gap capacitance.

(67) One part of the loop 810(a) generally adjacent the capacitive gap has a length that is appropriately chosen such that the input impedance of the loop matches the impedance of the radio chipset enabling maximum power transfer at a chosen resonant frequency. This part of the loop 810(a) is configured such that the first and second parts of the loop are appropriately located to minimize the phase difference between the electric field and the magnetic field that is generated by the antenna. Further, this part of the loop 810(a) is coupled the other part of the loop such that the first and second parts of the loop are appropriately located to generate multiple modes of transverse electric and transverse magnetic leaky waves such that the overall efficiency of the antenna is enhanced.

(68) The first part 810(a) of the magnetic field radiator portion 810 is formed at a ground end of the loop 810, whilst the second part 810(b) of the magnetic field radiator portion 810 is formed at a ground end of the loop 810. Thus, the end of the magnetic field radiator portion 810 at the end of the second part 810(b) provides a feed point that is typically connected to a feed pin, or the like. The other end of the magnetic field radiator portion 810 (at the end of the first part 810(a)) provides a ground connection that is connected to a ground point. The magnetic field radiator portion 810 has a shape configured to generate a transverse electric leaky wave in a frequency of interest, in the lossy body tissue of the implant patient, as the current flows from the feed pin to the ground point. The ground connection may be provided via a pin connected to a ground potential on a circuit board inside the housing, or the entire housing could form the ground plane.

(69) The electric field radiator 814 is the transverse magnetic (TM) component of the compound antenna 806 and is formed by a curved or meandered conducting member that is and located internally to the loop 810. The electric field radiator 814 is coupled to the loop portion 810 at a connection point (x.sub.i, y.sub.i) on the perimeter of the loop that is specifically designed to cause the magnetic current flowing through the loop 810 and incident on the electric field radiator 814 to be reflected with minimum amplitude such that it does not disrupt the magnetic current flowing in the loop 810 thereby resulting in a minimization of the phase difference between the electric and the magnetic fields generated in the lossy body tissue. Thus, the electric field radiator portion 814 is configured to generate a transverse electric and a transverse magnetic leaky wave, in a frequency of interest, in the lossy body tissue of the implant patient, thereby increasing the efficiency of the antenna structure.

(70) The antenna 806 is designed, in accordance with the principles described herein, to provide orthogonality of the of the TE and the TM polarization, to excite leaky wave modes in the fat and skin layer in operation, and to maximise far field gain of the transmitter 802 in either of the two bands of operation. Specifically, the antenna 802 has a geometry in which connection point (x.sub.i, y.sub.i) of the electric field radiator 814 along the perimeter of the loop portion 810 and the length of the electric field radiator 808 is configured to tune the antenna to provide the desired orthogonality, leaky wave modes and far field gain.

(71) Design Methodology

(72) FIG. 9 is a flow diagram showing, by way of example only, one possible procedure that may be used to design an antenna such as the antennas described with reference to FIGS. 7 and 8.

(73) After the process starts, at S900, a preliminary antenna design is generated at S912, based on a design specification 908 taking account of one or more design goals 910.

(74) The design specification 908 specifies, for example, one or more of the following: the required range of the implant; the size constraints on the implant antenna; the use case for the implant for which the antenna is being designed; any geometric constraints on the implant antenna; a tissue type at implant location; the size of the implant header; any additional metal structures such as a lead assembly that maybe present in the implant header; header material properties; approximate lead arrangement following surgery; and the final orientation of the implant in the patient pocket.

(75) The design goals 910 are set based on desired antenna performance requirements which are based on link budget calculations that take into account the use case scenarios, implant power budget and the transceiver radio chip characteristics. Typically, for example, the main antenna design goals are antenna input impedance, antenna field of view or half power beam width, and far field gain.

(76) It will be appreciated that not all of the input specifications (and design goals) may be available at the time of initial design (or they may change following the initial design) of the antenna and the method is not limited to having all the specification parameters and/or goals specified.

(77) The initial antenna geometry may similar to that shown in FIG. 7 or 8 or may be some other appropriate antenna design.

(78) Next, a representative body model for the target patient is generated, at S914, with appropriate electromagnetic properties based on appropriate information from the design specification 908 and any other appropriate information (where available) such as patient gender, weight, body characteristics or the like. It will be appreciated that this may be generated at any appropriate time (before, after or in parallel with the initial antenna design).

(79) The body model and initial design provide inputs to a simulator which simulates, at S916, the performance of the antenna in a body of the type represented by the body model. The simulator, in this example, comprises a high frequency simulator that uses finite element or finite difference methods to calculate amplitude and phase of the electric and magnetic fields vectors on the surface of a sphere of radius Δ/2π, where λ is the guided wavelength in the tissue of implantation. The antenna input impedance and the far field gain of the antenna are calculated and compared to the design goals, at S918, in the frequency band of interest.

(80) If the design goals are not met at S918, then the geometry of the antenna is altered, at S924, by changing: (1) the connection point (xi, yi) of the electric field radiator (or stub) along the perimeter of the loop and (2) changing the length of the electric field radiator (or stub) L.sub.i. This changes the phase B between the electric and magnetic dipole moments and the antenna impedances, Z.sub.E and Z.sub.H. The adjusted design is then simulated, at S916, and the simulation results compared with the design goals at S918 as described previously.

(81) The adjusting, simulation and comparison steps are repeated iteratively until the design goals are met.

(82) If the initial design, or when an adjusted design, meets the design goals at S918, then the antenna design meeting the goals is fabricated and tested at S920. The antenna is fabricated using standard metal stamping, laser cutting methods or the like. The materials commonly used for implant antennas is titanium, copper or alloys such as platinum-iridium. The fabricated antenna is placed in the header of the implant assembly in which it is to be used. Once fabricated, the implant assembly is tested in an anechoic chamber where the implant is placed in a body ‘phantom’ comprising a vat filled with tissue mimicking gels.

(83) At 2.4 GHz-2.5 GHz commonly used tissue mimicking gels include: For skin: SPEAG HBBL1900-3800V3 ϵ.sub.r=39.2 F/m; α=1.8 S/m For fat: SPEAG LCL 2450 V1 ϵ.sub.r=5 F/m; α=0.25 S/m For muscle: SPEAG MBBL1900-3800V3 ϵ.sub.r=52.7 F/m; α=1.95 S/m

(84) The measured performance parameters of the antenna are compared to the simulated results. If necessary, the antenna design is further adjusted (via steps S924, S916, S918) and modified designs fabricated (at S920) till the measured goals match or exceeds the design goals.

(85) Experimental/Simulation Results

(86) Table 1 below shows the percentages of power that is lost in the various body tissue layers and net radiated power coupled out of the body for the implant antenna described with reference to FIG. 7, and for a standard loop antenna at 2450 MHz.

(87) TABLE-US-00001 TABLE 1 Comparison of the radiated power of the compound field antenna design of FIG. 7 to a standard loop antenna having similar dimensions. Example 1 Dissipated power Loop antenna (FIG. 7) in the core Body 65.19%  61.43%  in the fat 34.61%  31.63%  In the skin 0.14% 2.73% Radiated power in air 0.06% 4.21% Radiated power in/through the body  100%  100%

(88) As seen in Table 1, the implantable compound antenna design results in significantly higher radiated power in the air than the standard loop antenna. It has been found that the radiation efficiency of the compound field antenna is greater than twenty times that of the loop antenna (linear scale).

(89) FIGS. 10(a) and 10(b) show measured antenna gain as a function of angle in the elevation plane for a compound antenna. The gain is measured in both the vertical (FIG. 10(a)) and horizontal (FIG. 10(b)) polarizations relative to the gravity vector. In FIG. 10, the two plots shown include a first plot 1010 representing active measurements, done using a CSR1010 Bluetooth SMART™ chipset, and a second plot 1020 representing passive measurements, done using a network analyzer. The receive antenna is a standard gain horn resonant in the 2300 MHz to 2500 MHz band. For the active measurement the receiver is a spectrum analyzer that measures peak received power in dBm. The plots clearly show that the antenna is linearly polarized in the vertical direction (along the gravity vector).

(90) FIG. 11 shows the real 1110 and imaginary 1120 parts of the input impedance of the antenna as a function of frequency.

(91) FIG. 12 shows the simulated 1210 and measured 1220 return loss of the antenna as a function of frequency. The return loss indicates how well an antenna's impedance is matched to the transmitter's impedance. In the example shown the common transmitter impedance is 50Ω.

Modifications and Alternatives

(92) Detailed embodiments have been described above. As those skilled in the art will appreciate, a number of modifications and alternatives can be made to the above embodiments whilst still benefiting from the inventions embodied therein.

(93) It will be appreciated, for example, that other shapes of the antenna may be used in dependence on requirements of the application in which the antenna is employed.

(94) Various other modifications will be apparent to those skilled in the art and will not be described in further detail here.

REFERENCES

(95) A general background to antenna design and the technology discussed herein can be found in the following documents: [1] Kiourti. A., K. A. Psathas, K. S. Nikita, Implantable and Ingestible Medical Devices with Wireless Telemetry Functionalities: A Review of Current Status and Challenges, Wiley Bioelectromagnetics, vol. 35, issue 1, pp. 1-15, January 2014. [2] Li et al., Inverted e antenna with capacitance loading for use with an implantable medical device, U.S. patent application Ser. No. 13/538,501, 2012, published as U.S. 2014/0002314 (2014). [3] Mashiach, A. and Mueller, C., Antenna providing variable communication with an implant, CA Patent App. CA 2,850,445, 2013. [4] Amundson, M. D. and Von Arx, J. A. and Linder, W. J. and Rawat, P. and Mass, W. R., Circumferential antenna for an implantable medical device, U.S. Pat. No. 6,614,406, 2003. [5] Renken, G. W., Implant device telemetry antenna, U.S. Pat. No. 6,009,350, 1999. [6] D. M. Grimes and C. A. Grimes, Minimum Q of electrically small antennas: a critical review, Microwave Optical Tech. Lett., vol. 28, pp. 172-177, February 2001. [7] M. J. Underhill and M. J. Blewett, Unidirectional tuned loop antennas using combined loop and dipole modes, in Proc. Inst. Elect. Eng. 8th Int. Conf. HF Radio Systems and Techniques, July 2000, Pub. 474, pp. 37-41. [8] D. H. Kwon, On the Radiation Q and the Gain of Crossed Electric and Magnetic Dipole Moments, IEEE Transactions on Antennas and Propagation, Vol. 53, No. 5, May 2005. [9] D. M. Pozar, New results for minimum Q, maximum gain, and polarization properties of electrically small arbitrary antennas, 3rd European Conference on Antennas and Propagation, pp. 1993, 23-27 Mar. 2009. [10] McLean, J. S., P×M antenna with improved radiation characteristics over a broad frequency range, U.S. Pat. No. 7,388,550, 2008. [11] Grimes, C. A., Grimes, D. M., Tefiku, F. and Lui, G., Electrically-small low Q radiator structure and method of producing EM waves therewith, U.S. Pat. No. 6,437,750, 2002. [12] Brown, F. J., Planar compound loop antenna, U.S. Pat. No. 8,144,065, 2012. [13] Orsi et. al., Capacitively coupled compound loop antenna, US 20130113666, May 9, 2013. [14] S. O. Driscoll, A. S. Y. Poon, T. H. Meng, Wireless power transmission for implantable medical devices, U.S. Pat. No. 8,634,928 B1, Jan. 21, 2014. [15] A. S. Y. Poon, Electromagnetic field focusing for short-range wireless power transmission, Proc. IEEE Radio and Wireless Symposium (RWS), Santa Clara, Calif., January 2012. [16] Ho, J. S., A. J. Yeh, E. Neofytou, S. Kim, Y. Tanabe, B. Patlolla, R. E. Beygui, and A. S. Y. Poon, Wireless power transfer to deep-tissue microimplants, PNAS, 111, 7974-7979 (2014). [17] S. Gabriel, R. W. Lau, and C. Gabriel, The Dielectric Properties of Biological Tissues: III. Parametric models for the Dielectric Spectrum of Tissues, Phys. Med. Bio., vol. 41, pp. 227193, November 1996. [18] Orfanidis, S. J., Electromagnetic waves and antennas, Online Publication, 21 Jun. 2004 [19] Karlsson, A., Physical limitations of antennas in a lossy medium, IEEE Trans. on Antennas and Propagation, Vol. 52, pp. 2027-2033, 2004. [20] R. W. P. King and G. S. Smith, Antennas in Matter, Cambridge, London, U.K.: MIT Press, 1981. [21] Harrington, R. F., Time Harmonic Electromagnetic Fields. New York: McGraw-Hill, 1961. [22] G. Lovat, P. Burghignoli, and D. R. Jackson, Fundamental properties and optimization of broadside radiation from uniform leaky-wave antennas, IEEE Trans. Antennas Propagat, vol. 54, no. 5, pp. 14421452, May 2006. [23] A. Ip, and D. R. Jackson, Radiation from cylindrical leaky waves, IEEE Trans. Antennas Propagat., vol. 38, no. 4, pp. 482488, April 1990. [24] Kim, K. Y., Guided and leaky modes for circular open electromagnetic waveguides: Dielectric, plasma and metamaterial column, Ph.D. Thesis, December 2004. [25] Hanson, G. W. and A. B. Yakovlev, An analysis of leaky wave dispersion phenomena in the vicinity of cutoff using complex frequency plane singularities, Radio Science, Vol. 33, No. 4, 803-819, 1998. [26] T. Needham, Visual Complex Analysis, Oxford University Press, 1999. [27] Schantz, H. G., Near Field Phase Behaviour, In proceeding of: Antennas and Propagation Society International Symposium, 2005 IEEE, Volume: 3B. [28] also cited as [8] Kwon, D. H., On the radiation Q and the gain of crossed electric and magnetic dipole moments, Antennas and Propagation, IEEE Transactions on, vol. 53, no. 5, pp. 1681, 1687, May 2005.