Aluminum nitride combined overtone resonators for the mmWave spectrum

Abstract

A resonator system is provided in which a combined overtone resonator device is excited with a two-dimensional mode of mechanical vibration in a cross sectional plane of a piezoelectric plate in response to an alternating voltage applied through an interdigitated electrode. The cross sectional plane extends along the width direction and the thickness direction, and the two-dimensional mode of mechanical vibration is a two-dimensional combined overtone mode of second and third order asymmetrical Lamb-wave overtones.

Claims

1. A resonator system comprising: a combined overtone resonator device comprising: a piezoelectric plate suspended from a substrate, the piezoelectric plate having a top surface, a bottom surface, a length direction, a width direction and a thickness direction between the top surface and the bottom surface, the top surface and the bottom surface extending along the length direction and the width direction, and a first interdigitated electrode comprising a plurality of conductive strips disposed on one of the top surface or the bottom surface of the piezoelectric plate, wherein each of the plurality of conductive strips has an electrode width substantially along the width direction, and wherein the plurality of conductive strips are arranged with a pitch substantially along the width direction; and circuitry in communication with the combined overtone resonator device to apply an alternating voltage through the first interdigitated electrode to excite a two-dimensional mode of mechanical vibration in a cross sectional plane of the piezoelectric plate, the cross sectional plane extending along the width direction and the thickness direction, the two-dimensional mode of mechanical vibration comprising a two-dimensional combined overtone mode of second and third order asymmetrical Lamb-wave overtones.

2. The system of claim 1, further comprising a second interdigitated electrode including a plurality of conductive strips disposed on another of the top surface or the bottom surface of the piezoelectric plate, wherein adjacent ones of the conductive strips on the top surface have opposite polarities, and ones of the conductive strips on the bottom surface have a same polarity as a most closely overlying one of the conductive strips on the top surface; and the circuitry is in communication with the combined overtone resonator device to apply the alternating voltage through the first interdigitated electrode and the second interdigitated electrode to excite the two-dimensional mode of mechanical vibration in the cross sectional plane.

3. The system of claim 1, further comprising a second plate electrode disposed on another of the top surface or the bottom surface of the piezoelectric plate.

4. The system of claim 1, wherein the pitch of the first interdigitated electrode is equal to the thickness of the piezoelectric plate.

5. The system of claim 1, wherein the pitch of the first interdigitated electrode is within 50% of the thickness of the piezoelectric plate.

6. The system of claim 1, wherein the electrode width of each conductive strip is in a range of 10 nm to 10 μm.

7. The system of claim 1, wherein the thickness of the piezoelectric plate is in a range of 100 nm to 100 μm.

8. The system of claim 1, wherein the piezoelectric plate is selected from aluminum nitride, lithium niobate, lithium tantalite, zinc oxide, gallium nitride, scandium nitride, aluminum scandium nitride, and quartz, and combinations thereof.

9. The system of claim 1, wherein the first interdigitated electrode is selected from aluminum, ruthenium, molybdenum, and tungsten and combinations thereof.

10. The system of claim 1, wherein a frequency of the two-dimensional combined overtone mode is in a range of 1 MHz to 100 GHz.

11. The system of claim 1, wherein the combined overtone resonator device has an electro-mechanical coupling efficiency of 0.1% to 10.0%.

12. The system of claim 1, wherein the combined overtone resonator device has a quality factor of at least 200 at a frequency of 6 GHz or greater.

13. The system of claim 1, wherein the combined overtone resonator device is operative in a frequency range of 6 GHz to 40 GHz.

14. The system of claim 1, wherein the circuitry comprises filter circuitry, circulator circuitry, or oscillator circuitry.

15. A filter comprising a plurality of combined overtone resonator devices of claim 1, the plurality of combined overtone resonator devices formed on the substrate, each device tuned to a different frequency bandwidth.

16. A filter comprising: a plurality of combined overtone resonator devices, the plurality of combined overtone resonator devices formed on a single substrate, each device tuned to a different frequency bandwidth, each device comprising: a piezoelectric plate suspended from a substrate, the piezoelectric plate having a top surface, a bottom surface, a length direction, a width direction and a thickness direction between the top surface and the bottom surface, the top surface and the bottom surface extending along the length direction and the width direction, and a first interdigitated electrode comprising a plurality of conductive strips disposed on one of the top surface or the bottom surface of the piezoelectric plate, wherein each of the plurality of conductive strips has an electrode width substantially along the width direction, and wherein the plurality of conductive strips are arranged with a pitch substantially along the width direction; and circuitry in communication with each of the combined overtone resonator devices to apply an alternating voltage through the first interdigitated electrode to excite a two-dimensional mode of mechanical vibration in a cross sectional plane of the piezoelectric plate, the cross sectional plane extending along the width direction and the thickness direction, the two-dimensional mode of mechanical vibration comprising a two-dimensional combined overtone mode of second and third order asymmetrical Lamb-wave overtones.

17. The filter of claim 16, wherein the pitch of the first interdigitated electrode of each device is different from the pitch of the first interdigitated electrodes of others of the plurality of devices.

18. The filter of claim 16, wherein the frequency bandwidth of each device ranges from 300 to 600 MHz.

19. A combined overtone resonator device comprising: a piezoelectric plate suspended from a substrate, the piezoelectric plate having a top surface, a bottom surface, a length direction, a width direction, and a thickness direction between the top surface and the bottom surface, the top surface and the bottom surface extending along the length direction and the width direction; and a first interdigitated electrode comprising a plurality of conductive strips disposed on the top surface of the piezoelectric plate, a second interdigitated electrode comprising a plurality of conductive strips disposed on the bottom surface of the piezoelectric plate, wherein each of the conductive strips has an electrode width substantially along the width direction, and wherein the plurality of conductive strips are arranged with a pitch substantially along the width direction; adjacent ones of the conductive strips on the top surface have opposite polarities, and each of the conductive strips on the bottom surface has a same polarity as a most closely overlying one of the conductive strips on the top surface; and wherein a two-dimensional mode of mechanical vibration is excited in a cross sectional plane of the piezoelectric plate in response to an alternating voltage applied through the interdigitated electrode and the second interdigitated electrode, the cross sectional plane extending along the width direction and the thickness direction, the two-dimensional mode of mechanical vibration comprising a two-dimensional combined overtone mode of second and third order asymmetrical Lamb-wave overtones.

20. A filter comprising a plurality of combined overtone resonator devices of claim 19, the devices formed on the substrate, each device tuned to a different frequency bandwidth.

Description

DESCRIPTION OF THE DRAWINGS

(1) FIG. 1A illustrates the FEM-simulated modeshape of two periodic units (N=2) of an AlN Combined Overtone Resonator (COR) for t.sub.AlN=1 μm and W=1 μm. This COR operates at 9771 MHz.

(2) FIG. 1B illustrates the displacement components in x- and z-directions of the simulations of FIG. 1A. The effect of electrodes was not included in the simulations. Normalized ; and along the lateral direction (left) and the thickness direction (right) of COR.

(3) FIG. 2 illustrates a dispersion curve and K.sup.2-values of the COM excited in a 1 μm-thick AlN plate for different values of its width (W). The highest K.sup.2 value and its corresponding frequency are marked by the stars. The modeshapes for W=1 μm and W=0.7 μm are shown in the insets.

(4) FIGS. 3A, 3B, and 3C illustrate electric field distributions for electrical configurations (FIG. 3A) LFE, (FIG. 3B) TEF-1 and (FIG. 3C) TFE-2. The directions and length of arrows represent the direction and strength of the electric field vectors. The bold lines represent the electric field component E.sub.z (in the mid-plane of each unit) across the thickness, and on the right side of the dash-dotted line indicates positive sign and vice versa.

(5) FIG. 3D illustrates the asymmetric stress field distributions for CORs without the mass loading of the IDT.

(6) FIG. 4A illustrates E.sub.z for TFE-2.

(7) FIGS. 4B and 4C illustrate S.sub.x distribution in the center of a periodic unit of CORs (FIG. 4B) without and (FIG. 4C) with the mechanical effects of electrodes.

(8) FIGS. 4D and 4E illustrate S.sub.z distribution in the center of a periodic unit of CORs (FIG. 4D) without and (FIG. 4E) with the mechanical effects of electrodes.

(9) FIGS. 5A and 5B illustrate k.sub.t.sup.2-values and resonant frequencies f.sub.r for different Al thickness (t.sub.Al) of CORs based on (FIG. 5A) TFE-2 and (FIG. 5B) LFE configurations. AlN thickness (t.sub.AlN) is fixed at 1 μm. W=1.1 μm, N=2, α=50%, t.sub.Al=110 nm for TFE-2. W=1.15 μm, N=2, α=35%, t.sub.Al=120 nm for LFE.

(10) FIG. 6 illustrates a microfabrication process for LFE CORs.

(11) FIG. 7 illustrates SEM image of (a) CORs, (b) zoom-in of COR IDT, and (c) etching sidewall of COR. The device comprises an array of 8 resonators to obtain a high static capacitance value for matching 50Ω termination.

(12) FIGS. 8A and 8B illustrate (FIG. 8A) measured and (FIG. 8B) FEM-simulated admittances of the sub-optimal COR design experimentally demonstrated: t.sub.AlN=1 μm, t.sub.Al=85 nm, W=1.15 μm, α=57%, SW angle=60°.

(13) FIG. 9 illustrates a summary of Q.sub.l-values for various piezoelectric resonator technologies past 6 GHz, where the star indicates the present technology. (M. Hara et al., “Super-High-Frequency Band Filters Configured with Air-Gap-Type Thin-Film Bulk Acoustic Resonators,” Jpn. J Appl. Phys., vol. 49, no. 7, p. 07HD13, July 2010. Y. Yoshino, “Piezoelectric thin films and their applications for electronics,” J. Appl. Phys., vol. 105, no. 6, p. 061623, March 2009. Y. Yang, R. Lu, T. Manzaneque, and S. Gong, “Toward Ka Band Acoustics: Lithium Niobate Asymmetrical Mode Piezoelectric MEMS Resonators,” in 2018 IEEE International Frequency Control Symposium (IFCS), Olympic Valley, C A, 2018, pp. 1-5. T. Yokoyama, M. Hara, M. Ueda, and Y. Satoh, “K-band ladder filters employing air-gap type thin film bulk acoustic resonators,” in 2008 IEEE Ultrasonics Symposium, Beijing, China, 2008, pp. 598-601. E. Iborra, M. Clement, J. Capilla, J. Olivares, and V. Felmetsger, “Optimization of thin AlN sputtered films for X-band BAW resonators,” in 2010 IEEE International Ultrasonics Symposium, San Diego, Calif., USA, 2010, pp. 1688-1691. G. Chen and M. Rinaldi, “High-Q X Band Aluminum Nitride Combined Overtone Resonators,” in 2019 Joint Conference of the IEEE International Frequency Control Symposium &European Frequency and Time Forum (IFCS-EFTF 2019), Orlando, Fla., 2019.)

(14) FIG. 10 illustrates operating frequencies versus AlN thickness for CORs and BAW resonators. The metal electrodes were not included in the FEM models used in the simulations.

(15) FIG. 11 illustrates simulated relative frequency shift Δf.sub.r for different relative variations of the resonator thickness, Δt.sub.AlN. Electrodes are not included in the simulations for a fair comparison.

(16) FIG. 12 illustrates simulated resonant frequency and k.sub.t.sup.2-values for different W-values for CORs using TFE-2 configuration around 24 GHz. The data points on W=360 nm and 540 nm are missing due to spurious modes.

(17) FIG. 13 illustrates simulated resonant frequency and k.sub.t.sup.2-values for different W-values for CORs using LFE configuration around 24 GHz. The data points on W=360 nm and 560 nm are missing due to spurious modes.

(18) FIG. 14 illustrates FEM simulated performance of 5 contiguous ladder filters synthesized using lithographically defined multi-frequency TFE-2 CORs operating at ˜24 GHz. Each filter has a BW>300 MHz covering an aggregated BW>1.6 GHz.

(19) FIG. 15 illustrates FEM simulated performance of 7 contiguous ladder filters synthesized using lithographically defined multi-frequency LFE CORs operating at ˜24 GHz. Each filter has a BW>150 MHz covering an aggregated BW>1 GHz.

(20) FIG. 16 illustrates simulated insertion loss (I.L.) for 3.sup.rd order ladder filers made by LFE (red triangles) and TFE-2 (blue squares) CORs for different Q.

(21) FIG. 17 illustrates SEM image of a fabricated 23.2 GHz COR.

(22) FIG. 18 is a graph of measured admittance for the COR of FIG. 17.

DETAILED DESCRIPTION

I. Introduction

(23) The development of wireless communication systems has been steadily growing and conventional sub-6 GHz frequency bands are too congested to meet the high data rate requirements of several emerging technologies. For example, connected-vehicle-to-everything communication (V2X) systems have been recently developed to allow vehicles to communicate with moving parts of the traffic system around them to increase awareness of all surroundings, reduce the risk of collisions, and maximize the transportation efficiency. This technology requires high-bandwidth, low-latency and high-reliability short-range wireless links to communicate with compatible systems on vehicles, pedestrians and infrastructures. The 5G cellular network can be considered the key enabler for such a ubiquitous and pervasive mobile internet connectivity. In particular, the use of the millimeter wave (mmWave) spectrum represents a major leap forward in the 5G network, as it enables improvements in data speed, capacity, quality and latency that are unimaginable in 3G and 4G networks. 5G frequency bands from 6 GHz to 40 GHz are typically referred to as cm-mmWave in the industry. Unlike sub-6 GHz spectrum with bandwidth (BW) typically between 5 MHz and 20 MHz, 5G cm-mmWave spectrum provides contiguous bandwidths (BWs) from 40 MHz up to 2 GHz. This high-band spectrum enables data rates in the tens of Gbps range with extremely low latency, providing significant opportunities for very high throughput services, such as Enhanced Mobile Broadband (eMBB), Ultra Reliable Low Latency Communications (URLLC), and Massive Machine Type Communications (mMTC). It is worth noting that the first set of high-band auctions, concluded by the Federal Communications Commission (FCC) in May 2019, offered more than 2,900 Upper Microwave Flexible Use Service licenses in the 24 GHz band: the lower segment of the 24 GHz band (24.25-24.45 GHz) is licensed as two 100-megahertz blocks, while the upper segment (24.75-25.25 GHz) is licensed as five 100-megahertz blocks. These 5G 24 GHz bands are closely located to the 23.8 GHz band that is used for sensitive meteorological and oceanographic measurements; therefore, the adoption of these 5G bands in communication systems requires the use of pass-band filters with a relatively small fractional bandwidth of ˜0.42% (i.e. 100 MHz) and a high Q>500 to achieve the steep roll-off and the large out-of-band rejection needed to enable coexistence with the adjacent band.

(24) Therefore, the technology described herein can provide a solution to address this 5G mmWave filter challenge. The technology provides high-performance filters for the mmWave that can enable the miniaturization and integration of 5G systems desirable for a ubiquitous and pervasive mobile internet connectivity.

(25) Although the electromagnetic (EM) wavelengths are significantly reduced at mmWave, the dimensions of conventional EM based filters are still more than 600 times larger than what could be achieved with the acoustic counterparts, making them not suitable for the implementation of next generation miniaturized mobile devices. Also, the low quality factor of EM filters results in poor roll-off which renders the system vulnerable to crosstalk from adjacent channels.

(26) Micro-electro-mechanical system (MEMS) components, especially Surface Acoustic Wave (SAW) and Bulk Acoustic Wave (BAW) resonators, have been employed as radio frequency (RF) filters in current mobile devices for frequency band selection due to the high-quality factor, high electromechanical coupling coefficient (corresponding to high fractional BW) and small form-factor that they can achieve in the sub-6 GHz range. Nevertheless, when scaled above 6 GHz, all the existing micro-acoustic resonator technologies resonators have suffered critical limitations associated with increased acoustic losses and aggressively scaled dimensions that have prevented the synthesis of high performance filters based on these technologies.

(27) SAW filters operating at 15 GHz were demonstrated employing 145 nm wide interdigital (IDT) electrodes. However, an insertion loss >40 dB was measured for these devices due to their heavily degraded Q (commercially available SAW filters show I.L.<1 dB in 1 GHz range). Aluminum nitride (AlN) BAW resonators in X band and K band were also demonstrated. Although these high frequency BAW resonators maintain an electromechanical coupling coefficient (k.sub.t.sup.2)>6% (enabled by the high piezoelectric coefficient, e.sub.33, employed to transduce vibration), they show very low Q-values (˜300), which are 10 times lower than the ones typically achieved by the same technology in the sub-6 GHz frequency bands, resulting in the synthesis of filters with high insertion loss of 3.8-11 dB (I.L.∝k.sub.t.sup.2.Math.Q). This drastic performance degradation is associated with the fact that ultra-thin piezoelectric and metal layers are required to achieve thickness-extensional vibration at mmWave frequencies. For example, the excitation of a 24 GHz thickness-extensional mode of vibration requires the use of an ultra-thin material stack composed of a 120 nm thick AlN piezoelectric layer and 17 nm thick ruthenium (Ru) electrodes. Studies have shown that the first few tens of nanometers of a deposited AlN thin-film are not properly oriented along the c-axis, resulting in degraded piezoelectric coupling and quality factor Q. In particular, it was experimentally demonstrated that the X-Ray diffraction (XRD) full width at half-maximum (FWHM, an indication of crystal quality with smaller values corresponding to better crystal orientation) increases dramatically when the deposited AlN film is thinner than 250 nm. For example, while the FWHM increases by 10% when the AlN film is scaled from 1000 nm to 250 nm, the degradation is more than 20% when it is scaled to 100 nm. In contrast, the present technology can employ AlN films thicker than 200 nm in the manufacture of micro-acoustic resonators in order to avoid the consequences of thin-film material degradation. In addition to the issues associated with the use of reduced quality ultra-thin piezoelectric films, the performance of the resonator is also significantly affected by the use of ultra-thin metal electrodes that unavoidably introduce a large electrical resistance that dramatically reduces the loaded Q of the resonator. For these reasons, to date AlN BAW resonators, or any other piezoelectric resonators, have not been successfully scaled to operate above 6 GHz without a significant performance degradation which has prevented the implementation of performing mmWave filters.

(28) In order to synthesize performing mmWave filters, resonators with high Q-values larger than 500 are required. The use of high Q resonators enables the achievement of the steep roll-off required for the implementation of contiguous filters separated by a minimum guard band for efficient spectrum usage. The use of high-Q resonators also enables more flexibility in the filter design with trade-off possibilities between filter rejection and bandwidth. Furthermore, the insertion loss (I.L.) heavily depends on the resonator Q when resonators with relatively modest k.sub.t.sup.2 values are employed to synthesize relatively narrow fractional bandwidth filters (FBW) (k.sub.t.sup.2<0.8% for 100 MHz BW at 24 GHz).

(29) In this context, overtone resonators are advantageous as they typically achieve high Q at very high frequencies due to reduced air damping and acoustic loss. However, their k.sub.t.sup.2-values are dramatically reduced as they scale inversely proportional to the cube of the order number.

(30) The technology described herein provides a class of acoustic resonators, aluminum nitride (AlN) combined overtone resonators (CORs), that can overcome the aforementioned fundamental challenges. CORs rely on the piezoelectric multimodal excitation of two higher-order Lamb waves (the 2.sup.nd and 3.sup.rd order Asymmetrical Lamb Waves, i.e. A2 and A3) to transduce a 2-dimensional (2D) mechanical mode of vibration in the cross-section of a suspended thin-film AlN plate. A combination of multiple overtone modes is employed to achieve a more efficient piezoelectric transduction. Exploiting the multimodal excitation of combined overtones, CORs can operate in the mmWave spectrum while maintaining relaxed lithographic requirements (minimum feature>100 nm) and relatively thick AlN films (>220 nm), which eliminate the performance degradation issues associated with ultra-thin film materials and directly translates in the achievement of high Q>1000. Furthermore, thanks to the coherent combination of the e.sub.33 and e.sub.15 piezoelectric coefficients of AlN employed to transduce the combined overtone 2D mechanical mode of vibration, CORs achieve relatively high electromechanical coupling coefficient k.sub.t.sup.2 (0.8% to 1.9%) despite the use of higher order modes (i.e. overtones) in the structure. Also, due to the dependence of this combined overtone 2D vibrational mode on both the vertical and the lateral dimensions of the structure, the resonant frequencies of CORs can be lithographically tuned to synthesize monolithic multi-frequency filters on the same chip with minimal fabrication complexity. Therefore, mmWave CORs can simultaneously achieve high Q-values >1000 and relatively high electromechanical coupling coefficient k.sub.t.sup.2 (0.8%˜1.9%) suitable for the implementation of contiguous filters (bandwidth from 300 to 592 MHz) for aggregated bandwidth from 1.6 to 2.6 GHz on the same substrate in the 24-40 GHz frequency range.

II. Principle of Operation

(31) Lamb wave theory has been studied to identify all vibration modes that can be described by a one-dimensional (1-D) displacement vector in a plate. This theoretical model for analyzing and optimizing Lamb wave resonators is adopted here to describe the 2-D motion of aluminum nitride (AlN) combined overtone resonators (CORs).

(32) AlN CORs are formed by a suspended AlN thin film and use one (i.e. top electrode-only configuration) or two (i.e. top and bottom electrodes configuration) interdigital transducer (IDT) electrodes for the piezoelectric multimodal excitation of the 2.sup.nd and 3.sup.rd order asymmetrical Lamb-wave overtones (A2 and A3 respectively), resulting into a 2-dimensional (2D) combined overtone mode (COM) of vibration in the cross-section of a suspended thin-film AlN plate. This COM is excited in the structure when the thickness of the AlN layer, t.sub.AlN, is approximately equal to the pitch, W, of the employed IDT. Such a COM is characterized by a 2D displacement vector with components along both the lateral () and vertical () directions of the plate. It is worth noting that , corresponds to the vibration modeshape of the A3 mode while , corresponds to the vibration modeshape of the A2 mode. The general expression of () and () for the A2-A3 COM is provided in (1), assuming infinite periodic boundary conditions in both directions:

(33) $\begin{array}{cc}\left[\begin{array}{c}\\ \end{array}\right]=\left[\begin{array}{c}\stackrel{\_}{X}A\left(x\right)B\left(z\right)\\ \stackrel{\_}{Z}C\left(x\right)D\left(z\right)\end{array}\right]=\left[\begin{array}{c}-\stackrel{\_}{X}\cos \left({\beta }_{x}x\right)\cos \left({\beta }_{z\_x}z\right)\\ -\stackrel{\_}{Z}\sin \left({\beta }_{x}x\right)\cos \left({\beta }_{z\_z}z\right)\end{array}\right]& \left(1\right)\end{array}$
where X and Z are the magnitudes of the displacement components along the x- and z-directions, respectively; β.sub.x is the wave-vector relative to the motion along the x-directions for both and ; β.sub.z_x and β.sub.z_z are the wave-vectors relative to the motion along the z-directions for and , respectively. The expressions describing the three wave-vectors are provided in (2), (3), and (4).

(34) $\begin{array}{cc}& \left(2\right)\\ & \left(3\right)\\ & \left(4\right)\end{array}$
where W is the pitch of interdigital transducer (IDT) electrodes employed to transduce the COM and t.sub.AlN is the thickness of the AlN plate. Note that all the three wave vectors have different mode orders, resulting in vortexes in the total displacements field.

(35) The vibration modeshape of the COM was simulated by finite-element method (FEM) in COMSOL, as shown in FIGS. 1A and 1B. The simulation was performed using two periodic unit cells (finger number N=2) with periodic boundary conditions along 2 and stress-free boundary conditions in {circumflex over (z)}. The FEM simulated modeshape of the COM in FIG. 1A matches well to the analytical description by equations 1-4 with small deviations along the z-direction due to the lack of periodic boundary conditions (i.e. stress-free boundaries on the top and bottom surfaces of the FEM model). The FEM simulated modeshapes of both and are also reported in FIG. 1A highlighting the clear matching with the displacement profiles of the A3 and A2 modes, respectively.

(36) A. Piezoelectric Coupling Efficiency

(37) Multiple parameters have been used in the literature to describe the piezoelectric coupling efficiency of resonators. Particularly important is the electromechanical coupling coefficient, k.sub.t.sup.2, which is a quantitative measure of the conversion between the electrical and mechanical energy in the electromechanical resonator:

(38) $\begin{array}{cc}{k}_t^2=\frac{{\pi }^2}{8}\frac{{f_p}^2-{f_s}^2}{{f_s}^2}& \left(5\right)\end{array}$
where f.sub.s and f.sub.p are the series and parallel resonant frequencies of the resonator, respectively.

(39) Another useful parameter is the piezoelectric coupling constant, K.sup.2, which identifies the maximum k.sub.t.sup.2 that can be achieved for any mode of vibration through optimal excitation. K.sup.2 can be evaluated using (5) with f.sub.s and f.sub.p replaced by f.sub.o and f.sub.m, as in (6), where f.sub.o and f.sub.m are the non-metallized and the metallized resonant frequencies of the resonator, respectively. Therefore, K.sup.2 can be readily computed using the dispersion characteristics of the resonator.

(40) $\begin{array}{cc}{K}^2=\frac{{\pi }^2}{8}\frac{{f_o}^2-{f_m}^2}{{f_m}^2}& \left(6\right)\end{array}$

(41) In this work, FEM simulations were performed in COMSOL Multiphysics to study the piezoelectric coupling constant of the COM in a suspended AlN plate. Both the resonant frequencies and the K.sup.2 values were computed for a 1 μm thick AlN while varying its lateral dimension W (i.e. the wave number k.sub.x) (FIG. 2). As evident, a maximum K.sup.2-value of ˜1.2% is achieved when the AlN thickness, t.sub.AlN, is approximately equal to W. Note that this K.sup.2-value is the highest ever predicted for any overtone mode excited in an AlN plate. Note also that the presence of the metal electrode, which is not included in this analysis, but it is required in an actual COR implementation, further increases the maximum achievable k.sub.t.sup.2 as explained in the following sections.

(42) Generally, AlN resonators rely on one or more piezoelectric coefficients such as, e.sub.31, e.sub.33, and e.sub.15, to transduce a mechanical mode of vibration. Since the COM is characterized by a 2D displacement vector with a shear vibration component along the lateral direction () and a longitudinal vibration component along the vertical () direction of the plate, the coherent combination of the e.sub.15 and e.sub.33 piezoelectric coefficients of AlN is exploited for its transduction.

(43) The analytical model proposed by Berlincourt et. al. was considered to investigate the effect of different excitation schemes on the k.sub.t.sup.2-values of a device with a volume Ω:

(44) $\begin{array}{cc}{k}_t^2=\frac{{\pi }^2}{8}\frac{U_{\mathrm{coupling}}^2}{U_m{U}_e}& \left(7\right)\\ \mathrm{where}& \\ U_m=\frac{1}{2}{\int }_{\Omega }{T}_i{S}_{\mathrm{ij}}^E{T}_{j}d\Omega & \left(8\right)\\ U_{\mathrm{coupling}}=\frac{1}{4}{\int }_{\Omega }\left(T_i{e}_{\mathrm{ni}}{E}_n+E_n{e}_{\mathrm{ni}}{T}_i\right)d\Omega & \left(9\right)\\ U_e=\frac{1}{2}{\int }_{\Omega }{E}_m{\mathrm{.Math.}}_{\mathrm{mn}}^T{E}_{n}d\Omega & \left(10\right)\end{array}$

(45) Here U.sub.m is the mechanical energy, U.sub.coupling is the energy associated with the coupled electrical mechanical domain (mutual energy) and U.sub.e is the electrical energy carried over the resonator volume Ω. m, n=1, 2, 3 and i, j=1, 2, 3, 4, 5, 6.

(46) E.sub.m is the component of the electric field vector. T.sub.i is the component of the stress. e.sub.mn, ε.sup.T.sub.mn and s.sup.E.sub.ij are the piezoelectric coefficient, the dielectric constant and the elastic compliance, respectively.

(47) Equations (7)-(10) provide further insights for the design and the optimization of the excitation scheme employed to transduce vibration in the structure. In particular, (9) indicates that the higher is the integral of the product of electric and stress fields in space the higher is the k.sub.t.sup.2. Therefore, the design of an excitation scheme providing an electric field distribution matching the stress field of the COM in the structure is critical to maximize the coupling.

(48) CORs can use two main kinds of excitation schemes: the Thickness-Field-Excited (TFE) configuration, using both top and bottom IDT electrode, and the Lateral-Field-Excited (LFE) configuration, using only a top IDT (or only a bottom IDT) electrode. The LFE configuration employs metal fingers with alternating electrical polarities on the top (or the bottom) of the resonator, as shown in FIG. 3A. The typical TFE configuration (referred as TFE-1) employs metal fingers with inversed electrical polarity between top and bottom, as shown in FIG. 3B. An unconventional TFE configuration (referred as TFE-2) is also proposed and examined here, which employs metal fingers with the same electrical polarities on the top and the bottom of the resonator, as shown in FIG. 3C.

(49) The choice of the excitation scheme affects the electric field distributions in the piezoelectric film and therefore the value of k.sub.t.sup.2 for a given mode of vibration (i.e. given stress field distribution). The electric field distributions for the 3 different excitation schemes were simulated by FEM and reported in FIGS. 3A-3C while the simulated stress field distribution of a COM, neglecting the IDT mass loading effect, is plotted in FIG. 3D. The simulation results indicate that the conventional symmetric electric field distribution provided by the TFE-1 scheme is not suitable for the transduction of the COM as it results in a mutual energy, (U.sub.coupling˜0. On the other hand, the asymmetrical electric field distribution provided by the LFE and TFE-2 configurations match the asymmetrical stress field of the COM resulting in a non-zero mutual energy value. In particular, the unconventional TFE-2 excitation scheme best matches the stress field of the COM enabling the efficient piezoelectric excitation of even-order overtones (i.e. A2, A4 etc.) which results in a high electromechanical coupling coefficient k.sub.t.sup.2 for the COM.

(50) The effect of the electrode mass loading on the stress field distribution was also analyzed by FEM. The simulated electric field component, E.sub.z, and stress field components, S.sub.x and S.sub.z, in the center of a periodic unit cell are plotted in FIGS. 4A-4E. The simulation results clearly show that the mechanical loading of the metal electrodes modifies the stress field distribution of the COM improving further the matching with the electric field distribution provided by the TFE-2 scheme (particularly in the middle region of the material stack). As a result, significantly higher k.sub.t.sup.2-values can be achieved in an actual COR implementation employing metal IDT electrodes.

(51) The choices of the electrode materials and thicknesses are also important factors affecting the device k.sub.t.sup.2. Ruthenium (Ru), molybdenum (Mo), and tungsten (W) are widely adopted electrode materials for BAW resonators as they provide good crystal interfaces for the growth of highly c-oriented AlN with excellent piezoelectric properties. However, the use of these metals significantly decreases the device resonant frequency due to their high density and stiffness. High quality AlN thin-films)(FWHM=1.05°, as good as the one obtained on top of Ru, Mo or W, can be attained also on top of lighter metals, such as aluminum (A1), representing a valid alternative for the implementation of resonators operating above 6 GHz. For this reason, A1 was chosen as electrode material for the analysis and the experiments presented in this work.

(52) Different thicknesses of the A1 electrodes were simulated to find the optimal value that maximizes the k.sub.t.sup.2-values for the TFE-2 configuration. As shown in FIG. 5A, a maximum k.sub.t.sup.2-value of 1.9% was found for an optimal ratio of AlN to A1 of 1000 nm to 110 nm for the TFE-2 CORs. This represents a more than 60% increase in k.sub.t.sup.2 compared to the electrode-less case (FIG. 2). As a comparison, resonators relying on fundamental modes of vibrations (such as conventional BAW resonators) typically show a 14% increment in V associated to the presence of metal electrodes. A similar study was performed for the LFE configuration (top A1 IDT electrode only) showing a maximum k.sub.t.sup.2-value of 0.8% for optimized vertical and lateral dimensions (IDT pitch W and metallic coverage a) of the resonator stack, as shown in FIG. 5B.

(53) B. Quality Factor Q

(54) The quality factor Q of a piezoelectric resonators is defined as the ratio between stored and dissipated energies. The total quality factor Q.sub.total can be generally expressed as in (11):

(55) $\begin{array}{cc}\frac{1}{Q_{total}}=\frac{1}{Q_{\mathrm{anchor}}}+\frac{1}{Q_{\mathrm{interface}}}+\frac{1}{Q_{\mathrm{material}}}+\frac{1}{Q_{\mathrm{electrical}}}+\frac{1}{Q_{\mathrm{dielectric}}}+\frac{1}{Q_{\mathrm{intrinsic}}}& \left(11\right)\end{array}$

(56) The anchor loss, I/Q.sub.anchor, describes the acoustic energy lost through the anchor to the substrate. The typical frame designs for energy confinement employed for lower frequency resonators cannot be readily applied to higher frequency devices due to their reduced dimensions. As can be seen from the modeshape in FIG. 1A, COM has a strong shear vibration which is characterized by high Q. Therefore, the energy is naturally well confined inside the resonator body. The interface loss, I/Q.sub.interface, is caused by the plane stress jump at the interface between the piezoelectric layer and the metal electrode. As shown in the COR modeshape simulation in FIG. 1A, the IDT electrodes are placed in the center of each periodic unit where the displacement is minimum, which minimizes the scattering of the acoustic wave at the electrode-AlN interface and the associated damping. The material loss, I/Q.sub.material, is related to the presence of defects in the crystal. Since high operating frequency is achieved in CORs while using an AlN layer thicker than the one employed by more conventional technologies operating in the same frequency range, the material loss is mitigated in CORs. The electrical loss, I/Q.sub.electrical, is determined by the ohmic losses of the metal electrodes and the electrical routing. The use of overtones in a relatively thick AlN layer allows the use of sufficiently thick metal electrodes with minimal electrical loading effect. The dielectric loss, I/Q.sub.dielectric, is determined by the inherent dissipations of electromagnetic energy in the dielectric material which are reduced in the sufficiently thick AlN films employed in CORs. In fact, it was experimentally shown that the dielectric loss increases rapidly only when the AlN film is scaled below 250 nm. Other dissipative mechanisms intrinsic to the resonating material, I/Q.sub.intrinsic, include thermoelastic, phonon-electron, and phonon-phonon interactions. The Akheiser theory predicts that the f-Q.sub.intrinsic product of AlN increases linearly with frequency for frequencies larger than ˜1 GHz, enabling the achievement of high Q.sub.intrinsic in cm-mmWave AlN resonators.

(57) In summary, it can be qualitatively concluded that most of the energy loss mechanisms are mitigated in CORs due to the use of overtones in a relatively thick material stack.

III. Contiguous COR Filters for 5G mmWave Spectrum

(58) Frequency scaling of existing micro-resonator technologies up to the mmWave range has not been successful to date due to the severe performance degradation associated with the required aggressive scaling in both the vertical and lateral dimensions of the resonator. This fundamental challenge, that has so far prevented the implementation of performing mmWave micro-acoustic filters, is overcome by the COR technology. In fact, thanks to the use of a combined overtone mode of vibration the operating frequency of CORs can be scaled up to mmWave frequencies while maintaining relaxed lithographic requirements (minimum feature>100 nm) and relatively thick AlN films (>220 nm) (Table I and Table II).

(59) TABLE-US-00001 TABLE I DIMENSIONS FOR LFE CORS OF DIFFERENT FREQUENCIES Freq W t.sub.AlN t.sub.Al [GHz] [nm] [nm] [nm] 24 431 375 45 25 414 360 43 26 398 346 42 27 383 333 40 28 370 321 39 29 357 310 37 30 345 300 36 31 334 290 35 32 323 281 34 33 314 273 33 34 304 265 32 35 296 257 31 36 288 250 30 37 280 243 29 38 272 237 28 39 265 231 28 40 259 225 27

(60) TABLE-US-00002 TABLE II DIMENSIONS FOR TFE-2 CORS OF DIFFERENT FREQUENCIES Freq W t.sub.AlN t.sub.Al [GHz] [nm] [nm] [nm] 24 422 383 42 25 405 368 40 26 389 354 39 27 375 341 37 28 361 329 36 29 349 317 35 30 337 307 34 31 326 297 33 32 316 288 32 33 307 279 31 34 298 271 30 35 289 263 29 36 281 256 28 37 274 249 27 38 266 242 27 39 259 236 26 40 253 230 25

(61) The FEM simulated operating frequencies of a COR and a BAW resonator, for different thicknesses of the AlN film, t.sub.AlN, are shown in FIG. 10. The operating frequency of the COR is 73% higher than the one of the BAW resonators. As evident, the COR technology can deliver resonators operating up to 49 GHz while maintaining an AlN thickness>200 nm (i.e. preserving good film quality of the piezoelectric film with <10% degradation compared to thicker films used in lower frequency devices) while the frequency of more conventional BAW resonators would be limited to 28 GHz under the same film thickness constraint.

(62) Another challenge associated with aggressive scaling of the film thickness is the highly increased sensitivity of the device operating frequency to process-related film thickness variations. This challenge is partially mitigated in CORs thanks to the transduction of a 2D mode of vibration with both lateral and vertical displacement components. As evident from the FEM simulation results in FIG. 11, CORs show a lower frequency sensitivity to thickness variation compared to more conventional BAW devices.

(63) The special 2-D modal characteristics of CORs also enable lithographic control of the device resonance frequency which is not possible in more conventional BAW resonators relying on a thickness extensional mode of vibration. In particular, the FEM simulations herein show that a lithographic frequency tuning of ˜4%, with <20% reduction in k.sub.t.sup.2, is possible for both TFE-2 and LFE CORs operating at ˜24 GHz (FIG. 12-13).

(64) The capability of lithographical tuning of the operating frequency of CORs offers advantages for the implementation of filter architectures. In fact, the synthesis of a micro-acoustic filter requires the coupling (electrically or mechanically) of multiple resonators operating at different frequencies. The COR technology is capable of delivering such multi-frequency resonators on the same chip with reduced fabrication complexity compared to more conventional resonator technologies for which lithographic frequency tuning is not possible requiring additional process steps (i.e. additional lithography masks and selective material deposition) to adjust the frequencies of individual resonators used to synthesize the filter. The bandwidth (BW) of a micro-acoustic ladder filter is directly proportional to the k.sub.t.sup.2-values of the resonators employed to synthesize the filter, while the filter Insertion Loss (I.L.) is inversely proportional to the product of the resonators k.sub.t.sup.2 and Q, as shown in (12-13).

(65) $\begin{array}{cc}\mathrm{BW}\propto \frac{3}{{\pi }^2}{k}_t^2& \left(12\right)\\ I.L.\propto \frac{1}{k_t^{2}Q}& \left(13\right)\end{array}$

(66) As discussed herein, AlN CORs provide k.sub.t.sup.2-values as high as 0.8%, for the LFE configuration, and 1.9% for the TFE-2 configuration, which directly translate into the capability of synthesizing mmWave filters with 100s MHz bandwidth, as reported in Table III and Table IV.

(67) TABLE-US-00003 TABLE III BANDWIDTH (BW) ACHIEVABLE BY 3.sup.RD ORDER TFE- 2 COR LADDER FILTERS IN THE MMWAVE SPECTRUM Freq BW [GHz] [MHz] 24 355 25 370 26 385 27 399 28 414 29 429 30 444 31 459 32 473 33 488 34 503 35 518 36 533 37 547 38 562 39 577 40 592

(68) TABLE-US-00004 TABLE IV BANDWIDTH (BW) ACHIEVABLE BY 3.sup.RD ORDER LFE COR LADDER FILTERS IN THE MMWAVE SPECTRUM Freq BW [GHz] [MHz] 24 150 25 156 26 163 27 169 28 175 29 181 30 188 31 194 32 200 33 206 34 213 35 219 36 225 37 231 38 238 39 244 40 250

(69) The performance level achievable by mmWave COR filters is here investigated by FEM simulations of contiguous 3.sup.rd order ladder filters synthesized by lithographically defined TFE-2 (FIG. 14) and LFE (FIG. 15) CORs operating at ˜24 GHz. The simulation results show that 5 lithographically defined multi-frequency TFE-2 COR filters achieve an individual filter BW>300 MHz capable of supporting an aggregated BW>1.6 GHz. Similarly, 7 lithographically defined multi-frequency LFE COR filters achieve an individual filter BW>150 MHz capable of supporting an aggregated BW>1 GHz. It is also worth noting that, assuming a resonator Q.sub.l of 750, an I.L.<1.2 dB is achieved with the TFE-2 configuration and an I.L.<2.0 is achieved with the LFE configuration. Even with a degraded Q of 500, an I.L.<1.5 dB is achieved for the TFE-2 configuration and an I.L.<2.5 is achieved with the LFE configuration, as shown in FIG. 16.

(70) Note that this simulated performance level of the mmWave AlN LFE CORs is suitable for the synthesis of miniaturized and low-cost (simple 2-mask fabrication process) filters meeting the frequency, loss, bandwidth, roll-off and out-of-band rejection requirements of the 24 GHz 5G bands that have been recently licensed.

IV. Fabrication and Experimental Results

(71) As a technological proof-of-concept LFE CORs were designed, fabricated and tested. Although the LFE configuration yields a lower k.sub.t.sup.2 than the TFE-2 one, it is characterized by the minimal fabrication complexity. The devices were fabricated using a simple 2-mask fabrication process as described in FIG. 6. The process started with the sputter deposition of a highly textured (1.1° FWHM) 1 μm thick AlN film directly on top of a polished high-resistivity silicon wafer. Then an aluminum (Al) layer was sputter deposited and patterned (using an i-line stepper) on top of the AlN to form the device top IDT electrode with a pitch of 1.15 μm. The AlN film was then dry etched in inductively coupled plasma (ICP) to define the shape of the suspended AlN plate. Finally, the devices were released from the substrate by XeF.sub.2 dry isotropic etching of silicon. A scanned electron microscope (SEM) picture of a fabricated COR is shown in FIG. 7. As evident in FIG. 7(c), a sloped sidewall (SW) angle around 60° was observed in the cross-sectional view due to the photoresist etching mask employed in the process.

(72) The performance of the fabricated prototypes were characterized using a Microwave Network Analyzer (Keysight N5221A PNA) in air and at room temperature. The measured admittance is shown in FIG. 8A. A strong resonance, corresponding to the transduction of the combined overtone mode of vibration, was measured near the frequency predicted by the FEM simulations with an error of 3%. The measured COR showed a loaded quality factor Q.sub.l of ˜750 and a mechanical quality factor Q.sub.m of ˜1100 (corresponding to f.Math.Q.sub.m˜1×10.sup.13). Such Q-values are the highest ever reported for piezoelectric micro-acoustic resonators operating >6 GHz. In addition, the measured f.Math.Q.sub.m product ˜1×10.sup.13 is the highest ever demonstrated for piezoelectric micro-acoustic resonators made out of a single AlN thin film employing a metal coverage of at least 50%, rivaling polycrystalline silicon capacitive resonators while exhibiting several orders of magnitude lower motional impedance.

(73) A summary of Q.sub.l-values reported in literature for piezoelectric resonators operating above 6 GHz is shown in FIG. 9. It is interesting to note that all demonstrated Q.sub.l-values are nearly constant and below 400 in the 6-30 GHz range, even though AlN devices are supposed to show an increased f−Q.sub.intrinsic product in this frequency range. This trend demonstrates how the increased loss associated with aggressive scaling of the piezoelectric film and the electrodes in conventional resonators poses a hard limit to the maximum achievable Q.sub.l-values. The COR technology described herein overcomes this fundamental limitation by exploiting the efficient transduction of a combined overtone mode of vibration in a relatively thick material stack, which directly translates into an improvement (>125%) in Q factor compared to the prior art piezoelectric resonator technologies.

(74) An electromechanical-coupling coefficient, k.sub.t.sup.2˜0.3% was also extracted from the measured electrical response of the devices. Note that this experimental value is lower than the highest one predicted by FEM simulation for the optimal design (t.sub.AlN=1 μm, t.sub.Al=120 nm, W=1.15 μm, α=35%, SW angle=90°, k.sub.t.sup.2˜0.78%). This is due to the fact that only a sub-optimal design (t.sub.AlN=1 μm, t.sub.Al=85 nm, W=1.15 μm, α=57%, SW angle=60°, k.sub.t.sup.2˜0.35%) could be experimentally demonstrated because of the limited fabrication capabilities available. State-of-the-art lithographic techniques in the integrated circuit (IC) industry would be sufficient to implement IDTs with pitches (W) and metallic coverage (a) suitable for optimal operation in the mm-Wave range (see Tables I and II). Furthermore, the use of a SiO.sub.2 hard mask for the AlN etch would be sufficient to improve the SW angle to >80°. Nevertheless, the FEM simulation of the sub-optimal design (actual fabricated dimensions) matches well with the experimental results (FEM-simulated k.sub.t.sup.2˜0.35%), proving the validity of FEM model. Note also demonstrated COR prototypes were properly sized to have a static capacitance, C.sub.0˜320 fF which corresponds to a termination impedance of 56Ω at the device operating frequency, making the devices suitable for direct interface with 50Ω radio frequency systems without using off-chip impedance matching networks.

(75) Additionally, a 23.2 GHz COR was fabricated using similar fabrication steps as previously described, except for employing electron beam lithography tool for patterning of the top electrode. A scanned electron micrograph of a fabricated 23.2 GHz COR is shown in FIG. 17. The 23.2 GHz COR was designed to have following design specifications: t.sub.AlN=360 nm, t.sub.Al=43 nm, W=414 nm, α=50%. Device performance was characterized using a Microwave Network Analyzer (KT-N5224A-200/W7X PNA) in air and at room temperature. The measured admittance is shown in FIG. 18. Resonance was observed near the frequency predicted by the FEA with an error of 7.5%. Measured mm wave COR showed a loaded quality factor, Q.sub.l, of ˜225 and a mechanical quality factor, Q.sub.m of ˜254 (corresponding to f.Math.Q.sub.m˜0.6×10.sup.13). Demonstrated Q-values are expected to improve with an optimized anchor design and improved fabrication process. An improved electromechanical-coupling coefficient of 0.46% was demonstrated, however, is still lower than the one predicted from the FEA model (0.73%). This discrepancy is attributed to sub-optimal design parameters (t.sub.AlN=383 nm, t.sub.Al=40 nm, W=415 nm, SW angle=70°) and can be mitigated with an unoptimized fabrication process.

V. Conclusions

(76) The technology described herein provides a class of AlN MEMS resonators, called combined overtone resonators (CORs), that use the coherent combination of the e.sub.33 and the e.sub.15 piezoelectric coefficients of AlN for the multimodal excitation of the 2.sup.nd and 3.sup.rd order asymmetrical Lamb-wave overtones (A2 and A3 respectively) resulting in the efficient transduction of a 2-dimensional (2D) combined overtone mode (COM) of vibration in the cross-section of a suspended thin-film AlN plate. A COR prototype operating at ˜9 GHz was fabricated and tested showing a high k.sub.t.sup.2˜0.3% (using a simple top-electrode-only configuration with a 2-mask process) and a Q˜1100, which is the highest ever achieved among piezoelectric resonators operating above 6 GHz. Detailed analytical and finite-element method (FEM) models were developed and used to demonstrate that a high electromechanical coupling coefficient, k.sub.t.sup.2˜1.9%, can be achieved by optimizing the device geometry and electrode configuration. The advantages of the COR technology for the implementation of a class of mmWave micro-acoustic filters is demonstrated by FEM analysis. 3.sup.rd order ladder filters, using lithographically defined CORs operating at ˜24 GHz, were designed and their performance simulated by FEM demonstrating the capability of synthesizing mmWave contiguous filters with 100s MHz bandwidths supporting aggregated bandwidths >1 GHz. The demonstrated performance level of the mmWave AlN CORs is suitable for the synthesis of miniaturized, single-chip and low-cost contiguous filters meeting the frequency, loss, bandwidth, roll-off and out-of-band rejection requirements of 24 GHz 5G bands. Therefore, the COR technology described herein addresses the challenges that have so far prevented the implementation of high performance mmWave micro-acoustic filters for ubiquitous and pervasive mobile internet connectivity enabled by highly miniaturization and integrated 5G systems.

(77) As used herein, “consisting essentially of” allows the inclusion of materials or steps that do not materially affect the basic and novel characteristics of the claim. Any recitation herein of the term “comprising,” particularly in a description of components of a composition or in a description of elements of a device, can be exchanged with “consisting essentially of” or “consisting of.”

(78) The present technology has been described in conjunction with certain preferred embodiments and aspects. It is to be understood that the technology is not limited to the exact details of construction, operation, exact materials or embodiments or aspects shown and described, and that various modifications, substitution of equivalents, alterations to the compositions, and other changes to the embodiments and aspects disclosed herein will be apparent to one of skill in the art.