Receiving spread spectrum signals

Abstract

A receiver for spread spectrum signals comprising a first part for preprocessing and digitizing a received signal, and a second part for tracking the digitized signal comprising a carrier loop and a code loop. The code loop comprises a generator for a reference receiver signal for correlation with the received signal and the code loop is configured to modify the reference signal to shape a correlation function between the received signal and the reference receiver signal. The first part is adapted to multiply the received spread spectrum signal with a first analog spectral offsetting signal provided for down-converting the received signal to an intermediate frequency and a sub-carrier frequency, selected from a set of sub-carrier frequencies, such that the received signal is down-converted and spectrally offset in the analog domain during a time interval covering at least one chip of a spreading code of the received signal.

Claims

1. A receiver for spread spectrum signals comprising a first part configured to preprocess and digitize a received spread spectrum signal, and a second part configured to track the digitized received spread spectrum signal and comprising a carrier loop and a code loop, wherein the code loop comprises a generator for a reference receiver signal for correlation with the received spread spectrum signal and the code loop is configured to modify the reference receiver signal in order to shape a correlation function between the received spread spectrum signal and the reference receiver signal, wherein one of the first part is adapted for multiplying the received spread spectrum signal with a first analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF0 and a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the received spread spectrum signal is down-converted and spectrally offset in the analog domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, or the first part is adapted for multiplying the received spread spectrum signal with a second analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF0 in the analog domain, for digitizing the down-converted received spread-spectrum signal, and for multiplying the digitized and down-converted received spread spectrum signal with a digital spectral offsetting signal provided for spectrally offsetting the digitized and down-converted received spread spectrum signal in the digital domain to a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the digitized and down-converted received spread spectrum signal is spectrally offset in the digital domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal.

2. The receiver of claim 1, wherein several spectral offsettings in the analog domain or in the digital domain are performed with different sub-carrier frequencies, f.sub.m, and/or the time interval covering L.sub.m chips varies from cycle to cycle of a frequency offsetting cycle C covering M sub-carrier frequencies {f.sub.m}.sub.1mM,C.

3. The receiver of claim 1, wherein the first analog spectral offsetting signal is defined by e.sup.j(2(f.sup.RF.sup.+IF.sup.0.sup.+f.sup.m.sup.)(t.sup.est,prec.sup.)+.sup.m.sup.), and the second analog spectral offsetting signal is defined by e.sup.j(2(f.sup.RF.sup.+IF.sup.0.sup.)(t.sup.est,prec.sup.)) and the digital spectral offsetting signal is defined by e.sup.j(2(f.sup.m.sup.)(t.sup.est,prec.sup.)+.sup.m.sup.) with t being a time variable, f.sub.m being a sub-carrier frequency selected from the set of M sub-carrier frequencies and applied during a time interval covering L.sub.m chips, .sub.est,prec being a time offset, which is used for synchronization and depends on the actual propagation delay of the transmitted signal up to the receiver and is estimated in a .sub.est,prec generator from at least one of the past outputs of the code loop and/or the carrier loop of the receiver, and/or is estimated by an external estimator of the propagation delay to the receiver and processed by the .sub.est,prec generator, and m being a phase which ensures that the argument of the exponential term is continuous at a transition between two consecutive time intervals corresponding to two sub-carrier frequencies f.sub.m and f.sub.m+1.

4. The receiver of claim 1, wherein the first part comprises a local oscillator, an analog digital converter and either a frequency generator configured for synthesizing frequency references for the local oscillator for down-converting the received spread spectrum signal to the intermediate frequency IF.sub.0 and the sub-carrier frequency f.sub.m and for the analog digital converter for digitizing the down-converted received spread spectrum signal, or a frequency generator configured for synthesizing frequency references for the local oscillator for down-converting the received spread spectrum signal to the intermediate frequency IF.sub.0 and for the analog digital converter for digitizing the down-converted received spread spectrum signal and an additional digital local oscillator to spectrally offset the down-converted digital spread spectrum signal to the sub-carrier frequency f.sub.m.

5. The receiver of claim 1, wherein the first part comprises a pre-processing unit for pre-processing the received spread spectrum signal before digitizing, wherein the pre-processing unit comprises filtering and an automatic gain control of the received spread spectrum signal.

6. The receiver of claim 1 being configured to receive as spread spectrum signal a signal with any pulse shape type such that the combination of the reference receiver signal multiplied with the offsetting signal can be replaced with a Binary Offset carrier (BOC) (K,N) reference receiver signal if the set of M sub-carrier frequencies contains only 2 frequencies +f.sub.K and f.sub.K and the duration of the time intervals when +f.sub.k is applied is identical to the duration of the time intervals when f.sub.k is applied.

7. The receiver of claim 6, wherein the receiver is configured to receive as spread spectrum signal a Binary-Phase-Shift-KeyingBPSK(N)modulated spread spectrum signal.

8. A device for positioning using spread spectrum signals transmitted by a Global Navigation Satellite System, the device comprising a receiver of claim 1 for receiving the spread spectrum signals.

9. A method for receiving spread spectrum signals comprising the following steps: preprocessing and digitizing a received spread spectrum signal, tracking the digitized received spread spectrum signal by means of a carrier loop and a code loop, wherein the tracking comprises generating a reference receiver signal for correlation with the received spread spectrum signal in the code loop and shaping a correlation function between the received spread spectrum signal and the reference receiver signal by modifying the reference receiver signal in the code loop, wherein one of the received spread spectrum signal is multiplied with a first analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF.sub.0 and a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the received spread spectrum signal is down-converted and spectrally offset in the analog domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, or the received spread spectrum signal is multiplied with a second analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF.sub.0 in the analog domain, for digitizing the down-converted received spread-spectrum signal, and for multiplying the digitized and down-converted received spread spectrum signal with a digital spectral offsetting signal provided for spectrally offsetting the digitized and down-converted received spread spectrum signal in the digital domain to a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the digitized and down-converted received spread spectrum signal is spectrally offset in the digital domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal; wherein the first analog spectral offsetting signal is defined by e.sup.j(2(f.sup.RF.sup.+IF.sup.0.sup.+f.sup.m.sup.)(t.sup.est,prec.sup.)+.sup.m.sup.), and the second analog spectral offsetting signal is defined by e.sup.j(2(f.sup.RF.sup.+IF.sup.0.sup.)(t.sup.est,prec.sup.)) and the digital spectral offsetting signal is defined by e.sup.j(2(f.sup.m.sup.)(t.sup.est,prec.sup.)+.sup.m.sup.) with t being a time variable, f.sub.m being a sub-carrier frequency selected from the set of M sub-carrier frequencies and applied during a time interval covering L.sub.m chips, .sub.est,prec being a time offset, which is used for synchronization and is a function of the actual propagation delay of the transmitted signal up to the receiver and is estimated in a .sub.est,prec generator from at least one of the past outputs of the code loop and/or carrier loop of the receiver, and/or is estimated by an external estimator of the propagation delay to the receiver and processed by the .sub.est,prec generator, and m being a phase which ensures that the argument of the exponential term is continuous at a transition between two consecutive sub-carrier frequencies f.sub.m and f.sub.m+1.

10. The method of claim 9, wherein several spectral offsettings in the analog domain or in the digital domain are performed with different sub-carrier frequencies, f.sub.m, and/or the time interval covering L.sub.m chips varies from cycle to cycle of a frequency offsetting cycle C covering M sub-carrier frequencies {f.sub.m}.sub.1mM,C.

11. The method of claim 9, wherein the step of preprocessing comprises filtering and an automatic gain control of the received spread spectrum signal.

12. A method for receiving spread spectrum signals comprising the following steps: preprocessing and digitizing a received spread spectrum signal, tracking the digitized received spread spectrum signal by means of a carrier loop and a code loop, wherein the tracking comprises generating a reference receiver signal for correlation with the received spread spectrum signal in the code loop and shaping a correlation function between the received spread spectrum signal and the reference receiver signal by modifying the reference receiver signal in the code loop, wherein one of the received spread spectrum signal is multiplied with a first analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF.sub.0 and a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the received spread spectrum signal is down-converted and spectrally offset in the analog domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, or the received spread spectrum signal is multiplied with a second analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF.sub.0 in the analog domain, for digitizing the down-converted received spread-spectrum signal, and for multiplying the digitized and down-converted received spread spectrum signal with a digital spectral offsetting signal provided for spectrally offsetting the digitized and down-converted received spread spectrum signal in the digital domain to a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the digitized and down-converted received spread spectrum signal is spectrally offset in the digital domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, wherein the steps of down-converting and digitizing of the received spread spectrum signal comprises either the synthesizing of frequency references for down-converting the received spread spectrum signal to the intermediate frequency IF.sub.0 and the sub-carrier frequency f.sub.m and for digitizing the down-converted received spread spectrum signal, or the synthesizing of frequency references for down-converting the received spread spectrum signal to the intermediate frequency IF.sub.0 and for digitizing the down-converted received spread spectrum signal and for spectrally offsetting the down-converted digital spread spectrum signal to the sub-carrier frequency fm.

13. The method of claim 12, wherein several spectral offsettings in the analog domain or in the digital domain are performed with different sub-carrier frequencies, f.sub.m, and/or the time interval covering L.sub.m chips varies from cycle to cycle of a frequency offsetting cycle C covering M sub-carrier frequencies {f.sub.m}.sub.1mM,C.

14. The method of claim 12, wherein the step of preprocessing comprises filtering and an automatic gain control of the received spread spectrum signal.

15. A method for receiving spread spectrum signals comprising the following steps: preprocessing and digitizing a received spread spectrum signal, tracking the digitized received spread spectrum signal by means of a carrier loop and a code loop, wherein the tracking comprises generating a reference receiver signal for correlation with the received spread spectrum signal in the code loop and shaping a correlation function between the received spread spectrum signal and the reference receiver signal by modifying the reference receiver signal in the code loop, wherein one of the received spread spectrum signal is multiplied with a first analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF.sub.0 and a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the received spread spectrum signal is down-converted and spectrally offset in the analog domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, or the received spread spectrum signal is multiplied with a second analog spectral offsetting signal provided for down-converting the received spread spectrum signal to an intermediate frequency IF.sub.0 in the analog domain, for digitizing the down-converted received spread-spectrum signal, and for multiplying the digitized and down-converted received spread spectrum signal with a digital spectral offsetting signal provided for spectrally offsetting the digitized and down-converted received spread spectrum signal in the digital domain to a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the digitized and down-converted received spread spectrum signal is spectrally offset in the digital domain during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, wherein as spread spectrum signal a signal with any pulse shape type and particularly a Binary-Phase-Shift-KeyingBPSK(N)modulated spread spectrum signal such that the combination of the reference receiver signal multiplied with the offsetting signal can be replaced with a Binary Offset Carrier (BOC) (K,N) reference receiver signal if the set of M sub-carrier frequencies contains only 2 frequencies +f.sub.K and f.sub.K and the duration of the time intervals when +f.sub.k is applied is identical to the duration of the time intervals when f.sub.k is applied, is received.

16. The method of claim 15, wherein several spectral offsettings in the analog domain or in the digital domain are performed with different sub-carrier frequencies, f.sub.m, and/or the time interval covering L.sub.m chips varies from cycle to cycle of a frequency offsetting cycle C covering M sub-carrier frequencies {f.sub.m}.sub.1mM,C.

17. The method of claim 15, wherein the step of preprocessing comprises filtering and an automatic gain control of the received spread spectrum signal.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) FIG. 1 shows a first embodiment of a first part of the receiver architecture according to the invention comprising a multiplication with the sub-carrier f.sub.m, in the analog domain;

(2) FIG. 2 shows a second embodiment of a first part of the receiver architecture according to the invention comprising a multiplication with the sub-carrier f.sub.m, in the digital domain;

(3) FIG. 3 shows an embodiment of a second part of the receiver architecture according to the invention which can be applied with the embodiments of FIG. 1 or FIG. 2;

(4) FIG. 4 shows a plot of the ACF of the BPSK modulated signal, and more exactly a BPSK(1) which means that the chip rate is 1 MChip/s;

(5) FIG. 5 shows a plot of the ACF and the modulation terms applied to the BPSK signal BPSK(1);

(6) FIG. 6 shows a plot of the multiplication of the ACF and the modulation terms of the BPSK signal BPSK(1);

(7) FIG. 7 shows a plot of the code tracking accuracy of a conventional receiver architecture and of the innovative receiver architecture according to the invention, in both cases when receiving a BPSK(1) signal;

(8) FIG. 8 shows a plot of the MP envelope of a BPSK CDMA signal of a conventional receiver architecture;

(9) FIG. 9 shows a plot of the MP envelope of a BPSK(1) signal of the innovative receiver architecture according to the invention;

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

(10) In the following, functionally similar or identical elements may have the same reference numerals. Absolute values are shown below by way of example only and should not be construed as limiting the invention.

(11) In FIG. 1, an embodiment of the inventive architecture applying a multiplication of the sub-carrier frequency in the analog domain (in an analog part of the receiver front-end) is considered for illustration. The mathematical derivations and the advantages in terms of robustness against noise and multipath would be similar when applying the multiplication of the sub-carrier frequency in the digital domain (as illustrated by the embodiment of the inventive architecture shown in FIG. 2).

(12) FIG. 1 shows a first, analog part 10 of the inventive receiver architecture for receiving spread spectrum signals transmitted by a GNSS, particularly by GNSS satellites. A GNSS (spread spectrum) signal is received through an antenna 100 and then filtered through a Band Pass Filter (BPF) at RF to reduce the effects of potential out-of-band interferers and to preserve thus the Low-Noise Amplifier (LNA) or other components of the receiver front-end 102. The signal is then amplified by a Low Noise Amplifier (LNA) with a minimal degradation of the receiver noise FIG. 104 to compensate for transmission

(13) A frequency generator 112 synthetizes a frequency reference for all the front end components and allows down-converting the signal at Radio Frequency (f.sub.RF) to an Intermediate Frequency (IF.sub.0) plus a sub-carrier frequency (f.sub.m) by means of a Local Oscillator (LO) 114. The multiplication with the first analog spectral offsetting signal e.sup.j(2(f.sup.RF.sup.+IF.sup.0.sup.+f.sup.m.sup.)(t.sup.est,prec.sup.)+.sup.m.sup.) ensures the additional spectral offset, f.sub.m, of the down-converted signal and constitutes the first part and condition of the proposed invention. In the argument of the exponential expression, t is the time variable, f.sub.m is a sub-carrier frequency among a set of M sub-carriers and applied during a time interval covering L.sub.m chips, wherein L.sub.m chips comprises at least one chip of a spreading code of the received spread spectrum signal, .sub.est,prec is a time offset, used for synchronization, which is function of the actual propagation delay, , of the transmitted signal up to the receiver, and which is estimated from the past outputs of the Code and Carrier loops embedded in the receiver, or by an external estimator of the propagation delay to the receiver, and m is a phase which ensures that the argument of the exponential term is continuous at the transition between two consecutive sub-carrier phase f.sub.m and f.sub.m+1. In the exemplary embodiment, one considers that M spectral offsettings are applied successively over a period covering P chips such that

(14) $P=\underset{m=1}{\overset{M}{.Math.}}{L}_m.$
Furthermore in the same exemplary embodiment, one considers that the spreading code is periodical with L chips, and that the duration covering the M spectral offsettings covers one spreading code period such that P=L. Finally in the same exemplary embodiment, one considers the special case where each sub-carrier frequency f.sub.m is applied over a duration covering a fraction of (1/M) of the spreading code period duration, which means that M is a divider of L, and that the number of chips L.sub.m is identical for all frequencies fm. In this exemplary embodiment, it means that the time intervals L.sub.m are identical to (L/M) chips for all sub-carrier frequencies and that the set {L.sub.m}.sub.1mM contains M times the same value (L/M) for different frequency offsetting cycles. The LO output signal is multiplied with the band-filtered and amplified received GNSS signal for down-conversion to the Intermediate Frequency IF.sub.0 plus the sub-carrier frequency fm. The Intermediate Frequency IF.sub.0 can be zero; in that case the signal is down-converted directly to the sub-carrier frequency f.sub.m.

(15) The sub-carrier frequency (f.sub.m(t)) takes different values among a set of M f.sub.m values during a period covering P chips which is equal to the spreading code period, L, of the GNSS signal, and depends consequently on time (for simplification, in the following mathematical description the time dependency of f.sub.m is not mentioned any more). The down-converted GNSS signal is then pre-processed before digitizing with an Analog-to-Digital Converter (ADC) 110. This signal pre-processing is performed by a pre-processing in 108, which can include filtering and an Automatic Gain Control (AGC), where the role of the filtering is to attenuate potential interfering signals appearing within the bandwidth of the first BPF and the role of the AGC is to guarantees that the amplitude of received signal is adapted to the dynamic of an Analog-to-Digital Converter (ADC) in block 110 to minimize the quantization losses. The analog down-converted and pre-processed GNSS signal is then digitized through the ADC 110, which outputs digital IQ (In-Phase and Quadrature) samples of the GNSS signal for processing by a digital part of the inventive receiver architecture.

(16) FIG. 2 shows a mixed analog/digital implementation of the first part 11 of the inventive receiver architecture for receiving spread spectrum signals transmitted by a GNSS, where the multiplication of the sub-carrier frequency is applied in the digital domain, i.e., after down-conversion and digitizing of a received GNSS signal. In this implementation, a frequency generator 113 synthetizes a frequency reference for all the front end components and allows down-converting the signal at Radio Frequency (f.sub.RF) to an Intermediate Frequency (IF.sub.0) by means of a Local Oscillator (LO) 114. The multiplication with the second analog spectral offsetting signal e.sup.j(2(f.sup.RF.sup.+IF.sup.0.sup.)(t.sup.est,prec.sup.)) ensures the down-conversion of the signal. In the argument of the exponential expression, t is the time variable, .sub.est,prec is a time offset, used for synchronization, which is function of the actual propagation delay, , of the transmitted signal up to the receiver, and which is estimated from the past outputs of the Code and Carrier loops embedded in the receiver, or by an external estimator of the propagation delay to the receiver. The LO output signal is multiplied with the band-filtered and amplified received GNSS signal for down-conversion to the Intermediate Frequency IF.sub.0.

(17) The down-converted GNSS signal is then pre-processed before digitizing with an Analog Digital Converter (ADC) 110. This signal pre-processing is performed by a pre-processing in 108, which can include filtering and an Automatic Gain Control (AGC), where the role of the filtering is to attenuate potential interfering signals appearing within the bandwidth of the first BPF and the role of the AGC is to guarantees that the amplitude of received signal is adapted to the dynamic of an Analog-to-Digital Converter (ADC) in block 110 to minimize the quantization losses. The analog down-converted and pre-processed GNSS signal is then digitized through the ADC 110, and the digitized and down-converted received spread spectrum signal is then in block 116 multiplied with a digital spectral offsetting signal e.sup.j(2(f.sup.m.sup.)(t.sup.est,prec.sup.)+.sup.m.sup.) for spectrally offsetting the digitized and down-converted received spread spectrum signal in the digital domain to a sub-carrier frequency f.sub.m selected from a set of M sub-carrier frequencies such that the digitized and down-converted received spread spectrum signal is spectrally offset in the digital domain during a time interval covering L.sub.m chips. In the argument of the exponential expression, t is the time variable, .sub.est,prec is a time offset, used for synchronization, which is function of the actual propagation delay, , of the transmitted signal up to the receiver, and which is estimated from the past outputs of the Code and Carrier loops embedded in the receiver, or by an external estimator of the propagation delay to the receiver, and m is a phase which ensures that the argument of the exponential term is continuous at the transition between two consecutive sub-carrier phase f.sub.m and f.sub.m+1. In the exemplary embodiment, one considers that M spectral offsettings are applied successively over a period covering P chips such that

(18) 0$P=\underset{m=1}{\overset{M}{.Math.}}{L}_m.$
Furthermore in the same exemplary embodiment, one considers that the spreading code is periodical with L chips, and that the duration covering the M spectral offsettings covers one spreading code period such that P=L. Finally in the same exemplary embodiment, one considers the special case where each sub-carrier frequency f.sub.m is applied over a duration covering a fraction of (1/M) of the spreading code period duration, which means that M is a divider of L, and that the number of chips L.sub.m is identical for all frequencies fm. In this exemplary embodiment, it means that the time interval L.sub.m is identical to (L/M) chips for all sub-carrier frequencies and that the set {L.sub.m}.sub.1mM contains M times the same value (L/M) for the different frequency offsetting cycles. Block 116 finally outputs digital IQ (In-Phase and Quadrature) samples of the GNSS signal for processing by a digital part of the inventive receiver architecture.

(19) The second digital part 20 of the inventive receiver architecture is shown in FIG. 3. The digitized IQ samples are transmitted from the first part (10 for the analog option, or 11 for the digital option), to the digital part 20 of the (tracking) receiver. The digital part 20 includes a code loop and a carrier loop. Both loops comprise a correlate, integrate & dump unit 202. The code loop additionally comprises a code discriminator 204, a code loop filter 206, a code numerical controlled oscillator (NCO) 208, a code and pulse generator 210 for generating a reference receiver signal, and a multiplier 212 for multiplying the reference receiver signal by e.sup.j(2(f.sup.m.sup.+IF.sup.0.sup.)(t.sup.est,cur.sup.)+.sup.m.sup.) as will be described below in more detail. The carrier loop additionally comprises a carrier discriminator 214, a carrier loop filter 216, a carrier NCO 218, and a multiplier 200 for multiplying the input signal (digitized IQ samples) with the carrier frequency output by the carrier NCO 218.

(20) The input signal Doppler frequency and phase are compensated by the carrier loop. The synchronization between the input signal and the receiver reference signal is done via the code loop.

(21) The second part of the invention, taking place in the digital part of the tracking receiver is that the reference receiver signal is multiplied by a second digital spectral offsetting signal e.sup.j(2(f.sup.m.sup.+IF.sup.0.sup.)(t.sup.est,cur.sup.)+.sup.m.sup.) with f.sub.m being a sub-carrier frequency among a set of M sub-carrier frequencies and applied during a time interval covering L.sub.m chips, t being the time variable, .sub.est,cur being the current estimate of the propagation delay , used for synchronization, and provided by the code loop or by an external estimator of the propagation delay to the receiver. The second digital spectral offsetting signal is generated by generator 220.

(22) The digital part 20 shown in FIG. 3 further comprises a .sub.est,prec generator 222 for generating the time offset .sub.est,prec used for synchronization in the analog or digital spectral offsetting performed in the receiver front-end, using as input the current estimate .sub.est,cur of the code phase due to propagation delay and/or a current estimate .sub.est,cur of the carrier phase due to the propagation delay, and/or an (optional) external estimator 224 for the time offset .sub.est,prec used for synchronization. The .sub.est,prec generator 222 for generating the time offset .sub.est,prec may either estimate the time offset .sub.est,prec by processing the inputted current estimate .sub.est,cur of the code phase due to propagation delay and/or the inputted current estimate .sub.est,cur of the carrier phase due to the propagation delay, or receive a time offset .sub.est,prec estimated by the external estimator 224 and process this received estimation of the time offset .sub.est,prec, for example by passing through received estimation of the time offset .sub.est,prec to its output, or by data fusing of the received time offset .sub.est,prec with its own estimated time offset .sub.est,prec, i.e. the time offset .sub.est,prec estimated in the .sub.est,prec generator 222 from the inputted current estimate .sub.est,cur of the code phase due to propagation delay and/or the inputted current estimate .sub.est,cur of the carrier phase due to the propagation delay. The data fusing particularly comprises a check for plausibility of the own estimated time offset .sub.est,prec by means of the received estimated time offset .sub.est,prec.

(23) In FIG. 3, the current estimate .sub.est,cur of the carrier phase due to the propagation delay is provided by the carrier loop or by the optional external estimator 224 of the propagation delay to the receiver.

(24) At least parts of the above described receiver architecture can also be implemented as steps of a method for receiving spread spectrum signals. The method can be for example implemented as computer program, which performs the steps for processing a received spread spectrum signal as described before with regard to the inventive receiver architecture, when the program is executed by a processor.

(25) In the following receiving and processing of a spread spectrum signal according to the invention is described in detail. Again, the description is proposed when the multiplication with the sub-carrier signal e.sup.j(2f.sup.m.sup.(t.sup.est,prec.sup.)+.sup.m.sup.) is performed in the analog domain, but similar derivation and results would be obtained when the multiplication is performed in the digital domain as represented in 116 on FIG. 2. For simplification, in the following equations, it is considered that the time offset .sub.est,prec used for synchronization and provided by the code loop or by an external estimator of the propagation delay to the receiver is error free, meaning that .sub.est,prec=. An error on the synchronization would introduce a slight degradation of the expected theoretical performances.

(26) Considering again the analog receiver architecture option, the expression of the received signal before multiplication with the e.sup.j(2f.sup.m.sup.(t.sup.est,prec.sup.)+.sup.m.sup.) is given by:

(27) $s_{\mathrm{rec}}\left(t\right)=p\left(t-\right).Math.\underset{l=1}{\overset{L}{.Math.}}{a}_l\left(t-{\mathrm{lT}}_c\right)$

(28) With

(29) p(t) being the pulse shape of the transmitted and received GNSS signal (e.g. a BPSK);

(30) L being the number of code chips during the PRN code period;

(31) Tc being the chip rate of the PRN code; and

(32) al being the lth PRN code chip.

(33) t is the time variable

(34) is the actual (physical) code delay due to the propagation between the transmitter and the receiver

(35) Then, the received signal is multiplied with the sub-carrier signal e.sup.j(2f.sup.m.sup.(t.sup.est,prec.sup.)+.sup.m.sup.) used to offset spectrally the received signal during a time interval covering L.sub.m chips, the resulting offset signal expression is given by:

(36) $s_{\mathrm{rec},m}\left(t\right)=p\left(t-\right)e^{j\left(2f_m\left(t-{}_{\mathrm{est},\mathrm{prec}}\right)+{}_m\right)}.Math.\underset{l=1}{\overset{L_m}{.Math.}}{a}_l\left(t-{\mathrm{lT}}_c\right)$

(37) With .sub.est,prec is an estimate of the actual code delay, , and derived from the past outputs of the Code and/or Carrier loops embedded in the receiver, or by an external estimator of the propagation delay to the receiver.

(38) The multiplications with the sub-carriers e.sup.j(2f.sup.m.sup.(t.sup.est,prec.sup.)+.sup.m.sup.) is performed over different time intervals covering L.sub.m chips, and the resulting signal is the time-aggregate of the elementary .sub.srec,m(t) signals. It should be noted again that in the proposed exemplary embodiment, the sum of the M intervals covering each L.sub.m chips equals P which is also equal to the spreading code period L.

(39) Then the digitized signal at the output of the ADC 110 and at the input of the digital part can be expressed as following when L.sub.m expressed in chips covers a constant fraction (1/M) of the spreading code period duration:

(40) $s_{\mathrm{input}}\left(t\right)=\underset{m=1}{\overset{M}{.Math.}}p\left(t-\right)e^{j\left(2f_m\left(t-{}_{\mathrm{est},\mathrm{prec}}\right)+{}_m\right)}.Math.\underset{k=\frac{\left(m-1\right)L}{M}+1}{\overset{\frac{mL}{M}}{.Math.}}{a}_k\left(t-{\mathrm{kT}}_c\right)$

(41) With:

(42) M being the number of sub-carrier frequencies fin applied during the period L of the PRN code used for coding the received GNSS signal;

(43) m being the index of the m.sup.th sub-code period to which fin is applied;

(44) p(t) being the pulse shape of the transmitted and received GNSS signal;

(45) L being the number of code chips during the PRN code period;

(46) Tc being the chip rate of the PRN code; and

(47) a.sub.k being the k.sup.th PRN code.

(48) t is the time variable

(49) In the previous equation, for mathematical simplification, no distortion at payload or receiver level has been considered, neither multipath nor atmospheric effect.

(50) The Cross Correlation Function (CCF) between the digitized signal s.sub.input and the reference receiver signal s.sub.ref can be expressed as following:

(51) $\mathrm{CCF}\left(,{}_{\mathrm{est},\mathrm{cur}}\right)=\frac{1}{T_{\mathrm{int}}}{}_0^{T_{\mathrm{int}}}{s}_{\mathrm{input}}\left(t\right)s_{\mathrm{ref}}^{*}\left(-{}_{\mathrm{est},\mathrm{cur}}\right)\mathrm{dt}$

(52) Herein Tint represents the coherent integration time used to generate the CCF and is equal to a duration covering a multiple of P=L chips, namely T.sub.int=KPT.sub.c which is also equal to KLT.sub.c chips in the proposed exemplary embodiment, and where K is an integer corresponding to the number of frequency offsetting cycles covered by the coherent integration Tint. In the following exemplary embodiment K is set to 1, but the mathematical developments would yield the same expression when integrating coherently over more than 1 frequency offsetting cycle.

(53) Note that if no distortions on the signal are considered, the CCF and Auto Correlation Function (ACF) are equivalent. Therefore both ACF and CCF will be used equivalently in the following description.

(54) Furthermore, .sub.est,cur represents the current estimation of the code delay due to the propagation between the transmitter and the receiver. It must be noted that .sub.est,cur has the same meaning as in a conventional receiver where the reference signal s.sub.ref which is also called replica, is shifted according to .sub.est,cur.

(55) The reference receiver signal s.sub.ref, including the shift .sub.est,cur applied during correlation, is equal to:

(56) $s_{\mathrm{ref}}\left(t\right)=\underset{m=1}{\overset{M}{.Math.}}p\left(t-{}_{\mathrm{est},\mathrm{cur}}\right)e^{j\left(2f_m\left(t-{}_{\mathrm{est},\mathrm{cur}}\right)+{}_m\right)}.Math.\underset{k=\frac{\left(m-1\right)L}{M}+1}{\overset{\frac{mL}{M}}{.Math.}}{a}_k\left(t-{\mathrm{kT}}_c\right)$

(57) The CCF can be expressed easily by:

(58) $\mathrm{CCF}\left(\right)=\left(\frac{1}{M}\underset{m=1}{\overset{M}{.Math.}}{e}^{j2f_m\left({}^{*}\right)}\right).Math.{\mathrm{CCF}}_p\left(\right)$

(59) With:

(60) CCF.sub.p() being the CCF of the pulse shape p, like for instance a BSPK.

(61) In the case of a BPSK, CCF.sub.BPSK() is equal to:

(62) ${\mathrm{CCF}}_{\mathrm{BPSK}}\left(\right)=\{\begin{array}{c}1-.Math..Math.,.Math..Math.T_c\\ 0,\mathrm{else}\end{array}$

(63) Where represents the tracking error for the code delay:
=.sub.est,cur

(64) Furthermore, *=.sub.est,prec.sub.est,cur. If one considers that .sub.est,prec is evaluated with the past information provided by the code and carrier loops or by an external estimator of the propagation delay to the receiver such like an inertial, a dead reckoning or an odometer system, all information eventually fed into a Kalman Filter, then .sub.est,prec is close or identical to t, leading to *=. In that case, the CCF can be expressed easily by:

(65) ${\mathrm{CCF}}_{\mathrm{BOC}}\left(\right)=\left(\frac{1}{M}\underset{m=1}{\overset{M}{.Math.}}{e}^{j2f_m\left(\right)}\right).Math.{\mathrm{CCF}}_p\left(\right)$

(66) Which is identical to the CCF of a signal using a BOC pulse shape, in case the pulse shape p is a BPSK.

(67) As an example, the ACF of a received BPSK GNSS signal BPSK(1) is shown in FIG. 4.

(68) By using the inventive receiver architecture, the CCF is the product of the CCF of the pulse shape CCFp which serves as envelop of the CCF and the modulation term:

(69) $\left(\frac{1}{M}\underset{m=1}{\overset{M}{.Math.}}{e}^{j2f_m\left({}^{*}\right)}\right).$

(70) This term is distortion independent; the shaping is the same for an ACF or for a CCF considering the signal distortions. The technique can also be applied to a distorted signal.

(71) Another embodiment of the proposed invention considers the extreme case when each sub-carrier frequency is applied during a single chip and therefore L.sub.m equals one chip for all sub-carrier frequencies fm. M sub-carrier frequencies will therefore cover P=M chips, which does not have to be equal to the number of chips within an entire spreading code period, L, but can be smaller or larger than L. For example, only two frequencies (P=M=2), f1 and f2 can be considered and can be alternated every chip (L.sub.m=1). Herein, the digitized signal at the output of the ADC 110 and at the input of the digital part can be expressed as following when Lm covers only one chip:

(72) 0$s_{\mathrm{input}}\left(t\right)=\underset{m=1}{\overset{M}{.Math.}}p\left(t-\right)e^{j\left(2f_m\left(t-{}_{\mathrm{est},\mathrm{prec}}\right)+{}_m\right)}.Math.a_m\left(t-{\mathrm{mT}}_c\right)$

(73) With:

(74) M being the number of sub-carrier frequencies f.sub.m;

(75) m being the index of the m.sup.th code chip to which f.sub.m is applied;

(76) p(t) being the pulse shape of the transmitted and received GNSS signal;

(77) Tc being the chip rate of the PRN code; and

(78) am being the m.sup.th PRN code.

(79) t is the time variable

(80) Here again, in the previous equation, for mathematical simplification, no distortion at payload or receiver level has been considered, neither multipath nor atmospheric effect.

(81) The Cross Correlation Function (CCF) between the digitized signal (s.sub.input and the reference receiver signal s.sub.ref can be expressed as following:

(82) $\mathrm{CCF}\left(,{}_{\mathrm{est},\mathrm{cur}}\right)=\frac{1}{T_{\mathrm{int}}}{}_0^{T_{\mathrm{int}}}{s}_{\mathrm{input}}\left(t\right)s_{\mathrm{ref}}^{*}\left(t-{}_{\mathrm{est},\mathrm{cur}}\right)\mathrm{dt}$

(83) Herein Tint represents the coherent integration time used to generate CCF and is equal to a duration covering several elementary durations of P=M=2 chips, namely T.sub.int=KMT.sub.c=KPT.sub.c and where K is an integer corresponding to the number of frequency offsetting cycles of M chips covered by the coherent integration T.sub.int. In the following exemplary embodiment K is set to 1, but the mathematical developments would yield the same expression when integrating coherently over more than 1 frequency offsetting cycle. In practice, K should be set to a larger value than 1 because for K=1 the coherent integration time would probably be not long enough to ensure good performances.

(84) Note that if no distortions on the signal are considered, the CCF and Auto Correlation Function (ACF) are equivalent. Therefore both ACF and CCF will be used equivalently in the following description.

(85) Furthermore, .sub.est,cur represents the current estimation of the code delay due to the propagation between the transmitter and the receiver. It must be noted that .sub.est,cur has the same meaning as in a conventional receiver where the reference signal s.sub.ref which is also called replica, is shifted according to .sub.est,cur.

(86) The reference receiver signal s.sub.ref, including the shift .sub.est,cur applied during correlation, is equal to:

(87) $s_{\mathrm{ref}}\left(t\right)=\underset{m=1}{\overset{M}{.Math.}}p\left(t-{}_{\mathrm{est},\mathrm{cur}}\right)e^{j\left(2f_m\left(t-{}_{\mathrm{est},\mathrm{cur}}\right)+{}_m\right)}.Math.a_m\left(t-{\mathrm{mT}}_c\right)$

(88) Calculation of the Cross Correlation Function (CCF) for this example application is the same as described with regards to the previous example application but now when L.sub.m represents one chip for all sub-carrier frequencies fm.

(89) It must be highlighted that the proposed applications of the invention can be extended to other cases where L.sub.m covers more than a single chip, but the sum of the M L.sub.m does not cover exactly one spreading code period, meaning

(90) $P=\underset{m=1}{\overset{M}{.Math.}}{L}_{m}L.$
This could be the case when L.sub.m covers for example 4 chips, and M equal 4 sub-carrier frequencies, letting P=16, while the spreading code period, L, encompasses more than 16 chips.

(91) As an example, a GPS L1CA code signal (BPSK(1)) is received when considering 2 sub-carrier frequencies (M=2) during the PRN code period, L, with 1023 code chips (corresponding to the Gold Code sequences used to generate the GPS L1 CA signals), and when the number of chips to which frequency f.sub.1 and f.sub.2 alternates from code period to code period in order to have an identical number of chips with frequency f.sub.1 and chips with frequency f.sub.2 over two successive GPS CA code periods covering 2046 chips. Here one cycle C covers P=L=1023 chips. It means that the set {L.sub.m}.sub.1m2,C per cycle C will differ from cycle to cycle as demonstrated hereafter.

(92) f.sub.1=1.023 MHz is applied during 512 chips over a code period, and is applied during 511 chips over the following code period, yielding the first set {L.sub.1=512, L.sub.2=511}.sub.1m2,C1 applicable during the first cycle C.sub.1.

(93) f.sub.2=1.023 MHz is applied during 511 chips over a code period, and is applied during 512 chips over the following code period, yielding the second set {L.sub.1=511, L.sub.2=512}.sub.1m2,C2 applicable during the second cycle C.sub.2.

(94) The BPSK(1) ACF as well as the modulation terms are shown in FIG. 5.

(95) The multiplication of the BPSK(1) ACF and modulation term leads to the ACF of the modulated BPSK signal mBPSK(1) visible on FIG. 6.

(96) In order to compare the performance tracking results in AWGN (Additive White Gaussian Noise), code jitter has been evaluated for the inventive receiver architecture and for a conventional receiver architecture. The code discriminator is a non-coherent early minus late receiver, a loop filter of order 1 with a bandwidth of 1 Hz and an integration time of 100 milliseconds and an Early Late spacing of 0.1 chip.

(97) The multipath robustness comparison is shown through the multipath envelope: FIG. 8 the envelope of the conventional receiver architecture, FIG. 9 shows the envelope of the inventive receiver architecture.

(98) It is must be noted that the same results would be obtained either if the frequency f.sub.1 would be applied during the first 512 chips of the first code period and to the first 511 chips of the second period, and the frequency f.sub.2 would be applied during the last 511 chips of the first code period and to the last 512 chips of the second period when the coherent integration time Tint would be computed with K=2, or if the frequencies f.sub.1 and f.sub.2 would alternate between consecutive chips (P=M=2 chips), as for example when f.sub.1 would be applied to the even chips and f.sub.2 to the odd chips, and when the coherent integration time T.sub.int would be computed with K=1023.

(99) As shown in the previous examples, the proposed inventive technique and related receiver architecture implementation leads to much better code jitter performance in AWGN and much more robust against multipath.

(100) If (f) denotes the PSD of the processing signal within one coherent integration (i.e. over the correlation time equal to the PRN code period), then the Gabor bandwidth is defined as (here (Rx denotes the two-sided receiver bandwidth):

(101) ${\mathrm{BW}}_{\mathrm{Gabor}}={}_{-{}_{\mathrm{rx}}/2}^{{}_{\mathrm{rx}}/2}{\left(\left(f\right)\right)}^2\mathrm{df}$

(102) It is proven that the code tracking performances and the multipath robustness performances are inversely proportional to the Gabor bandwidth. By using the proposed inventive technique BWGabor is synthetically increased, at receiver side, and this results in improved receiver performances.

(103) The improvement can even be better by choosing higher sub-carrier frequency, but the receiver sampling rate has to be updated accordingly (at least 2max(f.sub.m) plus the width of spectral occupancy, i.e. PSD, of the original pulse shape, for example the BPSK(1)).

(104) The invention allows reaching a major tracking performance improvement. This improvement can be reached without modifying the transmitted signal. Furthermore, when applying the multiplication with the sub-carrier in the digital domain, the required minimal bandwidth of the front end can be reduced to a minimum corresponding to the width of the original PSD for the pulse shape, such like a BPSK, and this represents an advantage in term of interference robustness. The methodology can be applied to any type of pulse shape.

(105) At least some of the functionalities of the invention may be performed by hard- or software. In case of an implementation in software, a single or multiple standard microprocessors or microcontrollers may be used to process a single or multiple algorithms implementing the invention. In case of an implementation in hardware, a FPGA (Field Programmable Gate Array) or ASIC (Application Specific Integrated Circuit) may be used.

(106) While at least one exemplary embodiment of the present invention(s) is disclosed herein, it should be understood that modifications, substitutions and alternatives may be apparent to one of ordinary skill in the art and can be made without departing from the scope of this disclosure. This disclosure is intended to cover any adaptations or variations of the exemplary embodiment(s). In addition, in this disclosure, the terms comprise or comprising do not exclude other elements or steps, the terms a or one do not exclude a plural number, and the term or means either or both. Furthermore, characteristics or steps which have been described may also be used in combination with other characteristics or steps and in any order unless the disclosure or context suggests otherwise. This disclosure hereby incorporates by reference the complete disclosure of any patent or application from which it claims benefit or priority.

REFERENCE NUMERALS AND ACRONYMS

(107) 10 first part of the receiver 11 first part of the receiver 100 antenna 102 band-pass filter 104 amplifier 106 multiplier 108 pre-processing unit 110 ADC 112 frequency generator 114 analog LO 116 digital LO 20 second part of the receiver 200 multiplier 202 correlate, integrate and dump unit 204 code discriminator 206 code loop filter 208 code NCO 210 code and pulse generator 212 multiplier 214 carrier discriminator 216 carrier loop filter 218 carrier NCO 220 second digital spectral offsetting signal generator 222 .sub.est,prec generator for the time offset .sub.est,prec used for synchronization 224 optional external estimator for the time offset .sub.est,prec used for synchronization ACF Auto Correlation Function ADC Analog Digital Converter AGC Automatic Gain Control ASIC Application Specific Integrated Circuit AWGN Additive White Gaussian Noise BPF Band Pass Filter BPSK Binary Phase Shift Keying BOC Binary Offset Carrier Modulation CF Correlation Function CCF Cross Correlation Function CDMA Code Division Multiple Access DSSS Direct Sequence Spread Spectrum FPGA Field Programmable Gate Array GNSS Global Navigation Satellite System GPS Global Positioning System LO Local Oscillator MP Multipath NCO Numerical Controlled Oscillator PRN Pseudo-Random Noise PSD Power Spectral Density