Wireless communication apparatus and wireless communication method

Abstract

A first wireless communication apparatus assigns a pilot signal without an effective signal component at least in an adjacent frequency component to a generated transmit signal, and transmits the transmit signal including the pilot signal. A second wireless communication apparatus converts the received signal or a frequency-converted signal obtained by frequency conversion of the signal into a signal in a frequency domain, sets an approximate value of the distance between the second wireless communication apparatus and the first wireless communication apparatus, calculates a coefficient γk, based on the approximate value of the distance, the effective bandwidth, the speed of light, the number of FFT points, and the frequency component number, extracts a signal in the frequency domain, generates a phase noise compensated sampling signal, and reproduces data transmitted by the first wireless communication apparatus.

Claims

1. A wireless communication apparatus in a wireless communication system including a first wireless communication apparatus and a second wireless communication apparatus, the first wireless communication apparatus comprising: a processor; and a storage medium having computer program instructions stored thereon, when executed by the processor, perform to: generate a transmit signal including information to be transmitted in a region or a portion of the region excluding an empty region within an effective bandwidth, the empty region being a predetermined frequency region within the effective bandwidth; generate a pilot signal without an effective signal component at least in an adjacent frequency component to a predetermined frequency component within the effective bandwidth and assign the pilot signal to the transmit signal; and transmit the transmit signal including the pilot signal generated by the pilot signal assigning section at a wireless frequency, and the second wireless communication apparatus comprising: a processor; and a storage medium having computer program instructions stored thereon, when executed by the processor, perform to: receive a signal of the wireless frequency; convert the received signal received by the reception section or a frequency-converted signal obtained by frequency conversion of the received signal from a sampling signal in a time domain to a signal in a frequency domain; set an approximate value L′ of a distance between the second wireless communication apparatus and the first wireless communication apparatus; calculate a coefficient γk given by Equation (1) below for the approximate value L′ of the distance, an effective bandwidth W, a speed of light c, a number of FFT points NFFT, and a frequency component number k; extract, a signal in a frequency domain including a frequency component of the pilot signal and a plurality of peripheral frequency components including at least adjacent frequency components of the pilot signal, and generate a replica of phase noise, based on a coefficient for each frequency component of the extracted signal and the coefficient γk; generate a phase noise compensated sampling signal by using the replica of the phase noise and the sampling signal in the time domain or a sampling signal modified based on the sampling signal in the time domain; and reproduce data transmitted by the first wireless communication apparatus, based on an output signal from the phase noise compensation section $\begin{array}{cc}\left[\mathrm{Math}.1\right]& \\ {\gamma }_k=e^{2\pi j\times \frac{L^{\prime }W}{c}\times \frac{k}{N_{\mathrm{FFT}}}}.& \left(1\right)\end{array}$

2. The wireless communication apparatus according to claim 1, wherein the computer program instructions further perform to allocate a subcarrier for a pilot signal to frequency components of both ends or to a frequency component of either end of the effective bandwidth, and set a neighboring subcarrier including an adjacent subcarrier as an empty subcarrier.

3. The wireless communication apparatus according to claim 1, wherein the first wireless communication apparatus includes: a memory that stores sampling data of a length of one cycle or an integer multiple times of the cycle of a sine wave signal of a predetermined frequency or a synthesized signal of a plurality of sine wave signals of predetermined frequencies; and wherein the computer program instructions further perform to output continuous time domain signals of the pilot signal by repeatedly reading the sampling data from the memory at predetermined intervals.

4. The wireless communication apparatus according to claim 1, wherein the computer program instructions further perform to generate sampling data at a time t by following Equation (2) or sampling data given by an inverse of Equation (2) as a replica of phase noise, based on a coefficient βk of the pilot signal of a k-th frequency component and a coefficient βk+k′ of a (k+k′)-th frequency component, for a positive integer NPN greater than or equal to 1 and an integer k′ which satisfies −NPN≤k′≤NPN $\begin{array}{cc}\left[\mathrm{Math}.2\right]& \\ {\Phi }_k\left(t\right)={\left(\stackrel{N_{\mathrm{PS}}}{\underset{k^{\prime }=-N_{\mathrm{PS}}}{.Math.}}\frac{{\alpha }_{k+k^{\prime }}}{{\alpha }_k}\times {\gamma }_{k+k^{\prime }}{e}^{2\pi {\mathrm{jk}}^{\prime }\Delta ft}\right)}^{-1}.& \left(2\right)\end{array}$

5. The wireless communication apparatus according to claim 1, wherein the computer program instructions further perform to generate sampling data at a time t by following Equations (3) and (4) or sampling data given by an inverse of Equation (4) as a replica of phase noise, based on a coefficient βk of the pilot signal of a k-th frequency component and a coefficient βk+k′ of a (k+k′)-th frequency component, for a positive integer NPN greater than or equal to 1 and an integer k′ which satisfies −NPN≤k′≤NPN $\begin{array}{cc}\left[\mathrm{Math}.3\right]& \\ {\Psi }_k\left(t\right)\approx \stackrel{N_{\mathrm{PS}}}{\underset{k^{\prime }=-N_{\mathrm{PS}}}{.Math.}}{\beta }_{k+k^{\prime }}\times {\gamma }_{k+k^{\prime }}{e}^{2\pi j\left(f_k+k^{\prime }\Delta f\right)t}& \left(3\right)\end{array}$ $\begin{array}{cc}\left[\mathrm{Math}.4\right]& \\ {{\Phi }_k\left(t\right)}^{-1}={\Psi }_k\left(t\right)\times \frac{1}{{\beta }_k}{e}^{-2\pi {\mathrm{jf}}_{k}t}.& \left(4\right)\end{array}$

6. The wireless communication apparatus according to claim 1, wherein the computer program instructions further perform to remove the pilot signal and predetermined frequency components around the pilot signal from a receive signal.

7. A wireless communication method performed by a wireless communication apparatus in a wireless communication system including a first wireless communication apparatus and a second wireless communication apparatus, the wireless communication method comprising: generating, by the first wireless communication apparatus, a transmit signal including information to be transmitted in a region or a portion of the region excluding an empty region within an effective bandwidth, the empty region being a predetermined frequency region within the effective bandwidth; generating, by the first wireless communication apparatus, a pilot signal without an effective signal component at least in an adjacent frequency component to a predetermined frequency component within the effective bandwidth and assigning the pilot signal to the transmit signal generated by the generating of the transmit signal; transmitting, by the first wireless communication apparatus, the transmit signal including the pilot signal generated by the assigning of the pilot signal at a wireless frequency; receiving, by the second wireless communication apparatus, a signal of the wireless frequency; converting, by the second wireless communication apparatus, the received signal received by the receiving or a frequency-converted signal obtained by frequency conversion of the received signal from a sampling signal in a time domain to a signal in a frequency domain; setting, by the second wireless communication apparatus, an approximate value L′ of a distance between the second wireless communication apparatus and the first wireless communication apparatus; calculating, by the second wireless communication apparatus, a coefficient γk given by Equation (5) below for the approximate value L′ of the distance, an effective bandwidth W, a speed of light c, a number of FFT points NFFT of the time/frequency conversion section, and a frequency component number k; extracting, by the second wireless communication apparatus, from an output in the converting of the signal, a signal in a frequency domain including a frequency component of the pilot signal and a plurality of peripheral frequency components including at least adjacent frequency components of the pilot signal, and generating a replica of phase noise, based on a coefficient for each frequency component of the extracted signal and the coefficient γk; generating, by the second wireless communication apparatus, a phase noise compensated sampling signal by using the replica of the phase noise and the sampling signal in the time domain or a sampling signal modified based on the sampling signal in the time domain; and reproducing, by the second wireless communication apparatus, data transmitted by the first wireless communication apparatus, based on an output signal in the compensating of the phase noise $\begin{array}{cc}\left[\mathrm{Math}.5\right]& \\ {\gamma }_k=e^{2\pi j\times \frac{L^{\prime }W}{c}\times \frac{k}{N_{\mathrm{FFT}}}}.& \left(5\right)\end{array}$

Description

BRIEF DESCRIPTION OF DRAWINGS

(1) FIG. 1 is a diagram illustrating a circuit configuration of a wireless station apparatus according to a first embodiment.

(2) FIG. 2 is a schematic block diagram illustrating an example of a configuration of a transmitter of the wireless station apparatus according to the first embodiment.

(3) FIG. 3 is a schematic block diagram illustrating an example of a configuration of a receiver of the wireless station apparatus according to the first embodiment.

(4) FIG. 4 is a schematic block diagram illustrating an example of a configuration of an extended FFT circuit according to the first embodiment.

(5) FIG. 5 is a schematic block diagram illustrating another example of a configuration of an extended FFT circuit according to the first embodiment.

(6) FIG. 6 is a schematic block diagram illustrating an example of a configuration of a transmitter of a wireless station apparatus according to a second embodiment.

(7) FIG. 7 is a schematic block diagram illustrating an example of a configuration of a receiver of the wireless station apparatus according to the second embodiment.

(8) FIG. 8 is a diagram illustrating a schematic block diagram illustrating an example of a configuration of a transmitter of a wireless station apparatus according to the third embodiment.

(9) FIG. 9 is a diagram illustrating a schematic block diagram illustrating an example of a configuration of a receiver of the wireless station apparatus according to a third embodiment.

(10) FIG. 10 is a diagram illustrating a configuration of an extended FFT circuit according to a fourth embodiment.

(11) FIG. 11 is a schematic block diagram illustrating another example of a configuration of an extended FFT circuit according to the fourth embodiment.

(12) FIG. 12 is a diagram illustrating a configuration example of a single carrier compensation circuit according to a fifth embodiment.

(13) FIG. 13 is a diagram illustrating another configuration example of a single carrier compensation circuit according to the fifth embodiment.

(14) FIG. 14 is a diagram illustrating a schematic block diagram illustrating an example of a configuration of a transmitter during single carrier transmission according to the fifth embodiment.

(15) FIG. 15 is a diagram illustrating a schematic block diagram illustrating an example of a configuration of a receiver during single carrier transmission according to the fifth embodiment.

(16) FIG. 16 is a diagram illustrating an overview of an opposing wireless station apparatus according to a sixth embodiment.

(17) FIG. 17 is a diagram illustrating a circuit configuration of a wireless station apparatus of related art.

(18) FIG. 18 is a schematic block diagram illustrating an example of a configuration of a transmitter in the wireless station apparatus.

(19) FIG. 19 is a schematic block diagram illustrating an example of a configuration of a receiver in the wireless station apparatus.

(20) FIG. 20 is a schematic block diagram illustrating an example of a configuration of a transmitter of a wireless station apparatus according to NPL 1.

(21) FIG. 21 is a schematic block diagram illustrating an example of a configuration of a receiver of a wireless station apparatus according to NPL 1.

(22) FIG. 22 is a schematic block diagram illustrating an example of a configuration of a transmitter of a wireless station apparatus according to NPL 2.

(23) FIG. 23 is a diagram illustrating a specific example of a waveform of an OFDM signal in related art.

(24) FIG. 24 is a schematic block diagram illustrating an example of a configuration of a receiver of a wireless station apparatus according to NPL 2.

(25) FIG. 25 is a diagram illustrating a specific example of a waveform of an OFDM signal according to related art.

(26) FIG. 26 is a diagram illustrating a relationship between a transmission training signal and a reception waveform thereof according to related art.

(27) FIG. 27 is a diagram illustrating a relationship between waveforms of a transmit signal and a receive signal according to related art.

DESCRIPTION OF EMBODIMENTS

(28) In the following description, the processing of the related inventions which are the background of the present invention will be cited for description.

Basic Principles of Present Invention

(29) The fundamental cause of the above problem is because the channel information used to estimate the phase noise includes a term of phase rotation proportional to the distance. For example, assuming that the approximate value of the distance of the two wireless station apparatuses is L, the cancellation of the phase rotation amount proportional to the distance can be performed by the coefficient γ.sub.k for the subcarrier number k indicated by the following Equation (10).

(30) $\begin{array}{cc}\left[\mathrm{Math}.10\right]& \\ {\gamma }_k=e^{2\pi j\times \frac{L^{\prime }W}{c}\times \frac{k}{N_{\mathrm{FFT}}}}& \left(10\right)\end{array}$

(31) Here, for the subcarrier number of k, the center frequency may be considered as zero, or the subcarrier having the lowest frequency may have k=1. In any case, all the subcarriers are only multiplied by a predetermined coefficient, and thus, for the subcarrier number of k, the subcarrier number may be assigned in a manner that is added one at a time from the lower to the higher of the frequency. With such a configuration, for the influence of the phase rotation proportional to the distance per subcarrier, the error can be suppressed to the value of the error of the estimated distance L instead of the actual distance L, that is, the degree of δL=(L−L). As an example, assuming that L=100 [m], δL=3 [m], and W=1 [GHz], the original phase rotation N.sub.p (=L*W/c) is approximately 333, and the time lag of 333 samples cannot be ignored. On the other hand, in a case that the above-described compensation is performed, the phase rotation amount remaining in the bandwidth is 2π*(L−L)*W/c, and in a case where this is let to be N.sub.p, N.sub.p is approximately 10. In a case where the error is approximately 10 samples, the amount of phase shift due to phase noise is expected to be within error margin, so the problem described above can be generally solved.

(32) Note that the measurement of this distance may be measured directly by using a laser pointer type distance measurement apparatus, or may be calculated based on the information of the installation location. In any case, in a case where the distance can be determined with an error of a few percent, the time lag of the replica of the phase noise will be significantly modified.

(33) The coefficient determined in this manner can be used, for example, by replacing Equation (5) with the following Equation (11).

(34) 0$\begin{array}{cc}\left[\mathrm{Math}.11\right]& \\ {\Phi }_k\left(t\right)={\left(\stackrel{N_{\mathrm{PS}}}{\underset{k^{\prime }=-N_{\mathrm{PS}}}{.Math.}}\frac{{\alpha }_{k+k^{\prime }}}{{\alpha }_k}\times {\gamma }_{k+k^{\prime }}{e}^{2\pi {\mathrm{jk}}^{\prime }\Delta \mathrm{ft}}.\right)}^{-1}& \left(11\right)\end{array}$

(35) Alternatively, the coefficient determined in this manner can be used by replacing Equation (7) with the following Equation (12).

(36) $\begin{array}{cc}\left[\mathrm{Math}.12\right]& \\ {\Psi }_k\left(t\right)\approx \stackrel{N_{\mathrm{PS}}}{\underset{k^{\prime }=-N_{\mathrm{PS}}}{.Math.}}{\beta }_{k+k^{\prime }}\times {\gamma }_{k+k^{\prime }}{e}^{2\pi j\left(f_k+k^{\prime }\Delta f\right)t}& \left(12\right)\end{array}$

(37) In the present invention, the coefficient γ.sub.k is calculated in the wireless station apparatus of the receiving side so as to cancel the phase rotation amount proportional to the distance as described above. The wireless station apparatus of the receiving side generates a replica signal of the phase noise by using the calculated coefficient γ.sub.k and the coefficient for each frequency component of the signal, and cancels the phase rotation amount by using the generated replica signal of the phase noise. The problem described above is solved by performing such a process.

(38) Features of the circuit configurations in each embodiment will be described below with reference to the drawings. For a representative example, an OFDM modulation scheme is described as an example herein, but extension is possible even in a case of other single carrier transmission, which will be described later. Although the transmission weight and the reception weight in the time domain are also referred to herein as the time axis transmission weight and the time axis reception weight, these “time axis” and “time domain” are synonymous.

First Embodiment

(39) Hereinafter, the first embodiment of the present invention will be described with mainly reference to diagrams related to circuit configurations.

(40) Circuit Configuration According to First Embodiment

(41) FIG. 1 is a diagram illustrating a circuit configuration of a wireless station apparatus 70 according to the first embodiment. As illustrated in FIG. 1, the wireless station apparatus 70 includes a transmitter 71, a receiver 75, an interface circuit 67, a Medium Access Control (MAC) layer processing circuit 68, and a communication control circuit 51.
The wireless station apparatus 70 inputs/outputs data from/to an external device or a network via the interface circuit 67. The interface circuit 67 detects data to be transferred on the wireless circuit from the data input, and outputs the detected data to the MAC layer processing circuit 68. The MAC layer processing circuit 68 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 51 configured to perform management control of the operation of the entire wireless station apparatus 70. In MIMO transmission, for spatially multiplexing and transmitting signals to one wireless station apparatus 70, signal sequences of a plurality of systems are output from the MAC layer processing circuit 68 to the transmitter 71.

(42) FIG. 2 is a schematic block diagram illustrating an example of a configuration of the transmitter 71 of the wireless station apparatus 70 according to the first embodiment. As illustrated in FIG. 2, the transmitter 71 includes: transmit signal processing circuits 311-1 to 311-N.sub.SDM; addition synthesis circuits 812-1 to 812-N.sub.Ant; IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant; D/A converters 814-1 to 814-N.sub.Ant; a local oscillator 815; mixers 816-1 to 816-N.sub.Ant; filters 817-1 to 817-N.sub.Ant; high power amplifiers 818-1 to 818-N.sub.Ant; antenna elements 819-1 to 819-N.sub.Ant; adders 320-1 to 320-N.sub.Ant; a pilot signal storage circuit 321; and a transmission weight processing unit 840. The transmit signal processing circuits 311-1 to 311-N.sub.SDM and the transmission weight processing unit 840 are connected to the communication control circuit 51.

(43) The transmission weight processing unit 840 includes a channel information acquisition circuit 841, a channel information storage circuit 842, and a transmission weight calculation circuit 843. Here, the subscript N.sub.SDM of the transmit signal processing circuits 311-1 to 311-N.sub.SDM in FIG. 2 represents the number of multiplexing for performing spatial multiplexing at the same time. N.sub.SDM representing the number of multiplexing for performing spatial multiplexing at the same time is the same in the following embodiments. The subscript N.sub.Ant of the circuits from the addition synthesis circuits 812-1 to 812-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant represents the number of antenna elements provided in the wireless station apparatus 70. N.sub.ANT representing the number of antenna elements provided in the wireless station apparatus 70 is the same in the following embodiments.

(44) Here, the transmit signal processing circuits 311-1 to 311-N.sub.SDM differ from the transmit signal processing circuits configured to generate the OFDM signal having the effective bandwidth W illustrated in FIG. 23 in that the transmit signal processing circuits 311-1 to 311-N.sub.SDM generate an OFDM signal having a reduced effective bandwidth W″ in FIG. 25, but other functions are the same. The transmit signal processing circuits 311-1 to 311-N.sub.SDM perform signal processing of the reduced effective bandwidth W″, and thus the transmission weight input from the transmission weight processing unit 840 is only for a subcarrier for allocation.

(45) In the background art, because a single wireless station apparatus 70 spatially multiplexes and transmits signals to other wireless station apparatuses 70, signal sequences of a plurality of systems are input from the MAC layer processing circuit 68 to the transmitter 71, and the input signal sequences of a plurality of systems are input to the transmit signal processing circuits 311-1 to 311-N.sub.SDM. The transmit signal processing circuits 311-1 to 311-N.sub.SDM perform modulation processing on the data (data input #1 to #N.sub.SDM) to be transmitted to a destination wireless station apparatus 70 when the data to be transmitted (wireless packets) is input from the MAC layer processing circuit 68 on a wireless circuit. Here, modulation processing is performed on signals of each signal sequence for each subcarrier in the reduced effective bandwidth W″ illustrated in FIG. 25, for example, in a case of using an OFDM modulation scheme. Furthermore, the transmit signal processing circuits 311-1 to 311-N.sub.SDM multiply the baseband signal after the modulation processing with the transmission weight for each subcarrier.

(46) The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819-N.sub.Ant is subjected to a remaining signal processing as necessary, and the signal is input from each of the transmit signal processing circuits 311-1 to 311-N.sub.SDM to the addition synthesis circuits 812-1 to 812-N.sub.Ant as a signal in the frequency domain of the transmit signal in the baseband. The signal input to the addition synthesis circuits 812-1 to 812-N.sub.Ant is synthesized for each subcarrier. The synthesized signal is converted from a signal on the frequency axis to a signal on the time axis in the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant, and is further subjected to processing such as insertion of a guard interval or waveform shaping in between OFDM symbols (between blocks of block transmission in a case of SC-FDE). Sampling data of the pilot signal is output from the pilot signal storage circuit 321 and is added to the output signal from the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant by the adders 320-1 to 320-N.sub.Ant. From the pilot signal storage circuit 321, the sampling data is repeatedly output in a form that each sine wave waveform of the pilot signal is continuous. The signal added by the adders is converted for each system of the antenna elements 819-1 to 819-N.sub.Ant from digital sampling data to analog signals in the baseband at the D/A converters 814-1 to 814-N.sub.Ant. Further, each analog signal is multiplied by a local oscillating signal input from the local oscillator 815 by the mixers 816-1 to 816-N.sub.Ant and up-converted to a wireless frequency signal. Here, the up-converted signal includes a signal in a region outside of the band of the channel to be transmitted, so that a signal outside of the band is removed at the filters 817-1 to 817-N.sub.Ant to generate a signal to be transmitted. The generated signal is amplified by the high power amplifiers 818-1 to 818-N.sub.Ant and transmitted from the antenna elements 819-1 to 819-N.sub.Ant.

(47) Note that, in FIG. 2, processing such as IFFT processing, insertion of a guard interval, waveform shaping, or the like is performed after the addition synthesis of the signal of each subcarrier is performed by the addition synthesis circuits 812-1 to 812-N.sub.Ant, but by causing these processes to be performed in the transmit signal processing circuits 311-1 to 311-N.sub.SDM, and by synthesizing the sampling signals on the time axis after IFFT by the addition synthesis circuits 812-1 to 812-N.sub.Ant, the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant may be omitted (precisely, the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant may be included in the transmit signal processing circuits 311-1 to 311-N.sub.SDM). In this case, remaining signal processing as necessary after the transmission weight multiplication in the transmit signal processing circuits 311-1 to 311-N.sub.SDM refers to processing such as IFFT processing, insertion of a guard interval, waveform shaping, or the like.

(48) The transmission weight multiplied by the transmit signal processing circuits 311-1 to 311-N.sub.SDM is acquired from the transmission weight calculation circuit 843 included in the transmission weight processing unit 840 during the signal transmission processing. The transmission weight processing unit 840 separately acquires the channel information acquired at the receiver 75 via the communication control circuit 51 in the channel information acquisition circuit 841, and stores the channel information in the channel information storage circuit 842 while sequentially updating the channel information. Upon transmission of the signal, in accordance with an instruction from the communication control circuit 51, the transmission weight calculation circuit 843 reads the channel information corresponding to the destination station from the channel information storage circuit 842, and calculates the transmission weight on the basis of the read channel information. The transmission weight calculation circuit 843 outputs the calculated transmission weight to the transmit signal processing circuits 311-1 to 311-N.sub.SDM. In a case that the wireless station apparatus is a base station, the communication control circuit 51 manages which terminal station apparatus the destination station is to communicate with a plurality of terminal station apparatuses.

(49) Note that the signals of the N.sub.SDM systems output from the transmit signal processing circuits 311-1 to 311-N.sub.SDM are synthesized in the addition synthesis circuits 812-1 to 812-N.sub.Ant, and the following D/A converters 814-1 to 814-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant are used together, but the signals may be implemented individually from the following D/A converters 814-1 to 814-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant without being synthesized at the addition synthesis circuits 812-1 to 812-N.sub.Ant, and a subarray may be configured by the antenna elements 819-1 to 819-N.sub.Ant in each of the antennas. Furthermore, in this case, the transmission weight calculation circuit 843 can use a virtual transmission line corresponding to the first singular value between the array antennas or the subarrays in the transmitter 71 of a wireless station apparatus 70 and in the receiver 75 of a wireless station apparatus 70 in the calculation of the transmission weight. There are several variations in the method of channel estimation and the method of calculating the transmission and/or reception weight in a case of utilizing the virtual transmission line corresponding to the first singular value. For example, a first right singular vector upon singular value decomposition may be used for the transmission weight vector for each channel matrix from the wireless station apparatus 70 towards the antenna elements 819-1 to 819-N.sub.Ant of another wireless station apparatus 70. In this case, the transmission weight calculation circuit 843 has the function of calculating this first right singular vector. Otherwise, various approaches to acquiring an approximate solution of such a singular vector may be used.

(50) For example, a first right singular vector upon singular value decomposition may be used for the transmission weight vector for each channel matrix from the wireless station apparatus 70 towards the antenna elements 819-1 to 819-N.sub.Ant of another wireless station apparatus 70. In this case, the transmission weight calculation circuit 843 has the function of calculating this first right singular vector. Otherwise, various approaches to acquiring an approximate solution of such a singular vector may be used.

(51) In the above description, the assignment of the pilot signal at both ends of the effective bandwidth is performed at the adders 320-1 to 320-N.sub.Ant between the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant and the D/A converters 814-1 to 814-N.sub.Ant. In contrast, a configuration may be adopted in which a signal allocated a pilot signal to the subcarriers on both ends of the effective bandwidth is generated in the transmit signal processing circuits 311-1 to 311-N.sub.SDM, and a predetermined signal processing such as multiplication of the transmission weight is performed on the signal. In this case, the adders 320-1 to 320-N.sub.Ant and the pilot signal storage circuit 321 are unnecessary, and equivalent processing is performed in the transmit signal processing circuits 311-1 to 311-N.sub.SDM.

(52) Next, FIG. 3 is a schematic block diagram illustrating an example of a configuration of the receiver 75 of the wireless station apparatus 70 according to the first embodiment. As illustrated in FIG. 3, the receiver 75 includes: antenna elements 851-1 to 851-N.sub.Ant; low noise amplifiers (LNA) 852-1 to 852-N.sub.Ant; a local oscillator 853: mixers 854-1 to 854-N.sub.Ant; filters 855-1 to 855-N.sub.Ant; A/D (analog to digital) converters 856-1 to 856-N.sub.Ant; extended FFT circuits 357-1 to 357-N.sub.Ant; receive signal processing circuits 345-1 to 345-N.sub.SDM; and a reception weight processing unit 844. The receive signal processing circuits 345-1 to 345-N.sub.SDM and the reception weight processing unit 844 are connected to the communication control circuit 51 illustrated in FIG. 1. The reception weight processing unit 844 includes a channel information estimation circuit 846 and a reception weight calculation circuit 847.

(53) First, a signal received at the antenna elements 851-1 to 851-N.sub.Ant is amplified by the low noise amplifiers 852-1 to 852-N.sub.Ant. The amplified signal and a local oscillating signal output from the local oscillator 853 are multiplied by the mixers 854-1 to 854-N.sub.Ant, and the amplified signal is down-converted from the wireless frequency signal to the baseband signal. The down-converted signal also includes signals outside of the frequency band to be received, so that the filters 855-1 to 855-N.sub.Ant remove out-of-band components. The signal from which the out-of-band components have been removed is converted to a digital baseband signal by the A/D converters 856-1 to 856-N.sub.Ant. For example, in a case that OFDM is used, the digital baseband signal is input to the extended FFT circuits 357-1 to 357-N.sub.Ant, the extended FFT circuits 357-1 to 357-N.sub.Ant perform the phase noise compensation processing described below, and a signal on the time axis is converted (separated into a signal of each subcarrier) to a signal on the frequency axis at a predetermined symbol timing determined by a circuit for timing detection of which description is omitted herein. The signal separated into each subcarrier is input to the receive signal processing circuits 345-1 to 345-N.sub.SDM, and is also input to the channel information estimation circuit 846.

(54) In the channel information estimation circuit 846, a channel vector of channel information between the antenna elements 819-1 to 819-N.sub.Ant on the transmitting station side and the antenna elements 851-1 to 851-N.sub.Ant on the receiving station side are estimated for each subcarrier, based on a known signal for channel estimation separated into each subcarrier (such as a preamble signal assigned to the head of the wireless packet), and the estimation result is output to the reception weight calculation circuit 847. In the reception weight calculation circuit 847, the reception weight to be multiplied is calculated for each subcarrier, based on the input channel information.

(55) For this reception weight, for example, a ZF type pseudo-inverse is utilized as described above, or an MMSE type reception weight matrix is utilized. At this time, the reception weight vectors for synthesizing the signals received at each of the antenna elements 851-1 to 851-N.sub.Ant are different from each other for each signal sequence, correspond to a row vector, such as the ZF type pseudo-inverse matrix or the MMSE type reception weight matrix described above, and are input to the receive signal processing circuits 345-1 to 345-N.sub.SDM corresponding to the signal sequence to be extracted.

(56) In the receive signal processing circuits 345-1 to 345-N.sub.SDM, the reception weight input from the reception weight calculation circuit 847 is multiplied by the signal input from the extended FFT circuits 357-1 to 357-N.sub.Ant for each subcarrier, and the signals received at each of the antenna elements 851-1 to 851-N.sub.Ant are added and synthesized for each subcarrier. The receive signal processing circuits 345-1 to 345-N.sub.SDM perform demodulation processing on the added and synthesized signals, and output the reproduced data to the MAC layer processing circuit 68.

(57) Here, different signal processing of signal sequences are performed in different receive signal processing circuits 345-1 to 345-N.sub.SDM. MLD or simple MLD using QR decomposition or the like may be used as the receive signal processing across the plurality of receive signal processing circuits 345-1 to 345-N.sub.SDM. The MAC layer processing circuit 68 performs processing related to the MAC layer (e.g., conversion of data input and output to and from the interface circuit 67, and data transmitted and/or received on the wireless circuit, i.e., wireless packets, termination of header information of the MAC layer, or the like). The receive data processed by the MAC layer processing circuit 68 is output to an external device or a network via the interface circuit 67. The communication control circuit 51 manages control related to the overall communication, such as overall timing control.

(58) Similarly to the transmitter 71, the receiver 75 uses the antenna elements 851-1 to 851-N.sub.Ant to the extended FFT circuits 357-1 to 357-N.sub.Ant together, and copies the output from the extended FFT circuits 357-1 to 357-N.sub.Ant into N.sub.SDM systems to input to the individual receive signal processing circuits 345-1 to 345-N.sub.SDM. In contrast, the antenna elements 851-1 to 851-N.sub.Ant to the extended FFT circuits 357-1 to 357-N.sub.Ant may be implemented individually, and each of the antenna elements 851-1 to 851-N.sub.Ant may be implemented to have a subarray configuration.

(59) Furthermore, in this case, a virtual transmission line corresponding to the first singular value may be used between the array antennas or the subarrays in the transmitter 71 of a wireless station apparatus 70 and in the receiver 75 of a wireless station apparatus 70 in the calculation of the reception weight. There are several variations in the method of channel estimation and the method of calculating the transmission and/or reception weight in a case of utilizing the virtual transmission line corresponding to the first singular value. For example, a first left singular vector upon singular value decomposition may be used for the reception weight vector for each channel matrix from another wireless station apparatus 70 towards the antenna elements 819-1 to 819-N.sub.Ant of the self-station. In this case, the reception weight calculation circuit 847 has the function of calculating this first left singular vector. Otherwise, various approaches to acquiring an approximate solution of such a singular vector may be used.

(60) The phase noise compensation processing performed by the extended FFT circuits 357-1 to 357-N.sub.Ant will be described below. FIG. 4 is a schematic block diagram illustrating an example of a configuration of the extended FFT circuit 357 according to the first embodiment. As illustrated in FIG. 4, a block 180 (FIG. 4(A)) corresponding to a functional block of an FFT circuit according to related art is changed to a block 190 (FIG. 4(B)) in the extended FFT circuit according to the first embodiment. As illustrated in FIG. 4(B), the extended FFT circuit 357 includes a replication circuit 181, an FFT circuit 182, a function ϕ (t) acquisition circuit 183, an IFFT circuit 184, a phase noise compensation circuit 185, a γ.sub.k setting circuit 193, a distance L setting circuit 194, and an FFT circuit 857. The extended FFT circuits 357 is connected to the receive signal processing circuit 345, the reception weight processing unit 844, and the A/D converter 856 in FIG. 3.

(61) First, when the wireless station apparatus is placed before the operation is started, the installer calculates an approximate value L of the distance between two wireless station apparatuses in any approach, and sets this by the distance L setting circuit 194. For this setting, there may be a section for numerical input on the apparatus, or a configuration may be used in which another apparatus for control (such as a PC) is connected and setting is performed from outside. The distance L′ set in this way is input to the γ.sub.k setting circuit 193, γ.sub.k is calculated by Equation (10) by using the L′ in the γ.sub.k setting circuit 193, and this is input to the function Φ (t) acquisition circuit 183 for preliminary preparation.

(62) Next, digital sampling data is input from the A/D converter 856 to the extended FFT circuits 357 (block 190), the sampling data is replicated in the replication circuit 181, and one of the replicated sampling data is input to the FFT circuit 182 and the other is input to the phase noise compensation circuit 185. The FFT circuit 182 performs FFT on the sampling data cut at a predetermined symbol timing determined by the timing detection unit not illustrated in the drawing, extracts components related to the pilot signal and the subcarrier regions 917 and 918 in the vicinity of this in FIG. 3, and inputs this information into the function Φ (t) acquisition circuit 183. The function Φ (t) acquisition circuit 183 calculates the value corresponding to the coefficient γ.sub.k+k*α.sub.k+k/α.sub.k of the function Φ.sub.k (t) of Equation (5) by the approach described above by the operation of the coefficient described above γ.sub.k+k*β.sub.k+k/β.sub.k, based on the output of the FFT circuit 182. Thus, the coefficient in the frequency domain of the inverse of the function Φ (t) is determined.

(63) Furthermore, based on this information, the IFFT circuit 184 converts the signal in the frequency domain into a signal in the time domain, and inputs the signal in the time domain into the phase noise compensation circuit 185. The phase noise compensation circuit 185 multiplies the values of the inverse of the time domain signal input from the IFFT circuit 184 (i.e., function ϕ (t)) and the time domain signal to be input from the replication circuit 181 (precisely, the time axis signal corresponding to the region from which the guard interval has been removed with FFT performed in the FFT circuit 182) for each sampling data as illustrated in Equation (6), and reproduces the signal in the time domain on which phase noise compensation has been performed. This reproduced signal in the time domain (sampling data) is input to the FFT circuit 857, and is again converted from the signal in the time domain to the signal in the frequency domain. However, the signal in the frequency domain is a signal compensated for phase noise, and the leakage of power between subcarriers is suppressed. The signal in the frequency domain is input to the receive signal processing circuit 345, and the receive signal processing circuit 345 performs predetermined receive signal processing.

(64) Although the signal processing performed by the receive signal processing circuit 345 is basically equivalent to the conventional receive signal processing circuit, the receive signal processing circuit of related art processes the OFDM signal of the effective bandwidth W illustrated in FIG. 23, whereas in the embodiment of the present invention, the receive signal processing circuit 345 performs receive signal processing of the reduced effective bandwidth W″ illustrated in FIG. 25. At this time, the signal components of the pilot signal or an empty subcarrier region are neglected to perform processing and thus the signal components are illustrated as different functional blocks precisely, but while the range of the effective subcarrier is different, the signal processing other than that is equivalent at all. Thus, the configuration is equivalent at all to the known configuration except that the effective bandwidth W s changed to the reduced effective bandwidth W″ in this manner, including the output of the information to the reception weight processing unit 844.

(65) For another configuration to achieve a similar effect, the configuration of FIG. 5 may be used in which processing is performed in Equation (12) instead of Equation (11). In FIG. 5, in addition to FIG. 4, a function ψ (t) acquisition circuit 195, and a function ϕ (t) acquisition circuit 196 are added, and the function ϕ (t) acquisition circuit 183 is deleted.

(66) In FIG. 5, the configuration is the same as FIG. 4 until γ.sub.k is calculated according to Equation (10), but is only different in that this result is input to the function ψ (t) acquisition circuit 195 for preliminary preparation.

(67) As a difference of processing other than this, the FFT circuit 182 performs FFT on the sampling data cut at a predetermined symbol timing determined by the timing detection unit not illustrated in the drawing, extracts components related to the pilot signal and the subcarrier regions 917 and 918 in the vicinity of this in FIG. 3, and inputs this information into the function ψ (t) acquisition circuit 195. The function ψ (t) acquisition circuit 195 calculates the value corresponding to the coefficient γ.sub.k+k*β.sub.k+k of the function ψ.sub.k (t) of Equation (12), based on the output of the FFT circuit 182. Thus, the coefficient in the frequency domain of the function ψ (t) is determined.

(68) Furthermore, based on this information, the IFFT circuit 184 converts the signal in the frequency domain into a signal in the time domain, and inputs the signal to the function ϕ (t) acquisition circuit 196. The function ϕ (t) acquisition circuit 196 calculates the function ϕ (t) by using Equation (8) and inputs the signal in the time domain into the phase noise compensation circuit 185. Processing other than this is similar to FIG. 4.

(69) According to the wireless station apparatus 70 configured as described above, the phase rotation amount is canceled by using an approximate value of the distance between the two wireless station apparatuses 70 obtained by using some approach (for example, it may be measured directly by using a laser pointer type distance measurement apparatus, or calculated based on the information of the installation location). In this way, the time lag of the replica of the phase noise can be significantly modified.

(70) As a result, a decrease in throughput can be suppressed.

Second Embodiment

(71) In the first embodiment, a description has been given of a case that phase noise compensation is performed for a circuit configuration in which an FFT circuit is implemented for each antenna element. However, in the background art, there is a signal processing technique that aggregates signals of antennas of a plurality of elements for each signal sequence to be transmitted, and limits the signal processing of FFT or IFFT to the number of signal sequences to be spatially multiplexed. Thus, in the second embodiment, an embodiment in a case that the present invention is applied to the technique described in NPL 1 will be described.

(72) Circuit Configuration According to Second Embodiment

(73) In the second embodiment of the present invention, the configuration of the wireless station apparatus 70 takes an equivalent configuration as the wireless station apparatus 70 illustrated in FIG. 1. The difference from FIG. 1 is only that the transmitter 71 is replaced with a transmitter 72a, the receiver 75 is replaced with a receiver 76a, and the communication control circuit 51 is replaced with a communication control circuit 52, so that the overall functions and features are in accordance with FIG. 1, and the details of the drawings and descriptions are omitted herein.

(74) FIG. 6 is a schematic block diagram illustrating an example of a configuration of the transmitter 72a of the wireless station apparatus 70 according to the second embodiment. As illustrated in FIG. 6, the transmitter 72a includes: transmit signal processing circuits 411-1 to 411-N.sub.SDM; addition synthesis circuits 812-1 to 812-N.sub.Ant; IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM; D/A converters 814-1 to 814-N.sub.Ant; a local oscillator 815; mixers 816-1 to 816-N.sub.Ant; filters 817-1 to 817-N.sub.Ant; high power amplifiers 818-1 to 818-N.sub.Ant; antenna elements 819-1 to 819-N.sub.Ant; a transmission weight processing unit 740; and time axis transmission weight multiplication circuits 761-1 to 761-N.sub.SDM; adders 420-1 to 420-N.sub.SDM; and a pilot signal storage circuit 321. The transmit signal processing circuits 411-1 to 411-N.sub.SDM and the transmission weight processing unit 740 are connected to the communication control circuit 52. The transmission weight processing unit 740 includes a channel information acquisition circuit 741, a channel information storage circuit 742, and a time axis transmission weight calculation circuit 743.

(75) The difference between the configurations of the transmitters described in FIG. 6 and NPL 1 is that the adders 420-1 to 420-N.sub.SDM are disposed between the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM and the time axis transmission weight multiplication circuits 761-1 to 761-N.sub.SDM, the pilot signal storage circuit 321 is further added, and the transmit signal processing circuits 711-1 to 711-N.sub.SDM are changed to the transmit signal processing circuits 411-1 to 411-N.sub.SDM.

(76) This is similar to the first embodiment in that the difference between the configuration of related art and FIG. 2 is that the adders 320-1 to 320-N.sub.Ant are added between the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant and the D/A converters 814-1 to 814-N.sub.Ant, the pilot signal storage circuit 321 is further added, and the transmit signal processing circuits 811-1 to 811-N.sub.SDM are changed to the transmit signal processing circuits 311-1 to 311-N.sub.SDM.

(77) In other words, while performing the equivalent operation as the transmitter according to NPL 1, the transmitter 72a of FIG. 6 changes the processing for assigning the pilot signal at both ends of the effective bandwidth W to the transmit signal to the processing for assigning the pilot signal for each signal sequence to be spatially multiplexed, instead of performing for each antenna system. Furthermore, the difference corresponds to the change that, in the transmit signal processing circuit according to NPL 2, a signal allocated with the user data is generated within the effective bandwidth W illustrated in FIG. 25, whereas in FIG. 26, a signal allocated with the user data is generated within the reduced effective bandwidth W″ illustrated in FIG. 25. However, while the multiplication processing of the transmission weight is performed in the transmit signal processing circuits 311-1 to 311-N.sub.SDM in FIG. 2, the multiplication processing of the transmission weight is not performed in the transmit signal processing circuits 411-1 to 411-N.sub.SDM in FIG. 6. The processing corresponding to the multiplication processing of the transmission weight is performed by the time axis transmission weight multiplication circuits 761-1 to 761-N.sub.SDM as a feature of the technique according to NPL 1.

(78) In the above description, the assignment of the pilot signal at both ends of the effective bandwidth is performed at the adders 420-1 to 420-N.sub.SDM between the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM and the time axis transmission weight multiplication circuits 761-1 to 761-N.sub.SDM. In contrast, in the same manner as in the first embodiment, a configuration may be adopted in which a signal allocated a pilot signal to the subcarriers on both ends of the effective bandwidth is generated in the transmit signal processing circuits 411-1 to 411-N.sub.SDM. In this case, the adders 420-1 to 420-N.sub.SDM and the pilot signal storage circuit 321 are unnecessary, and equivalent processing is performed in the transmit signal processing circuits 411-1 to 411-N.sub.SDM.

(79) Note that, in the second embodiment, a configuration is taken in which the signals of the N.sub.SDM systems output from the time axis transmission weight multiplication circuits 761-1 to 761-N.sub.SDM are synthesized in the addition synthesis circuits 812-1 to 812-N.sub.Ant, and the following D/A converters 814-1 to 814-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant are used together. In contrast, the signals may be implemented individually from the following D/A converters 814-1 to 814-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant without being synthesized at the addition synthesis circuits 812-1 to 812-N.sub.Ant, and a subarray may be configured by the antenna elements 819-1 to 819-N.sub.Ant in each of the antennas.

(80) FIG. 7 is a schematic block diagram illustrating an example of a configuration of the receiver 76a of the wireless station apparatus 70 according to the second embodiment. As illustrated in FIG. 7, the receiver 76a includes: antenna elements 851-1 to 851-N.sub.Ant; low noise amplifiers 852-1 to 852-N.sub.Ant; a local oscillator 853; mixers 854-1 to 854-N.sub.Ant; filters 855-1 to 855-N.sub.Ant; A/D converters 856-1 to 856-N.sub.Ant; extended FFT circuits 157-1 to 157-N.sub.SDM; receive signal processing circuits 445-1 to 445-N.sub.SDM; a reception weight processing unit 744; time axis reception weight multiplication circuits 755-1 to 755-N.sub.SDM; and a time axis reception weight calculation circuit 757. The receive signal processing circuits 445-1 to 445-N.sub.SDM, the reception weight processing unit 744, and the reception weight calculation circuit 747 are connected to the communication control circuit 52. The reception weight processing unit 744 includes a channel information estimation circuit 746 and a reception weight calculation circuit 747.

(81) The difference between the configurations of the receivers described in FIG. 7 and NPL 1 is that the FFT circuits 257-1 to 257-N.sub.SDM are changed to the extended FFT circuits 157-1 to 157-N.sub.SDM, and that the receive signal processing circuits 745-1 to 745-N.sub.SDM are changed to the receive signal processing circuits 445-1 to 445-N.sub.SDM.

(82) This is similar to the first embodiment in that the FFT circuits 257-1 to 257-N.sub.SDM are changed to the extended FFT circuits 157-1 to 157-N.sub.SDM, and the receive signal processing circuits 845-1 to 845-N.sub.SDM are further changed to the receive signal processing circuits 345-1 to 345-N.sub.SDM.

(83) In other words, the difference corresponds to the change that the receiver 76a in FIG. 7 adds a function of performing phase noise compensation by using the pilot signal at both ends of the effective bandwidth W on the transmit signal to the FFT circuits 257-1 to 257-N.sub.SDM (corresponding to the extended FFT circuits 157-1 to 157-N.sub.SDM) while performing the equivalent operation as the receiver 66a in FIG. 21, and in the receive signal processing circuits 745-1 to 745-N.sub.SDM of FIG. 21, the receive signal processing is performed on the signal allocated with the user data within the effective bandwidth W illustrated in FIG. 23, whereas in FIG. 7, the receive signal processing is performed on the signal allocated with the user data within the reduced effective bandwidth W″ illustrated in FIG. 25. Furthermore, other features are the same as those described with respect to the receiver 66a according to NPL 1 illustrated in FIG. 21.

(84) According to the wireless station apparatus 70 according to the second embodiment configured as described above, the same effects as those of the first embodiment can be obtained in other configurations.

Third Embodiment

(85) In the background art described in NPL 1, the time axis transmission weight multiplication circuits 761-1 to 761-N.sub.SDM digitally multiply the time axis transmission weight, and the time axis reception weight multiplication circuits 755-1 to 755-N.sub.SDM digitally multiply the time axis reception weight. However, the multiplication of the transmission and/or reception weight in the time domain corresponds to multiplying the entire frequency band by a common coefficient, and is equivalent to the process of rotating the phase by using a phase shifter directly to analog transmit signals or analog receive signals. Using this feature, in NPL 2, a method is adopted in which the calculation of the transmission and/or reception weight to be digitally multiplied is performed by the approach described in NPL 1, while processing corresponding to the multiplication of the actual time axis transmission and/or reception weight is performed in a phase shifter. In the following description, first, a description will be given of another circuit configuration in a case that the technique described in NPL 2 is applied as a background art of the present invention, and then the third embodiment will be described as a difference with it.

(86) An embodiment in a case that the present invention is applied to the technique described in NPL 2 will be described below.

(87) Circuit Configuration According to Third Embodiment

(88) In the third embodiment of the present invention, the configuration of the wireless station apparatus takes an equivalent configuration as the wireless station apparatus 70 illustrated in FIG. 1. The difference from FIG. 1 is only that the transmitter 71 is replaced with a transmitter 72b, the receiver 75 is replaced with a receiver 76b, and the communication control circuit 51 is replaced with a communication control circuit 53, so that the overall functions and features are in accordance with FIG. 1, and the details of the drawings and descriptions are omitted herein.

(89) FIG. 8 is a diagram illustrating a schematic block diagram illustrating an example of a configuration of the transmitter 72b of the wireless station apparatus 70 according to the third embodiment. As illustrated in FIG. 8, the transmitter 72b includes: transmit signal processing circuits 411-1 to 411-N.sub.SDM; IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM; D/A converters 414-1 to 414-N.sub.SDM; a local oscillator 815; mixers 316-1 to 316-N.sub.SDM: filters 317-1 to 317-N.sub.SDM; high power amplifiers 818-1 to 818-N.sub.Ant; antenna elements 819-1 to 819-N.sub.Ant; synthesizers 671-1 to 671-N.sub.Ant; phase shifter groups 681-1 to 681-N.sub.SDM; distributors 673-1 to 673-N.sub.SDM; a phase control circuit 688; a time axis transmission weight calculation circuit 642; and a pilot signal storage circuit 321. The transmit signal processing circuits 411-1 to 411-N.sub.SDM and the time axis transmission weight calculation circuit 642 are connected to the communication control circuit 53.

(90) The difference from FIG. 22 is that the D/A converters 314-1 to 314-N.sub.SDM are changed to the D/A converters 414-1 to 414-N.sub.SDM, the pilot signal storage circuit 321 is further added, and that the transmit signal processing circuits 711-1 to 711-N.sub.SDM are changed to the transmit signal processing circuits 411-1 to 411-N.sub.SDM. The D/A converters 414-1 to 414-N.sub.SDM have functions to add a signal in the time domain from the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM and a signal in the time domain from the pilot signal storage circuit 321 for each sampling data, and performs D/A conversion on the added value. This is equivalent to a configuration in which the adders 420-1 to 420-N.sub.SDM are disposed after the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM (and prior to the D/A converters 314-1 to 314-N.sub.SDM) similar to FIG. 8, and here the signal in the time domain from the pilot signal storage circuit 321 is added for each sampling data.

(91) This is similar to the first embodiment in that the difference between FIG. 18 and FIG. 2 is that the adders 320-1 to 320-N.sub.Ant are added between the IFFT & GI assigning circuits 813-1 to 813-N.sub.Ant and the D/A converters 814-1 to 814-N.sub.Ant, the pilot signal storage circuit 321 is further added, and the transmit signal processing circuits 811-1 to 811-N.sub.SDM are changed to the transmit signal processing circuits 311-1 to 311-N.sub.SDM.

(92) In other words, while performing the equivalent operation as the transmitter 62b in FIG. 22, the transmitter 72b of FIG. 8 changes the processing for assigning the pilot signal at both ends of the effective bandwidth W to the transmit signal to the processing for assigning the pilot signal for each signal sequence to be spatially multiplexed, instead of performing for each antenna system. Furthermore, the difference corresponds to the change that, in the transmit signal processing circuits 711-1 to 711-N.sub.SDM of FIG. 22, a signal allocated with the user data is generated within the effective bandwidth W illustrated in FIG. 23, whereas in FIG. 8, a signal allocated with the user data is generated within the reduced effective bandwidth W″ illustrated in FIG. 25.

(93) In the above description, the assignment of the pilot signal at both ends of the effective bandwidth is performed at the D/A converters 414-1 to 414-N.sub.SDM. In contrast, in the same manner as the supplemental description of the second embodiment, a configuration may be adopted in which a signal allocated a pilot signal to the subcarriers on both ends of the effective bandwidth is generated in the transmit signal processing circuits 411-1 to 411-N.sub.SDM. In this case, while the pilot signal storage circuit 321 is omitted, the D/A converters 414-1 to 414-N.sub.SDM are replaced with the D/A converters 314-1 to 314-N.sub.SDM that do not have an addition function with the pilot signal, and equivalent processing is performed in the transmit signal processing circuits 411-1 to 411-N.sub.SDM.

(94) Note that, in the third embodiment, in the same manner as the background art illustrated in FIG. 22, a configuration is taken in which N.sub.SDM systems from the distributors 673-1 to 673-N.sub.SDM to the phase shifter groups 681-1 to 681-N.sub.SDM are synthesized by the synthesizers 671-1 to 671-N.sub.Ant, and the high power amplifiers 818-1 to 818-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant are used together for each signal sequence. In contrast, the signals may be implemented individually from the following high power amplifiers 818-1 to 818-N.sub.Ant to the antenna elements 819-1 to 819-N.sub.Ant without being synthesized by the synthesizers 671-1 to 671-N.sub.Ant, and a subarray may be configured by the antenna elements 819-1 to 819-N.sub.Ant in each of the antennas. Furthermore, other features are the same as those described with respect to the transmitter 62b of the background art illustrated in FIG. 22.

(95) FIG. 9 is a diagram illustrating a schematic block diagram illustrating an example of a configuration of the receiver 76b of the wireless station apparatus 70 according to the third embodiment. As illustrated in FIG. 9, the receiver 76b includes: antenna elements 851-1 to 851-N.sub.Ant; low noise amplifiers 852-1 to 852-N.sub.Ant; a local oscillator 853; mixers 254-1 to 254-N.sub.SDM; filters 255-1 to 255-N.sub.SDM; A/D converters 256-1 to 256-N.sub.SDM; extended FFT circuits 157-1 to 157-N.sub.SDM; receive signal processing circuits 445-1 to 445-N.sub.SDM; a reception weight processing unit 744; distributors 672-1 to 672-N.sub.Ant; phase shifter groups 682-1 to 682-N.sub.SDM; synthesizers 674-1 to 674-N.sub.SDM; a time axis reception weight calculation circuit 657; and a phase control circuit 678. The receive signal processing circuits 445-1 to 445-N.sub.SDM, the reception weight processing unit 744, and the time axis reception weight calculation circuit 657 are connected to the communication control circuit 53. The reception weight processing unit 744 includes a channel information estimation circuit 746 and a reception weight calculation circuit 747.

(96) The difference from FIG. 24 is that the FFT circuits 257-1 to 257-N.sub.SDM are changed to the extended FFT circuits 157-1 to 157-N.sub.SDM, and that the receive signal processing circuits 745-1 to 745-N.sub.SDM are changed to the receive signal processing circuits 445-1 to 445-N.sub.SDM. This is similar to the second embodiment in that the FFT circuits 257-1 to 257-N.sub.SDM are changed to the extended FFT circuits 157-1 to 157-N.sub.SDM, and the receive signal processing circuits 845-1 to 845-N.sub.SDM are further changed to the receive signal processing circuits 445-1 to 445-N.sub.SDM when the receiver 66a illustrated in FIG. 21 of the background art is changed to the receiver 76a illustrated in FIG. 7 of the second embodiment.

(97) In other words, the difference corresponds to the change that the receiver 76b in FIG. 9 adds a function of performing phase noise compensation by using the pilot signal at both ends of the effective bandwidth W on the transmit signal to the FFT circuits 257-1 to 257-N.sub.SDM (corresponding to the extended FFT circuits 157-1 to 157-N.sub.SDM) while performing the equivalent operation as the receiver 66b in FIG. 24, and in the receive signal processing circuits 745-1 to 745-N.sub.SDM of FIG. 24, the receive signal processing is performed on the signal allocated with the user data within the effective bandwidth W illustrated in FIG. 23, whereas in FIG. 9, the receive signal processing is performed on the signal allocated with the user data within the reduced effective bandwidth W″ illustrated in FIG. 25. Furthermore, other features are the same as those described with respect to the receiver 66b according to NPL 2 illustrated in FIG. 24.

(98) According to the wireless station apparatus 70 according to the third embodiment configured as described above, the same effects as those of the first embodiment can be obtained in other configurations.

Fourth Embodiment

(99) In the first to third embodiments of the present invention, the configurations of the extended FFT circuits 357 and 157 in the receiver uses the configuration illustrated in FIG. 4(B). Here, the input signal to the FFT circuit 857 is performed in a form that includes the pilot signal at both ends of the effective bandwidth W.
However, because the FFT circuit 857 and the receive signal processing circuits 345-1 to 345-N.sub.SDM or the receive signal processing circuits 445-1 to 445-N.sub.SDM do not require the pilot signal at both ends of the effective bandwidth W, it is possible to exclude the pilot signal on both ends of the effective bandwidth W before inputting into the FFT circuit 857. The configuration of the extended FFT circuit according to the fourth embodiment with this function is illustrated in FIG. 10.

(100) As illustrated in FIG. 10, the extended FFT circuit 191 includes an FFT circuit 182, a function ϕ (t) acquisition circuit 183, an IFFT circuit 184, an IFFT circuit 187, a phase noise compensation circuit 185, a γ.sub.k setting circuit 193, a distance L setting circuit 194, a pilot signal removal circuit 186, and an FFT circuit 857.

(101) The difference from FIG. 4 is that the input to the phase noise compensation circuit 185 is a signal from which the pilot signal at both ends of the effective bandwidth W Chas been removed by the pilot signal removal circuit 186 and the IFFT circuit 187, rather than the input signal to the extended FFT circuit 190 duplicated by the replication circuit 181 itself. The operation of the pilot signal removal circuit 186 is to insert zero into the components of the subcarrier of the region 917 and the region 198 in FIG. 3 when the signal converted to the signal in the frequency domain is input by the FFT circuit 182. This signal is converted from the signal in the frequency domain to the signal in the time domain by the IFFT circuit 187. As a result, by canceling not only the pilot signal at both ends of the effective bandwidth W, but also the pilot signal including the components leaked to the neighboring subcarriers due to the influence of the phase noise, and by returning the signal in the frequency domain canceled to the signal in the time domain, it is possible to perform signal processing on data including of the effective region including the user data within the reduced effective bandwidth W″ (precisely, within the band also including some subcarriers where signals have been leaked from the reduced effective bandwidth W″) at the receive signal processing circuits 345-1 to 345-N.sub.SDM or the receive signal processing circuits 445-1 to 445-N.sub.SDM.

(102) Similarly, corresponding to FIG. 5, the configuration of FIG. 11 can be used. The difference from FIG. 5 is similar to FIG. 10 in that the input to the phase noise compensation circuit 185 is a signal from which the pilot signal at both ends of the effective bandwidth W Chas been removed by the pilot signal removal circuit 186 and the IFFT circuit 187, rather than the input signal to the extended FFT circuit 190 duplicated by the replication circuit 181 itself.

(103) According to the wireless station apparatus 70 according to the fourth embodiment configured as described above, the same effects as those of the first embodiment can be obtained in other configurations.

Fifth Embodiment

(104) In the description of the first to fourth embodiments, the case that the OFDM modulation scheme is applied is illustrated as a representative example, but the same process can be applied to a system of single carrier transmission. Systems of single carrier transmission include a SC-FDE scheme that performs equalization processing in the frequency domain, and the like, in addition to a common single carrier transmission. In the SC-FDE scheme among them, a common SC-FDE processing may be performed instead of signal processing of the OFDM modulation scheme in the receive signal processing circuits 345-1 to 345-N.sub.SDM or the receive signal processing circuits 445-1 to 445-N.sub.SDM, for involving signal processing of the frequency domain. These are common techniques, and thus detailed description thereof is omitted. Similarly, the signal processing of the OFDM modulation scheme may be changed to a common SC-FDE processing on the transmitter side. For example, by generating a normal signal of a single carrier in the time domain in the transmit signal processing circuits 311-1 to 311-N.sub.SDM or the transmit signal processing circuits 411-1 to 411-N.sub.SDM, and by implementing a circuit configured to assign GI without performing IFFT in the part of the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM, the processing can correspond to single carrier transmission.

(105) However, in a case of a pure single carrier transmission rather than the SC-FDE scheme, it is not necessary to convert the signal in the time domain into the signal in the frequency domain by using the FFT circuit. Thus, the FFT circuit 857 implemented in the extended FFT circuit 357 is not needed, so that the extended FFT circuit will be replaced by a single carrier compensation circuit, and the assignment of GI in the part of the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM is also not needed. FIG. 12 is a diagram illustrating a configuration example of a single carrier compensation circuit according to the fifth embodiment.

(106) As illustrated in FIG. 12, the single carrier compensation circuit 192 includes an FFT circuit 182, a function ϕ (t) acquisition circuit 183, an IFFT circuit 184, a pilot signal removal circuit 186, an IFFT circuit 187, a phase noise compensation circuit 189, a γ.sub.k setting circuit 193, a distance L setting circuit 194, a ψ (t) acquisition circuit 195, and a function ϕ (t) acquisition circuit 196. The difference from FIG. 10 is that the FFT circuit 857 is omitted and the phase noise compensation circuit 185 is replaced by phase noise compensation circuit 189. The difference between the phase noise compensation circuit 185 and the phase noise compensation circuit 189 is that the output is changed from one system to N.sub.SDM systems of the same contents while having the same function. As illustrated in the fourth embodiment, the output signal from the phase noise compensation circuit 189 does not include the pilot signal at both ends of the effective bandwidth W, and a single carrier signal compensated for phase noise is output.

(107) Similarly, corresponding to FIG. 11, the configuration of FIG. 13 can be used. The difference from FIG. 11 is similar to the difference between FIG. 10 and FIG. 12 in that the FFT circuit 857 is omitted and the phase noise compensation circuit 185 is replaced by phase noise compensation circuit 189.

(108) A configuration example of a transmitter and a receiver in single carrier transmission according to the fifth embodiment is illustrated in FIGS. 14 and 15 in a manner that reflects the changes described above. In the fifth embodiment of the present invention, the configuration of the wireless station apparatus takes an equivalent configuration as the wireless station apparatus 70 illustrated in FIG. 1. The difference from FIG. 1 is only that the transmitter 71 is replaced with a transmitter 74, the receiver 75 is replaced with a receiver 78, and the communication control circuit 51 is replaced with a communication control circuit 54, so that the overall functions and features are in accordance with FIG. 1, and the details of the drawings and descriptions are omitted herein. Hereinafter, a configuration example of the transmitter 74 and the receiver 78 is illustrated.

(109) As illustrated in FIG. 14, the transmitter 74 according to the fifth embodiment include: transmit signal processing circuits 511-1 to 511-N.sub.SDM; D/A converters 314-1 to 314-N.sub.SDM; a local oscillator 815, mixers 316-1 to 316-N.sub.SDM; filters 317-1 to 317-N.sub.SDM; high power amplifiers 818-1 to 818-N.sub.Ant; antenna elements 819-1 to 819-N.sub.Ant; synthesizers 671-1 to 671-N.sub.Ant; phase shifter groups 682-1 to 682-N.sub.SDM; distributors 673-1 to 673-N.sub.SDM; a phase control circuit 688; a time axis transmission weight calculation circuit 642; adders 420-1 to 420-N.sub.SDM; and a pilot signal storage circuit 321. The transmit signal processing circuits 511-1 to 511-N.sub.SDM and the time axis transmission weight calculation circuit 642 are connected to the communication control circuit 54.

(110) The difference from FIG. 8 is that the IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM are omitted, and the transmit signal processing circuits 411-1 to 411-N.sub.SDM configured to perform processing of the frequency domain such as an OFDM modulation scheme are replaced with the transmit signal processing circuits 511-1 to 511-N.sub.SDM configured to perform signal processing of single carrier transmission in the time domain. Although the D/A converters 414-1 to 414-N.sub.SDM are described in divided portions as the D/A converters 314-1 to 314-N.sub.SDM and the adders 420-1 to 420-N.sub.SDM, this is only describing a functionally equivalent circuit explicitly in divided portions, and essentially there is no difference.

(111) Thus, it is changed from generating a transmit signal in the time domain of the digital baseband using an OFDM modulation scheme in combination of the transmit signal processing circuits 411-1 to 411-N.sub.SDM and IFFT & GI assigning circuits 313-1 to 313-N.sub.SDM, to generating a transmit signal in the time domain of the digital baseband for single carrier transmission by the transmit signal processing circuits 511-1 to 511-N.sub.SDM. Signal processing of single carrier transmission performed by the transmit signal processing circuits 511-1 to 511-N.sub.SDM is a common signal processing, and the signal such as a unique word is assigned as an overhead in advance of a signal that is subjected to modulation processing on the user data, although the details are omitted herein. The signal in the time domain including these will be output.

(112) Note that, in the fifth embodiment as well, similar to the third embodiment, the high power amplifiers 818-1 to 818-N.sub.Ant, the antenna elements 819-1 to 819-N.sub.Ant, or the like may be implemented individually for each signal sequence spatially multiplexed, and a subarray may be configured for each of the signal sequences. Furthermore, other features are the same as those described with respect to the third embodiment.

(113) As illustrated in FIG. 15, the receiver 78 according to the fifth embodiment includes: antenna elements 851-1 to 851-N.sub.Ant; low noise amplifiers 852-1 to 852-N.sub.Ant; a local oscillator 853; mixers 254-1 to 254-N.sub.SDM; filters 255-1 to 255-N.sub.SDM; A/D converters 256-1 to 256-N.sub.SDM; single carrier compensation circuits 557-1 to 557-N.sub.SDM (the SC compensation circuit in FIG. 15); receive signal processing circuits 545-1 to 545-N.sub.SDM; a reception weight processing unit 744; a time axis reception weight calculation circuit 757, distributors 672-1 to 672-N.sub.Ant; phase shifter groups 682-1 to 682-N.sub.SDM; and synthesizers 674-1 to 674-N.sub.SDM. The receive signal processing circuits 545-1 to 545-N.sub.SDM, the reception weight processing unit 744, and the time axis reception weight calculation circuit 757 are connected to the communication control circuit 54. The reception weight processing unit 744 includes a channel information estimation circuit 746 and a reception weight calculation circuit 747.

(114) The difference from FIG. 9 is that the extended FFT circuits 157-1 to 157-N.sub.SDM are changed to the single carrier compensation circuits 557-1 to 557-N.sub.SDM, and the receive signal processing circuits 445-1 to 445-N.sub.SDM configured to perform processing of the frequency domain such as the OFDM modulation scheme are changed to the receive signal processing circuits 545-1 to 545-N.sub.SDM configured to perform processing of the time domain of single carrier transmission.

(115) Thus, it is changed from performing the signal detection processing of the time domain of the digital baseband using the OFDM modulation scheme by the receive signal processing circuits 445-1 to 445-N.sub.SDM, to performing the signal detection processing of the time domain of the digital baseband in single carrier transmission by the receive signal processing circuits 545-1 to 545-N.sub.SDM. Signal processing of single carrier transmission performed by the receive signal processing circuits 545-1 to 545-N.sub.SDM is a common signal processing, and the signal such as a unique word assigned in advance of a signal that is subjected to modulation processing on the user data is detected, the signal detection processing is started at an appropriate timing, and the signal on the transmitting side is reproduced and output to the MAC layer processing circuit 68 side, although the details are omitted herein.

(116) The signal processing performed by the single carrier compensation circuits 557-1 to 557-N.sub.SDM is as described above, which performs the compensation processing of the phase noise, and outputs a single carrier signal within the reduced effective bandwidth W″ from which the pilot signal has been removed to the receive signal processing circuits 545-1 to 545-N.sub.SDM.

(117) Note that, in the fifth embodiment as well, similar to the third embodiment, low noise amplifiers 852-1 to 852-N.sub.Ant, antenna elements 851-1 to 851-N.sub.Ant, or the like may be implemented individually for each signal sequence spatially multiplexed, and a subarray may be configured for each of the signal sequences. Furthermore, other features are the same as those described with respect to the third embodiment.

(118) According to the wireless station apparatus 70 according to the fifth embodiment configured as described above, the same effects as those of the first embodiment can be obtained even in a configuration in which single carrier transmission is performed.

Sixth Embodiment

(119) In the first to fifth embodiments, in the transmitters 71, 72a, 72b, and 74 of each of the wireless station apparatuses 70, the antenna elements 819-1 to 819-N.sub.Ant, the high power amplifiers 818-1 to 818-N.sub.Ant, and the like may be implemented individually for each signal sequence spatially multiplexed, and a subarray may be configured for each of the signal sequences. Similarly, in the receivers 75, 76a, 76b, and 78, the antenna elements 851-1 to 851-N.sub.Ant, the low noise amplifiers 852-1 to 852-N.sub.Ant, and the like may be implemented individually for each signal sequence spatially multiplexed, and a subarray may be configured for each of the signal sequences. In a case that the subarray antennas of the transmitters 71, 72a, 72b, and 74 and the receivers 75, 76a, 76b, and 78 are physically installed, for example, at positions separated by several m, each of the local oscillator 815 and the local oscillator 853 may be individually implemented, and different phase noises may be added between each of the subarrays.

(120) FIG. 16 is a diagram illustrating an overview of a counterpart wireless station apparatus 70 in the sixth embodiment. As illustrated in FIG. 16, the wireless station apparatuses 70a and 70b include transmitters 71a and 71b, receivers 75a and 75b, interface circuits 67a and 67b, MAC layer processing circuits 68a and 68b, and communication control circuits 51a and 51b. The wireless station apparatus 70b takes the subarray configuration as described above, and the transmitter 71b includes subarrays 92-1 to 92-N.sub.SDM, and the receiver 75b includes subarrays 91-1 to 91-N.sub.SDM. The wireless station apparatus 70a includes antennas used together without taking a subarray configuration, and forms transmit directivity beams 93-1 to 93-N.sub.SDM by multiplication of the transmission weight to face the subarrays 91-1 to 91-N.sub.SDM, and forms receive directivity beams 94-1 to 94-N.sub.SDM similarly by multiplication of the reception weight to face the subarrays 92-1 to 92-N.sub.SDM.

(121) For example, as described with reference to the third embodiment, processing is performed in which transmit directivity beams 93-1 to 93-N.sub.SDM are formed in the synthesizers 671-1 to 671-N.sub.Ant, the phase shifter groups 681-1 to 681-N.sub.SDM, and the distributors 673-1 to 673-N.sub.SDM in the transmitter 72b of FIG. 8 corresponding to the transmitter 71a, and receive directivity beams 94-1 to 94-N.sub.SDM are formed in the distributors 672-1 to 672-N.sub.Ant, the phase shifter groups 682-1 to 682-N.sub.SDM, and the synthesizers 674-1 to 674-N.sub.SDM in the receiver 76b of FIG. 9 corresponding to the receiver 75a.

(122) At this time, because the receiver 75a and 75b are applied with the local oscillator 815 and/or the local oscillator 853 different for each signal sequence to be spatially multiplexed, individual independent phase noises are added. At this time, for example, with the third embodiment as an example, the interference components are expected to be suppressed to some extent in the input signal to each of the extended FFT circuits 157-1 to 157-N.sub.SDM because the mutual signal separation is generally made by the above-described directivity formation. In this case, in a case where the above-described processing is performed individually on each of the extended FFT circuits 157-1 to 157-N.sub.SDM, appropriate phase noise compensation can be performed in each signal sequence.

(123) Similarly, even in a multi-user MIMO environment in which one base station apparatus performs spatial multiplex transmission with a plurality of terminal station apparatuses, the same phase noise compensation may be performed by each of the individual extended FFT circuits 157-1 to 157-N.sub.SDM because the mutual signal separation is generally made by the above-described directivity formation. Here, in a case that the number of users of the multi-user MIMO is two, it is possible to operate to further reduce the mutual interference by allocating one of the pilot signals at both ends of the effective bandwidth W to the user #1, and the other to the user #2. Otherwise, while taking a subarray configuration, in a case that the local oscillator 815 and/or the local oscillator 853 are used together between subarrays, it is expected that the phase noise added to each signal sequence will be completely equivalent phase noise. In FIG. 1, a method of placing a pilot signal on both ends of the effective bandwidth W and averaging information obtained from a plurality of pilot signals to increase the extraction accuracy of the phase noise is as described above, but similar averaging processing can be performed across the extended FFT circuits 157-1 to 157-N.sub.SDM. In this case, signal lines are required to exchange the mutual information between the extended FFT circuits 157-1 to 157-N.sub.SDM.

(124) In this way, the embodiments of the present invention can be applied even in a case of taking a subarray configuration or a case of performing multi-user MIMO transmission in addition to single user MIMO.

(125) Although the embodiments of the present invention have been described above with reference to the drawings, it is clear that the above embodiments are merely examples of the present invention, and the present invention is not limited to the embodiments described above. For example, in the descriptions of the embodiments of the present invention, the apparatus configurations and signal processing are mainly described assuming the OFDM modulation scheme, the single carrier transmission scheme, the SC-FDE scheme, and the like, but in order to apply other schemes, an apparatus configuration of the scheme according to related art may be reflected in the embodiments of the present invention. Although the present invention has been described in the context of a general Point-to-Point type of spatial multiplex transmission without clearly expressing in the present invention, the same discussion holds true even in a case of a Point-to-Multipoint type of communication configuration equipped with a plurality of wireless station apparatuses. Furthermore, at this time, the configuration can also be extended to a configuration in which communication is performed concurrently in parallel with a plurality of wireless station apparatuses by the multi-user MIMO transmission.

(126) The processing of the assignment on the transmitting side of the pilot signal used in the embodiment of the present invention may be achieved by separately adding the information (sampling data) in the time domain of the pilot signal stored in the memory to the transmit signal generated by any section. In a case that signal processing in the frequency domain is involved on the transmitting side, such as an OFDM modulation scheme, an SC-FDE scheme, or the like, it can be managed by allocating the components of the pilot signals in the frequency domain, and collectively generating the pilot signals. In this sense, the pilot signal is essentially assigned to the transmit signal, and the manner in which the pilot signal is assigned can be achieved by various variations.

(127) While terms such as frequency domain, time domain, time axis, and the like are used herein. the digital sampling data can be converted into signals of a plurality of frequency components or subcarriers by FFT processing performed when applying an OFDM modulation scheme or a SC-FDE scheme, for example, and the signal of each frequency component or each subcarrier is referred to herein as the signal in the frequency domain (or frequency axis). In contrast to this, the digital sampling data is a signal of time sequence, which is referred to as a signal in the time domain (or time axis), and in this sense, because the analog signal is a temporally continuous signal, and the rotation processing of the complex phase performed by the phase shifter, for example, in the third embodiment or the like, corresponds to signal processing in the time domain. Basically, signal processing in the time domain does not require FFT processing unlike the signal processing in the frequency domain, and thus basically has a feature of being easy to avoid the problem of signal leakage to adjacent frequency components. Thus, even in a case where other signal processing of the time domain not described herein may be added additionally, the present invention can be operated without any influence.

(128) Here, the descriptions have been made mainly of the embodiments provided with a plurality of antenna elements, assuming a case that phase noise compensation is performed for spatial multiplex transmissions that cannot be managed by the known background arts. However, the essence of the present invention is a feature that a pilot signal in the band (the feature is not to allocate signal in the frequency domain around the pilot signal) is assigned on the transmitting side, a replica of phase noise is generated on the receiving side by using the pilot signal and the frequency components around the pilot signal, and the phase noise is removed from the receive signal by using these, as illustrated in FIG. 25. In this sense, it is not necessary to use a plurality of antenna elements, and even in a case that spatial multiplex transmission is not used, phase noise compensation can be performed by the application of the present invention. At this time, in the embodiments of the present invention, the pilot signal is disposed at both ends of the effective bandwidth, but this is in order to actively utilize the guard band originally with no signal allocation, and the pilot signal is not necessary disposed at both ends of the effective bandwidth, but it is possible to allocate a pilot signal to any location within the band. Further, although an example of assigning two pilot signals is illustrated in the embodiments of the present invention, it is possible to utilize only one or three or more of the pilot signals. In particular, in a case that a plurality of wireless stations and one wireless station (base station apparatus) perform spatial multiplex transmissions at the same time, or in a case that a wireless station using a subarray configuration performs spatial multiplex transmission between subarrays, a configuration may be adopted in which each of the signal sequences with independent phase noise is allocated an individual pilot subcarrier, and the wireless stations other than the wireless station does not perform signal transmission to the pilot subcarrier. In this case, phase noise compensation is performed individually for each signal sequence corresponding to the individual wireless stations or subarrays on the receiving station side. Thus, it is also within the scope of the present invention to appropriately modify and implement the individual parameters described in the embodiments of the present invention.

(129) Thus, addition, omission, substitution, and other modifications of the constituent components may be made without departing from the spirit and scope of the present invention.

REFERENCE SIGNS LIST

(130) 51, 51a, 51b, 52, 53 Communication control circuit 60 Wireless station apparatus 61, 62 Transmitter 65, 66 Receiver 67, 67a, 67b Interface circuit 68, 68a, 68b MAC layer processing circuit 70 Wireless station apparatus 71, 71a, 71b, 72a, 72b, 74 Transmitter 75, 75a, 75b, 76a, 76b, 78 Receiver 157-1 to 157-N.sub.SDM Extended FFT circuit 181 Replication circuit 182 FFT circuit 183 Function Φ (t) acquisition circuit 184 IFFT circuit 185 Phase noise compensation circuit 186 Pilot signal removal circuit 187 IFFT circuit 189 Phase noise compensation circuit 191 Extended FFT circuit 193 γ.sub.k setting circuit 194 Distance L setting circuit 195 Function ψ (t) acquisition circuit 196 Function Φ (t) acquisition circuit 192 Single carrier compensation circuit 254-1 to 254-N.sub.SDM Mixer 255-1 to 255-N.sub.SDM Filter 256-1 to 256-N.sub.SDM A/D (analog to digital) converter 257-1 to 257-N.sub.SDM FFT circuit 311-1 to 311-N.sub.SDM Transmit signal processing circuit 313-1 to 313-N.sub.SDM IFFT & GI assigning circuit 314-1 to 314-N.sub.SDM D/A converter 316-1 to 316-N.sub.SDM Mixer 317-1 to 317-N.sub.SDM Filter 320-1 to 320-N.sub.Ant Adder 321 Pilot signal storage circuit 345-1 to 345-N.sub.SDM Receive signal processing circuit 357-1 to 357-N.sub.Ant Extended FFT circuit 411-1 to 411-N.sub.SDM Transmit signal processing circuit 414-1 to 414-N.sub.SDM D/A converter 420-1 to 420-N.sub.SDM Adder 445-1 to 445-N.sub.SDM Receive signal processing circuit 511-1 to 511-N.sub.SDM Transmit signal processing circuit 642 Time axis transmission weight calculation circuit 657 Time axis reception weight calculation circuit 671-1 to 671-N.sub.Ant Synthesizer 672-1 to 672-N.sub.Ant Distributor 673-1 to 673-N.sub.SDM Distributor 674-1 to 674-N.sub.SDM Synthesizer 678 Phase control circuit 681-1 to 681-N.sub.SDM Phase shifter group 682-1 to 682-N.sub.SDM Phase shifter group 688 Phase control circuit 740 Transmission weight processing unit 741 Channel information acquisition circuit 742 Channel information storage circuit 743 Transmission weight calculation circuit 744 Reception weight processing unit 746 Channel information estimation circuit 747 Reception weight calculation circuit 755-1 to 755-N.sub.SDM Time axis reception weight multiplication circuit 757 Time axis transmission weight calculation circuit 761-1 to 761-N.sub.SDM Time axis transmission weight multiplication circuit 812-1 to 812-N.sub.Ant Addition synthesis circuit 813-1 to 813-N.sub.Ant IFFT & GI assigning circuit 814-1 to 814-N.sub.Ant D/A converter 815 Local oscillator 816-1 to 816-N.sub.Ant Mixer 817-1 to 817-N.sub.Ant Filter 818-1 to 818-N.sub.Ant High power amplifier (HPA) 819-1 to 819-N.sub.Ant Antenna element 840 Transmission weight processing unit 841 Channel information acquisition circuit 842 Channel information storage circuit 843 Transmission weight calculation circuit 844 Reception weight processing unit 846 Channel information estimation circuit 847 Reception weight calculation circuit 851-1 to 851-N.sub.Ant Antenna element 852-1 to 852-N.sub.Ant Low noise amplifier (LNA) 853 Local oscillator 854-1 to 854-N.sub.Ant Mixer 855-1 to 855-N.sub.Ant Filter 856-1 to 856-N.sub.Ant A/D (analog to digital) converter 857-1 to 857-N.sub.Ant FFT circuit 887 Local oscillator