Passive mixer with reduced second order intermodulation

Abstract

The present disclosure generally relates to the field of receiver structures in radio communication systems and more specifically to passive mixers in the receiver structure and to a technique for converting a first signal having a first frequency into a second signal having a second frequency by using a third signal having a third frequency. A passive mixer for converting a first signal having a first frequency into a second signal having a second frequency by using a third signal having a third frequency comprises a cancellation component 220 for generating a first cancellation signal for cancelling second order intermodulation components by superimposing the first signal weighted by a cancellation value on the third signal; and a mixing component 231 having a first terminal 232 for receiving the first signal, a second terminal 234 for outputting the second signal, and a third terminal 236 for receiving the first cancellation signal, wherein the mixing component 231 is adapted to provide the second signal as output at the second terminal 234 by mixing the first signal provided as input at the first terminal 232 and the first cancellation signal provided as input at the third terminal 236.

Claims

1. A passive mixer, not comprising an active component biased by an external power source which is not an input signal, adapted to convert a first signal having a first frequency into a second signal having a second frequency by using a third signal having a third frequency, comprising: a first cancellation component adapted to generate a first cancellation signal operative to substantially cancel second order intermodulation components by adding the first signal weighted by a first cancellation value on the third signal; a mixing component having a first terminal adapted to receive the first signal, a second terminal adapted to output the second signal, and a third terminal adapted to receive the first cancellation signal, wherein the mixing component is adapted to provide the second signal as output at the second terminal by mixing the first signal provided as input at the first terminal and the first cancellation signal provided as input at the third terminal; and a first active sensing component adapted to sense a voltage at the first terminal and to adapt the first cancellation value based on the sensed voltage at the first terminal.

2. The passive mixer according to claim 1, wherein the mixing component comprises a pair of two complementary voltage controlled switches, wherein the two complementary voltage controlled switches are connected in parallel and share the first terminal, and each have a distinct third terminal.

3. The passive mixer according to claim 1, wherein the mixing component comprises a field effect transistor switch having its drain operatively connected to the first terminal, its gate operatively connected to the third terminal and its source operatively connected to the second terminal.

4. The passive mixer according to claim 1, wherein the mixing component comprises a field effect transistor switch having its source operatively connected to the first terminal, its gate operatively connected to the third terminal and its drain operatively connected to the second terminal.

5. The passive mixer according to claim 1, further comprising a second sensing component adapted to sense the voltage at the second terminal, wherein the first cancellation component is adapted to generate the first cancellation signal by additionally considering the sensed voltage at the second terminal.

6. The passive mixer according to claim 1, further comprising two or more mixing components and a corresponding number of cancellation components, and a more-phase generator adapted to generate the third signal with two or more different phases and to individually feed the different phases of the third signal into one or more of the two or more mixing components.

7. The passive mixer according to claim 6, wherein the first cancellation component is adapted to generate the first cancellation signal by superimposing the first signal weighted by the cancellation value on one phase of the third signal and by superimposing the first signal weighted by the same or an adapted cancellation value on another phase of the third signal.

8. The passive mixer according to claim 6, wherein the more-phase generator comprises a four-phase generator, and wherein the cancellation components may be weighted with the same or different cancellation values.

9. The passive mixer according to claim 1, wherein the first signal is a radio frequency signal, the third signal is a local oscillator signal, and the second signal is one of an intermediate frequency signal and a baseband signal.

10. The passive mixer according to claim 1, wherein the first signal is one of an intermediate frequency signal and a baseband signal, the third signal is a local oscillator signal, and the second signal is a radio frequency signal.

11. A transceiver apparatus comprising a transmitter adapted to transmit a radio frequency transmit signal and a receiver adapted to receive a radio frequency receive signal, wherein the receiver comprises: a low noise amplifier adapted to amplify the high frequency receive signal; and a passive mixer, not comprising an active component biased by an external power source which is not an input signal, comprising: a local oscillator adapted to generate a local oscillator signal; a cancellation component adapted to generate a first cancellation signal to substantially cancel second order intermodulation components by superimposing the amplified radio frequency receive signal weighted by a cancellation value on the local oscillator signal; a mixing component having a first terminal adapted to receive the amplified radio frequency receive signal, a second terminal adapted to output one of an intermediate frequency signal and a baseband signal, and a third terminal adapted to receive the first cancellation signal, wherein the mixing component is adapted to provide one of the intermediate frequency signal and the baseband signal as output at the second terminal by mixing the amplified radio frequency receive signal provided as input at the first terminal and the first cancellation signal provided as input at the third terminal; and a first active sensing component adapted to sense a voltage at the first terminal and to adapt the first cancellation value based on the sensed voltage at the first terminal.

12. The transceiver apparatus according to claim 11, wherein the receiver further comprises one of a bandpass filter and a lowpass filter connected to the second terminal, wherein the bandpass filter has a passband of a predetermined frequency range adapted to filter the intermediate frequency signal and the lowpass filter has a passband of a predetermined frequency range adapted to filter the baseband signal.

13. A transceiver apparatus comprising a transmitter adapted to transmit a radio frequency transmit signal and a receiver adapted to receive a radio frequency receive signal, wherein the transmitter comprises: an amplifier adapted to amplify one of an intermediate frequency signal and a baseband signal to be transmitted; and a passive mixer, not comprising an active component biased by an external power source which is not an input signal, comprising: a local oscillator adapted to generate a local oscillator signal; a cancellation component adapted to generate a first cancellation signal to substantially cancel second order intermodulation components by superimposing the amplified intermediate frequency signal or baseband signal weighted by a cancellation value on the local oscillator signal; a mixing component having a first terminal adapted to receive the amplified intermediate frequency signal or baseband signal, a second terminal adapted to output a radio frequency transmit signal, and a third terminal adapted to receive the first cancellation signal, wherein the mixing component is adapted to provide the radio frequency transmit signal as output at the second terminal by mixing the amplified intermediate frequency signal or baseband signal provided as input at the first terminal and the first cancellation signal provided as input at the third terminal; and a first active sensing component adapted to sense a voltage at the first terminal and to adapt the first cancellation value based on the sensed voltage at the first terminal.

14. A method, performed by a passive mixer not comprising an active component biased by an external power source which is not an input signal, of converting a first signal having a first frequency into a second signal having a second frequency by using a third signal having a third frequency, comprising: sensing, by an active sensing component, a voltage at a first terminal of a mixing component; adapting, by the active sensing component, a first cancellation value based on the sensed voltage at the first terminal; generating, by a cancellation component, a first cancellation signal to substantially cancel second order intermodulation components by superimposing the first signal weighted by the cancellation value on the third signal; receiving, at a first terminal of a mixing component, the first signal; receiving, at a third terminal of the mixing component, the first cancellation signal; and outputting, at a second terminal of the mixing component, the second signal by mixing the first signal provided as input at the first terminal and the first cancellation signal provided as input at the third terminal.

15. The method of claim 14, wherein a voltage at the second terminal is sensed by a second sensing component and the first cancellation signal is generated by additionally considering the sensed voltage at the second terminal.

16. A passive mixer, not comprising an active component biased by an external power source which is not an input signal, adapted to convert a first signal having a first frequency into a second signal having a second frequency by using a third signal having a third frequency, comprising: a first cancellation component adapted to generate a first cancellation signal operative to substantially cancel second order intermodulation components by adding the first signal weighted by a predetermined cancellation value on the third signal; and a mixing component having a first terminal adapted to receive the first signal, a second terminal adapted to output the second signal, and a third terminal adapted to receive the first cancellation signal, wherein the mixing component is adapted to provide the second signal as output at the second terminal by mixing the first signal provided as input at the first terminal and the first cancellation signal provided as input at the third terminal.

17. A method, performed by a passive mixer not comprising an active component biased by an external power source which is not an input signal, of converting a first signal having a first frequency into a second signal having a second frequency by using a third signal having a third frequency, comprising: generating, by a cancellation component, a first cancellation signal to substantially cancel second order intermodulation components by superimposing the first signal weighted by a predetermined cancellation value on the third signal; receiving, at a first terminal of a mixing component, the first signal; receiving, at a third terminal of the mixing component, the first cancellation signal; and outputting, at a second terminal of the mixing component, the second signal by mixing the first signal provided as input at the first terminal and the first cancellation signal provided as input at the third terminal.

Description

BRIEF DESCRIPTION OF THE DRAWINGS

(1) In the following, the invention will further be described with reference to exemplary embodiments illustrated in the figures, in which:

(2) FIG. 1 is a block diagram schematically illustrating a transceiver apparatus embodiment;

(3) FIG. 2 is a schematic illustration of a receiver of the transceiver embodiment of FIG. 1;

(4) FIG. 3 is a schematic illustration of a first passive mixer embodiment of the receiver shown in FIG. 2;

(5) FIG. 4 is a schematic illustration of a second passive mixer embodiment of the receiver shown in FIG. 2;

(6) FIG. 5 is a flow chart illustrating a first method embodiment;

(7) FIG. 6 is a schematic illustration of a third passive mixer embodiment of the receiver shown in FIG. 2;

(8) FIG. 7 is a flow chart illustrating a second method embodiment;

(9) FIG. 8 is a schematic illustration of a fourth passive mixer embodiment of the receiver shown in FIG. 2;

(10) FIG. 9 is a schematic illustration of a fifth passive mixer embodiment of the receiver shown in FIG. 2;

(11) FIG. 10 is a schematic illustration of a sixth passive mixer embodiment of the receiver shown in FIG. 2;

(12) FIG. 11 is a schematic illustration of a seventh passive mixer embodiment of the receiver shown in FIG. 2;

(13) FIG. 12 is a schematic illustration of an eighth passive mixer embodiment of a receiver shown in FIG. 2;

(14) FIG. 13 is a schematic illustration of a ninth passive mixer embodiment of a receiver shown in FIG. 2;

(15) FIG. 14 is a schematic illustration of a current-mode passive mixer embodiment of a transmitter shown in FIG. 1; and

(16) FIG. 15 is a schematic illustration of a voltage-mode passive mixer embodiment of a transmitter shown in FIG. 1.

DETAILED DESCRIPTION

(17) In the following description, for purposes of explanation and not limitation, specific details are set forth, such as specific circuitries including particular components, elements etc., in order to provide a thorough understanding of the present invention. It will be apparent to one skilled in the art that the present invention may be practiced in other embodiments that depart from these specific details. For example, the skilled person will appreciate that the present invention, although explained below with respect to a Metal Oxide Semiconductor (MOS) Field Effect Transistor (FET), may make use of other transistors like a Junction Field Effect Transistor (JFET), a Metal Semiconductor Field Effect Transistor (MESFET), an Insulated Gate Bipolar Transistor (IGBT) or the like. For example, the invention may make use of n-channel MOSFETs, p-channel MOSFETs, n-channel JFETs or p-channel JFETs.

(18) Those skilled in the art will further appreciate that functions explained hereinbelow may be implemented using individual hardware circuitry and/or using an application specific integrated circuit (ASIC). The ASIC may be built from Field-programmable gate arrays (FPGAs), programmable logic devices (PLDs), like complex programmable logic devices (CPLDs), or any other standard parts known to those skilled in the art. It will also be appreciated that when the present invention is described as a method, this method may also be embodied on the ASIC.

(19) FIG. 1 shows a block diagram of a radio frequency (RF) transceiver apparatus 110 for use in a mobile communication device 100. As schematically illustrated in FIG. 1, the mobile communication device 100 comprising the RF transceiver apparatus 110 is adapted to transmit a radio frequency transmit signal 102 from an antenna 112 and is adapted to receive a radio frequency receive signal 104 with the antenna 112. The mobile terminal comprises a duplexer 114 connected to the RF transceiver apparatus 110 via an impedance matching network 116 and a power amplifier 118, so that the transceiver apparatus 110 can be used for both transmitting the radio frequency transmit signal by means of a transmitter 130 and receiving the radio frequency receive signal by means of a receiver 120. The impedance matching network 116 is connected to the receiver 120 of the RF transceiver apparatus 110 and the power amplifier 118 is connected to the transmitter 130 of the RF transceiver apparatus 110.

(20) The transmitter 130 of the RF transceiver apparatus 110 comprises a driver amplifier 310, a mixer 330, a local oscillator (LO) 340 and a baseband (BB) filter 350. The receiver 120 of the RF transceiver apparatus 110 comprises a low noise amplifier (LNA) 210, a mixer 230, a local oscillator (LO) 240 and a baseband (BB) filter 250. When the RF transceiver apparatus 110 is used in transmit mode, a data signal is passed to the BB filter 350, filtered by the BB filter 350 and passed to the mixer 330, where the BB data signal is up-converted into an RF signal using an LO signal generated by the LO 340. The RF signal is then passed to the driver amplifier 310, the power amplifier 118, the duplexer 114 and finally to the antenna 112 for transmitting the RF transmit signal 102.

(21) When the transceiver apparatus 110 is used in receive mode, an RF receive signal 104 is received by the antenna 112, is passed by the duplexer 114 to the impedance matching network 116 and then to the receiver 120 of the RF transceiver apparatus 110. In the receiver 120, the LNA 210 amplifies the RF receive signal, the mixer 230 directly down-converts the amplified RF receive signal into a BB signal by mixing the amplified RF receive signal with an LO signal generated by the LO 240 and then passes the BB signal to the BB filter 250 for further BB filtering and amplification.

(22) FIG. 2 illustrates the receiver 120 of the RF transceiver apparatus 110 shown in FIG. 1 comprising the LNA 210 for amplifying the RF receive signal, the mixer 230 for down-converting the amplified RF receive signal into the BB signal by using the LO signal generated by the LO 240 and the BB amplifier 250 for filtering and amplifying the down-converted BB signal.

(23) FIG. 3 schematically illustrates a first passive mixer 230 embodiment of the receiver 120 shown in FIG. 2. As shown in FIG. 3, the passive mixer 230 comprises a MOSFET 231 as a mixing component having a drain, a gate and a source (the bulk is grounded for ease of simplicity), the drain being operatively connected to a first terminal 232, the source being operatively connected to a second terminal 234 and the gate being operatively connected to a third terminal 236 of the passive mixer 230. The first terminal 232 is adapted to receive the RF receive signal (amplified by the LNA 210), the second terminal 234 is adapted to output the BB signal and the third terminal is adapted to receive a third signal, a first cancellation signal, the generation of which will be described in more detail below.

(24) The current through an N-channel MOSFET in its linear region, i.e. when 0<V.sub.ds<V.sub.gs−V.sub.th, can to a first order be given as

(25) $\begin{array}{cc}{I}_{\mathrm{ds}}=\beta \times V_{\mathrm{ds}}\times \left(V_{\mathrm{gs}}-V_{\mathrm{th}}-\frac{V_{\mathrm{ds}}}{2}\right)& \left(1\right)\end{array}$
where I.sub.ds is the drain-source current, β is a geometry dependent constant, V.sub.gs is the gate-source voltage, V.sub.ds is the drain-source voltage and V.sub.th is the MOSFET threshold voltage. Here, an N-channel device has been assumed but similar relations can easily be derived for P-channel devices. Assuming, without loss of generality, that the source and the second terminal 234 are grounded, there will be two scenarios depending on the polarity of V.sub.ds, namely the first scenario for V.sub.ds≧0 and the second scenario for V.sub.ds<0.

(26) For V.sub.ds≧0, the second terminal 234 will act as the source, i.e. the voltage at the source V.sub.s will be equal to zero (grounded) (V.sub.s=0), and the first terminal 232 will act as the drain, i.e. the voltage at the drain V.sub.d will be equal to the voltage of the RF signal V.sub.rf (V.sub.d=V.sub.rf). For V.sub.ds<0, the drain and source are swapped (V.sub.d=0 and V.sub.s=V.sub.rf).

(27) For the first scenario (V.sub.ds≧0), the drain-source current I.sub.ds becomes by means of equation (1)

(28) $\begin{array}{cc}{I}_{\mathrm{ds}}=\beta \times V_{\mathrm{rf}}\times \left(V_{\mathrm{lo}}-V_{\mathrm{th}}-\frac{V_{\mathrm{rf}}}{2}\right)& \left(2\right)\end{array}$
and for the second scenario (V.sub.ds<0), the drain-source current I.sub.ds becomes by means of equation (1)

(29) $\begin{array}{cc}\begin{array}{c}{I}_{\mathrm{ds}}=-\beta \times V_{\mathrm{rf}}\times \left(V_{\mathrm{lo}}-V_{\mathrm{rf}}-V_{\mathrm{th}}+\frac{V_{\mathrm{rf}}}{2}\right)\\ =-\beta \times V_{\mathrm{rf}}\times \left(V_{\mathrm{lo}}-V_{\mathrm{th}}-\frac{V_{\mathrm{rf}}}{2}\right)\end{array}& \left(3\right)\end{array}$
where V.sub.lo is the voltage of the LO signal which is equal to the gate-source voltage V.sub.gs since the source is grounded. The sign reversal in the drain-source current I.sub.ds reflects the change in reference direction due to the terminal swapping.

(30) As can be seen from equations (2) and (3), in both scenarios, there is one linear current component (V.sub.rf×V.sub.lo) and one second order modulation (IM2) component (V.sub.rf×V.sub.rf/2). Since the MOSFET switch primarily works in the linear region, the above equations (2) and (3) describe the main influence of the nonlinear channel conductance on the mixer current.

(31) By superimposing a fraction of the RF signal on the LO signal by weighting the RF signal with a cancellation value α, i.e. when the gate voltage V.sub.g becomes
V.sub.g=V.sub.lo+α×V.sub.rf  (4)
the cancellation of the IM2 term can be achieved by choosing the cancellation value α appropriately. Since the source is grounded V.sub.g will equal V.sub.gs. As shown above, the IM2 component is proportional to V.sub.rf*V.sub.rf/2.

(32) Thus, by selecting α=½ the IM2 component can be cancelled as equation (1) then yields in combination with equation (4)

(33) $\begin{array}{cc}\begin{array}{c}{I}_{\mathrm{ds}}=\beta \times V_{\mathrm{ds}}\times \left(V_{\mathrm{gs}}-V_{\mathrm{th}}-\frac{V_{\mathrm{ds}}}{2}\right)\\ =\beta \times V_{\mathrm{rf}}\times \left(V_{\mathrm{lo}}+\frac{V_{\mathrm{rf}}}{2}-V_{\mathrm{th}}-\frac{V_{\mathrm{rf}}}{2}\right)\\ =\beta \times V_{\mathrm{rf}}\times \left(V_{\mathrm{lo}}-V_{\mathrm{th}}\right)\end{array}& \left(1\right)+\left(4\right)\end{array}$
which is now proportional to V.sub.rf, i.e. is now linear, when V.sub.lo and V.sub.th can be considered constant.

(34) Thus, by merely setting the cancellation value α=½, by weighting the RF signal with the cancellation value α and by superimposing (adding) the weighted RF signal on (to) the LO signal, the IM2 component can be canceled.

(35) The latter is exemplarily shown in FIG. 3, where the RF signal is weighted by the amplifier 222 with the cancellation value α (by scaling the drive strength of the first sense amplifier 222 in relation to the LO) and is then added to the LO signal in order to generate the first cancellation signal at the cancellation component 220. The first cancellation signal is then provided to the third terminal 236 (connected to the gate), the RF signal is provided to the first terminal 232 (connected to the drain) and the BB signal is generated as output at the second terminal 234 (connected to the source) by mixing the RF signal and the first cancellation signal.

(36) The above will cancel the IM2 due to the MOSFET 231 switch channel conductance, which covers most of the switch conduction angle. At the switching threshold, the MOSFET 231 will start in the sub-threshold region and will enter the saturation region as soon as any significant current starts to flow through the MOSFET 231. The drain-source current I.sub.ds in the saturation region can be described as

(37) $\begin{array}{cc}{I}_{\mathrm{ds}}=\frac{\beta }{2}\times {\left(V_{\mathrm{gs}}-V_{\mathrm{th}}\right)}^2.& \left(5\right)\end{array}$

(38) For V.sub.ds≧0, equation (5) yields

(39) $\begin{array}{cc}{I}_{\mathrm{ds}}=\frac{\beta }{2}\times {\left(V_{\mathrm{lo}}-V_{\mathrm{th}}\right)}^2& \left(6\right)\end{array}$
which is proportional to the square of V.sub.lo and for V.sub.ds<0, equation (5) yields

(40) $\begin{array}{cc}{I}_{\mathrm{ds}}=\frac{\beta }{2}\times {\left(V_{\mathrm{lo}}-V_{\mathrm{rf}}-V_{\mathrm{th}}\right)}^2& \left(7\right)\end{array}$
which also has an IM2 term proportional to the square of V.sub.rf.

(41) In the sub-threshold region, the drain-source current I.sub.ds is much smaller and the characteristic is exponential also contributing with some IM2.

(42) When the cancellation value α is selected to deviate slightly from the linear cancellation criterion, i.e. the cancellation value α would be selected to not equal 0.5, the IM2 generated in the sub-threshold region and the saturation region can be compensated by allowing some residual IM2 in the linear region. In other words, the cancellation value α can be tuned such that it nulls the sum of all IM2 contributions but does not null all individual IM2 components separately, e.g. the one in the linear region.

(43) As shown in FIG. 3, the cancellation component 220 is adapted to generate the first cancellation signal for cancelling IM2 components by superimposing the RF receive signal weighted by the cancellation value α on the LO signal. Alternatively to setting the cancellation value α to a fixed value, the first sense amplifier 222 can be used in order to sense the voltage at the first terminal 232. By sensing the voltage at the first terminal 232 the appropriate cancellation value for cancelling the IM2 component can be determined by evaluating equation (1). The unweighted RF receive signal is provided to the first terminal 232 of the MOSFET 231 and the first cancellation signal is provided to the third terminal 236 of the MOSFET 231. By mixing the amplified RF receive signal with the first cancellation signal, the MOSFET 231 switch outputs a BB signal at its source and thus at the second terminal 234 of the mixing component. The BB signal is then filtered and amplified by the BB amplifier 250 comprising, for example as shown in FIG. 3, an amplifier 252, a resistor 254 and a capacitor 256, wherein the resistor 254 and the capacitor are operatively connected to the input and the output of the amplifier 252 for feedback control.

(44) FIG. 4 shows a second mixer embodiment of the receiver shown in FIG. 2, which is used in voltage mode. In voltage mode, the source of the MOSFET 231 and thus the second terminal 234 are not directly connected to virtual ground but are connected via an impedance, e.g. as shown in FIG. 4 a capacitor 258, to ground, i.e. the MOSFET 231 is loaded by the capacitor 258. In contrast to the first passive mixer embodiment in FIG. 3, the voltage at the second terminal 234 is not close to zero since the second terminal 234 is not connected to virtual ground. Therefore, the voltage at the second terminal 234 has to be considered in equations (1) to (4) in order to determine the drain-source voltage V.sub.ds and the gate-source voltage V.sub.gs. The drain-source voltage V.sub.ds is unlike in the first embodiment not merely equal to the RF voltage V.sub.rf but is equal to the difference between the RF voltage V.sub.rf at the first terminal 232 and the IF voltage V.sub.s at the second terminal 234. Likewise, the gate-source voltage V.sub.gs (without any additional cancellation signal) is not merely equal to the voltage of the LO signal V.sub.lo but is equal to the difference between the voltage of the LO signal and the source voltage V.sub.s (the voltage at the second terminal 234). Thus, in order to select the cancellation value α, both the drain voltage V.sub.d at the first terminal 232 and the source voltage V.sub.s at the second terminal 234 have to be sensed in order to determine the drain-source voltage V.sub.ds and the gate-source voltage V.sub.gs of the MOSFET 231.

(45) Since the RF voltage at the first terminal 232 and the IF voltage at the second terminal 234 are widely separated in frequency, they can be sensed independently. In order to sense the voltage at the first terminal 232, the first sense amplifier 222 is used and in order to sense the voltage at the second terminal 234, a second sense amplifier 224 is used. Then, the sensed voltage at the first terminal 232 and the sensed voltage at the second terminal 234 are used to adapt the cancellation value α. The RF signal is weighted by the cancellation value α.sub.1 and the IF signal is weighted by the cancellation value α.sub.2 and the weighted signals are provided to the cancellation component 220. The cancellation values α.sub.1 and α.sub.2 may be the same for simplicity or individually set to maximize performance. At the cancellation component 220, the LO signal is superimposed on both the weighted RF signal and the weighted IF signal, in order to generate a first cancellation signal which is then provided to the third terminal 236 and the gate of the MOSFET 231. The output of the second terminal 234 is then again provided to the BB amplifier 252 for amplification and filtering.

(46) The two passive mixer embodiments described above with respect to FIGS. 3 and 4 are further illustrated by the flow chart of FIG. 5 which shows a first method embodiment 300. The method 300 comprises the following steps: Generate first cancellation signal (step 302); Receive first signal at first terminal (step 304); Receive first cancelation signal at third terminal (step 306); and Output second signal at second terminal by mixing first signal and first cancellation signal (step 308).

(47) FIG. 6 shows a third passive mixer embodiment for cancelling second order intermodulation components. The passive mixer is operated in current mode, like in the first passive mixer embodiment of FIG. 3, but in contrast to the first passive mixer embodiment shown in FIG. 3 (and also to the second passive mixer embodiment shown in FIGS. 4), the bulk terminal of the MOSFET 231, which is operatively connected to a fourth terminal 238 of the passive mixer, is additionally considered. When additionally considering the bulk terminal 238, the threshold voltage V.sub.th is proportional to the bulk-source voltage V.sub.bs
V.sub.th=V.sub.th0−γ×V.sub.bs  (8)
where V.sub.th is the unmodulated threshold voltage and V.sub.bs is the bulk-source voltage. By injecting (superimposing) a suitably scaled second cancellation signal at the bulk terminal 238, unwanted IM2 components can also be suppressed. When V.sub.bs=α×V.sub.rf, α×γ=½ and V.sub.g=V.sub.lo (since the source and the second terminal 234 are grounded), the drain-source current I.sub.ds yields using equation (1)

(48) $\begin{array}{cc}\begin{array}{c}{I}_{\mathrm{ds}}=\beta \times V_{\mathrm{rf}}\times (\left(V_{\mathrm{lo}}-\left(V_{\mathrm{th}0}-\gamma \times \alpha \times V_{\mathrm{rf}}\right)-\frac{V_{\mathrm{rf}}}{2}\right)\\ =\beta \times V_{\mathrm{rf}}\times \left(V_{\mathrm{lo}}-V_{\mathrm{th}0}\right)\end{array}& \left(9\right)\end{array}$

(49) Thus, the IM2 term is again cancelled with the above assumptions.

(50) Since, in practice, V.sub.th is a complex nonlinear function of the bulk, source and drain voltages, the above linearized model is thus not exact, but provides a good estimation. Also because of the small moderate and weak inversion conduction angles, the cancellation criterion α×γ can be selected to slightly deviate from 0.5 in order to minimize the aggregate IM2 components.

(51) As shown in FIG. 6, the amplified RF receive signal is provided to the drain of the MOSFET 231 and the first sense amplifier 242 is adapted to sense the voltage of the RF signal at the first terminal 232 in order to the select the appropriate cancellation value. The RF receive signal is weighted with the determined cancellation value and is superimposed on a bias voltage at the cancellation component 240 in order to generate the second cancellation signal. The second cancellation signal is provided to the bulk terminal of the MOSFET 231. The MOSFET 231 switch is adapted to mix the amplified RF receive signal provided as input at the drain and the local oscillator signal provided as input at the gate together with the second cancellation signal provided as input at the bulk of the MOSFET 231 in order to generate and output a BB signal at the source. The BB signal is then amplified by the BB amplifier 250.

(52) The third passive mixer embodiment described above with respect to FIG. 6 is further illustrated by the flow chart of FIG. 7 which shows a second method embodiment 400. The method 400 comprises the following steps: Generate second cancellation signal (step 402); Receive first signal at first terminal (step 404); Receive third signal at third terminal (step 408); Receive second cancelation signal at fourth terminal (step 410); and Output second signal at second terminal by mixing first signal and third signal together with second cancellation signal (step 412).

(53) The first passive mixer embodiment of FIG. 3 and the third passive mixer embodiment of FIG. 6 can be combined to a fourth passive mixer embodiment as shown in FIG. 8. In this embodiment, the voltage at the first terminal 232 is sensed by two distinct sense amplifiers 222, 242, a first one 222 connected to the first terminal 232 and the third terminal 236 (via a first cancellation component 220) and a second one 242 connected to the first terminal 232 and the fourth terminal 238 (via a second cancellation component 240). In this way two distinct cancellation values can be determined by the two sense amplifiers 222, 242, i.e. a first cancellation value determined by the first sense amplifier 222 and a second cancellation value determined by the second sense amplifier 242. Then, a first cancellation signal is determined by adding the RF signal weighted by the first cancellation value to the LO signal and a second cancellation signal is determined by adding the RF signal weighted by the second cancellation value on a bias voltage. As shown in FIG. 8, the first cancellation signal is input to the gate of the MOSFET 231 via the third terminal 236 and the second cancellation signal is input to the bulk of the MOSFET 231 via the fourth terminal 238. By mixing the first signal provided as input at the drain and the first cancellation signal provided as input at the gate together with the second cancellation signal provided as input at the bulk, a BB signal is provided at the source of the MOSFET 231 and the second terminal 234. The BB signal is finally filtered and amplified by a BB amplifier 250.

(54) In the fifth passive mixer embodiment of FIG. 9, like in the fourth embodiment of FIG. 8, also an additional sense amplifier 244 is added to the third passive mixer embodiment of FIG. 6. Unlike the fourth embodiment shown in FIG. 8, the additional (second) sense amplifier 244 is connected between the second terminal 234 and the second cancellation component 240 rather than to the first terminal 232 and the first cancellation component 220. In this way, the fifth passive mixer embodiment shown in FIG. 9 operates in voltage mode. The RF receive signal is weighted with a first cancellation value (determined by the first sense amplifier 242 by sensing the voltage at the first terminal 232) and, in addition thereto, the voltage (BB voltage) at the second terminal 234 is sensed by the second sense amplifier 244 so that the BB signal is weighted by a second cancellation value (determined by the second sense amplifier 244 by sensing the voltage at the second terminal 234). Both the RF signal weighted by the first cancellation value and the IF signal (BB signal) weighted by the second cancellation value are added to a bias voltage at the cancellation component 240 in order to generate the second cancellation signal. The second cancellation signal is provided to the bulk terminal of the MOSFET 231. The MOSFET 231 switch is adapted to mix the amplified RF receive signal provided as input at the drain and the local oscillator signal provided as input at the gate together with the second cancellation signal provided as input at the bulk of the MOSFET 231 in order to generate and output a BB signal at the source. The BB signal is then amplified by the BB amplifier 250

(55) In the sixth passive mixer embodiment shown in FIG. 10, a further sense amplifier 224 is added to the fifth passive mixer embodiment of FIG. 9. The second cancellation signal which is provided to the bulk of the MOSFET 231, is determined in the way as described with respect to FIG. 9 above. In addition thereto, the further sense amplifier 224 is adapted to sense the voltage at the second terminal (like the sense amplifier 244). Although the sense amplifier 244 and the sense amplifier 224 are both adapted to sense the voltage at the second terminal 234, the BB signal can be weighted with different cancellation values by the two amplifiers 224, 244. The BB signal weighted by one cancellation value is supplied to the second cancellation component 240 (to generate the second cancellation signal) and the BB signal weighted by the same or a different cancellation value is supplied to the first cancellation component 220 to be added to the LO signal in order to generate the first cancellation signal. By mixing the amplified RF receive signal provided as input at the drain and the first cancellation signal provided as input at the gate together with the second cancellation signal provided as input at the bulk of the MOSFET 231, a BB signal is generated and output at the source and finally filtered and amplified by a BB amplifier 250.

(56) The seventh passive mixer embodiment shown in FIG. 11 is a combination of the previously described first to sixth passive mixer embodiments. In this embodiment, a first cancellation signal is generated by superimposing the RF signal weighted by a cancellation value and the BB signal weighted by a cancellation value on the LO signal at the first cancellation component 220 and a second cancellation signal is generated by superimposing the RF signal weighted by a cancellation value and the BB signal weighted by a cancellation value on the bulk voltage at the second cancellation component 240. The first and second cancellation signals are then provided to the respective terminals of the MOSFET 231 (the gate and bulk) in order to generate the BB signal at the source as described above with respect to the first to sixth embodiments.

(57) An eighth passive mixer embodiment is shown in FIG. 12. According to this embodiment, a more-phase generator, exemplarily shown as a 4-phase generator 270 in FIG. 12, is provided. The 4-phase generator comprises two opposing signal sources in order to generate different phases of the LO signal received by the 4-phase generator 270 as input. The RF receive signal (amplified by the LNA 210) is provided at the drain of four MOSFETs 231 shown in FIG. 12. In the same manner as described above, the voltage of the RF signal (the voltage at the drain of the four MOSFETs 230) is sensed by the first sense amplifier 222 and the RF receive signal is weighted with a cancellation value by the first sense amplifier 222. The weighted RF receive signal is then added to the different phases generated by the 4-phase generator 270 at the respective cancellation components 220. The RF signals applied to the respective cancellation components 220 may be weighted with the same or with different cancellation values. If required by the operating conditions (mismatch, process speed, temperature and the like), the RF signal can be weighted with a first cancellation value and can be added (at a first cancellation component 220) to the first phase of the LO signal generated by the 4-phase generator 270 and the RF signal can be weighted by a second cancellation value (different from the first cancellation value) and can be added (at a second cancellation component 220) to the second phase of the LO signal generated by the 4-phase generator and so on. In this way the appropriate cancellation signals are determined.

(58) The RF signal is then mixed by the four MOSFETs 231 with the respective cancellation signals in the same manner as described above to output BB (or IF) signals at the source of the MOSFETs 231 which are finally filtered by the BB amplifiers 250 in order to generate I and Q quadrature components.

(59) The ninth passive mixer embodiment illustrated in FIG. 13 differs from the eighth passive mixer embodiment of FIG. 12 in that the weighted RF signal is superimposed on the bias voltage of the MOSFETs 231 rather than on the different phases of the LO signal. The different phases of the LO signal are instead supplied to the gate of the MOSFETs 231. At each MOSFET 231 the RF signal provided as input at the drain is mixed with the respective phase of the LO signal provided as input at the gate together with the cancellation signal provided as input at the bulk, in order to generate the BB (or IF) signal.

(60) Although described herein primarily in the context of a receiver circuit, the IM2-suppressing passive mixer of the present invention is not limited to receiver implementations, but additionally finds utility in reducing the IM2, and consequently raising the IP2, in transmitter circuits. For example, FIG. 14 shows one embodiment of a passive mixer 330, deployed in the transmitter 130 shown in FIG. 1. As shown in FIG. 3, the passive mixer 330 comprises a MOSFET 331 as a mixing component having a drain, a gate and a source (the bulk is grounded for ease of simplicity), the drain being operatively connected to a first terminal 332, the source being operatively connected to a second terminal 334 and the gate being operatively connected to a third terminal 336 of the passive mixer 330. The first terminal 332 is adapted to receive a BB or IF signal (amplified by a BB or IF amplifier 310), the second terminal 334 is adapted to output the RF signal and the third terminal is adapted to receive a third signal, which is a first cancellation signal.

(61) In one embodiment, the first cancellation signal is generated by setting a cancellation value α=½ , by weighting the BB or IF signal with the cancellation value α and by superimposing (adding) the weighted BB or IF signal on (to) the LO signal. The first cancellation signal thus cancels the IM2 component, in a manner similar to that described with respect to the receiver circuit of FIG. 3.

(62) In FIG. 14, the BB or IF signal is weighted by the amplifier 322 with the cancellation value α (by scaling the drive strength of the first sense amplifier 322 in relation to the LO) and is then added to the LO signal in order to generate the first cancellation signal at the cancellation component 320. The first cancellation signal is then provided to the third terminal 336 (connected to the gate), the BB or IF signal is provided to the first terminal 332 (connected to the drain) and the RF signal is generated as output at the second terminal 334 (connected to the source) by mixing the BB or IF signal and the first cancellation signal.

(63) As described above with respect to the passive mixer in the receiver circuit 120 of FIG. 3, when the cancellation value α is selected to deviate slightly from the linear cancellation criterion, i.e. the cancellation value α would be selected to not equal 0.5, the IM2 generated in the sub-threshold region and the saturation region can be compensated by allowing some residual IM2 in the linear region. In other words, the cancellation value α can be tuned such that it nulls the sum of all IM2 contributions but does not null all individual IM2 components separately, e.g. the one in the linear region.

(64) As shown in FIG. 14, the cancellation component 320 is adapted to generate the first cancellation signal for cancelling IM2 components by superimposing the BB or IF signal weighted by the cancellation value α on the LO signal. In one embodiment, alternatively to setting the cancellation value α to a fixed value, the first sense amplifier 322 can be used in order to sense the voltage at the first terminal 332. By sensing the voltage at the first terminal 332 the appropriate cancellation value for cancelling the IM2 component can be determined by evaluating equation (1). The unweighted BB or IF signal is provided to the first terminal 332 of the MOSFET 331 and the first cancellation signal is provided to the third terminal 336 of the MOSFET 331. By mixing the amplified BB or IF signal with the first cancellation signal, the MOSFET 331 switch outputs an RF signal at its source and thus at the second terminal 334 of the mixing component. The RF signal is then filtered and amplified by the RF amplifier 350 comprising, for example as shown in FIG. 14, an amplifier 352 having a resistor 354 operatively connected to the input and the output of the amplifier 352 for feedback control.

(65) FIG. 15 shows an embodiment of a passive mixer 330, deployed in the transmitter 130 shown in FIG. 1, which is used in voltage mode. In voltage mode, the source of the MOSFET 331 and thus the second terminal 334 are not directly connected to virtual ground, but rather are connected via an impedance, e.g. as shown in FIG. 15, a capacitor 358, to ground. That is, the MOSFET 331 is loaded by the capacitor 358. In contrast to the passive mixer embodiment in FIG. 14, the voltage at the second terminal 334 is not close to zero, since the second terminal 334 is not connected to virtual ground. Therefore, the voltage at the second terminal 334 has to be considered in equations (1) to (4) in order to determine the drain-source voltage V.sub.ds and the gate-source voltage V.sub.gs. The drain-source voltage V.sub.ds is, unlike in the embodiment of FIG. 14, not merely equal to the RF voltage V.sub.rf, but it is equal to the difference between the BB or IF voltage V.sub.bb or V.sub.if at the first terminal 332 and the RF voltage V.sub.s, at the second terminal 334. Likewise, the gate-source voltage V.sub.gs(without any additional cancellation signal) is not merely equal to the voltage of the LO signal V.sub.lo but is equal to the difference between the voltage of the LO signal and the source voltage V.sub.s(the voltage at the second terminal 334). Thus, in order to select the cancellation value α, both the drain voltage V.sub.d at the first terminal 332 and the source voltage V.sub.s, at the second terminal 334 have to be sensed in order to determine the drain-source voltage V.sub.ds and the gate-source voltage V.sub.gsof the MOSFET 331.

(66) Since the BB or IF voltage at the first terminal 332 and the RF voltage at the second terminal 334 are widely separated in frequency, they can be sensed independently. In order to sense the voltage at the first terminal 332, the first sense amplifier 322 is used and in order to sense the voltage at the second terminal 334, a second sense amplifier 324 is used. Then, the sensed voltage at the first terminal 332 and the sensed voltage at the second terminal 334 are used to adapt the cancellation value α. The BB or IF signal is weighted by the cancellation value α.sub.1 and the RF signal is weighted by the cancellation value α.sub.2 and the weighted signals are provided to the cancellation component 230. The cancellation values α.sub.1 and α.sub.2 may be the same for simplicity or individually set to maximize performance. At the cancellation component 320, the LO signal is superimposed on both the weighted BB or IF signal and the weighted RF signal, in order to generate a first cancellation signal which is then provided to the third terminal 336 and the gate of the MOSFET 331. The output of the second terminal 334 is then again provided to the RF amplifier 352 for amplification and filtering.

(67) Those of skill in the art will readily recognize that all embodiments of the passive mixer described herein in the context of a receiver (e.g., as depicted in FIGS. 3-4, 6, 8-13) may similarly be advantageously deployed in a transmitter circuit to reduce IM2 and hence raise the IP2.

(68) The technique described above results in noticeable IM2 improvements on the order of 20 dB, in particular for the current mode mixer. This significant improvement leads to decreased duplexer requirements and therefore to lower costs, smaller size and smaller losses. For frequency bands where the duplexer Tx-Rx isolation is below the typically required 50 dB, the described technique eliminates the need for a SAW interstage filter between the LNA and the mixer, which also improves band flexibility and reduces the costs and size of the receiver structures and the devices.