METHOD FOR REDUCING INTERFERENCE EFFECTS IN A RADAR SYSTEM

Abstract

The invention describes a method for reducing interference effects in a radar system, which has at least two transceiver units (S1, S2), which are in particular spatially separated from one another, wherein the method comprises the following steps: —a transmission step (VS1), in which a first transmission signal (sigTX1) of the first transceiver unit (S1) is sent and received to and by a second transceiver unit (S2) and a second transmission signal (sigTX2) of the second transceiver unit (S2) is sent and received to and by the first transceiver unit (S1) via a radio channel (T), wherein the transmission signals (sigTX1, sigTX2) are modulated according to an orthogonal frequency multiplex method; and—a pre-correction step (VS2), in which correction values (γ1, γn, γ2) are determined from the received transmission signals (sigTX1, sigTX2) and in particular are exchanged between the transceiver stations (S1, S2), wherein the received transmission signals (sigRX1, sigRX2) are postprocessed on the basis of the correction values (γ1, γn, γ2), so that influences of interference variables, in particular of phase noise and/or a time offset and/or unknown initial phase positions, are reduced.

Claims

1. A method for reducing interference effects in a radar system, which has at least two transceiver units (S1, S2), which are in particular spatially separated from one another, wherein the method comprises the following steps: a transmission step (VS1), in which a first transmission signal (sigTX1) of the first transceiver unit (S1) is sent and received to and by a second transceiver unit (S2) and a second transmission signal (sigTX2) of the second transceiver unit (S2) is sent and received to and by the first transceiver unit (S1) via radio channel (t), wherein the transmission signals (sigTX1, sigTX2) are modulated according to an orthogonal frequency multiplex method; and a pre-correction step (VS2), in which correction values (γ1, γn, γ2) are determined from the received transmission signals (sigR1, sigRX2) and in particular are exchanged between the transceiver stations (S1, S2), wherein the received transmission signals (sigRX1, sigRX2) are postprocessed on the basis of the correction values (γ1, γn, γ2), so that influences of interference variables, in particular of phase noise and/or a time offset and/or unknown initial phase positions, are reduced.

2. The method of claim 1, which furthermore comprises a comparison step (VS3), in which: in the second transceiver unit (S2), a comparison signal (sigC21) is formed from a first corrected, received transmission signal (sigCOR1) and the second transmission signal (sigTX2), wherein the comparison signal (sigC21) is transmitted, in particular communicated, from the second transceiver unit (S2) to the first transceiver unit (S1); and in the first transceiver unit (S1), a comparison signal (sigC12) is formed from a second corrected, received transmission signal (sigCOR2) and the first transmission signal (sigTX1), wherein the comparison signal (sigC12) is transmitted, in particular communicated, from the first transceiver unit (S1) to the second transceiver unit (S2); wherein in a first step (VS31), deviations of the comparison signals (sigC21, sigC12), which are induced by systematic deviations in the transceiver units (S2, S1), are reduced, wherein in a second step (VS32), at least one complex value from a first of the two comparison signals or from a signal which was derived from this first comparison signal is used to adapt at least one complex value of the second of the two comparison signals or a value of a signal which was derived from this second comparison signal and thus to form an adapted signal (sigCC), wherein the adaptation is carried out in such a way that the vectorial sum or the difference of the complex values is formed by a mathematical operation or the sum or the difference of the phases of the complex values is formed.

3. The method of claim 1, wherein in the pre-correction step (VS2), in the transceiver unit (S1) and in the transceiver unit (S2), a frequency spectrum is calculated, in each of which a frequency peak (k1, k2) is determined, and preferably the frequency peaks (k1, k2) are exchanged between the stations, wherein the correction values (γ1, γn, γ2) are calculated on the basis of the frequency peaks (k1, k2).

4. The method of claim 3, wherein the correction values (γ1, γn, γ2) are calculated as follows: one correction value (γ1) as the division of the frequency peak k1 by the number of the subcarriers (N) and multiplication by pi; a further correction value (γn) for each subcarrier (n) as the difference of the frequency peaks (k1, k2), division by the number of the subcarriers (N), and multiplication by pi and a subcarrier number (n); and a time-dependent correction value (γ2) as the division of the present time by the symbol duration and multiplication by 2*pi.

5. The method of claim 1, which furthermore comprises a reconstruction step (VS4), in which distances and/or relative velocities and/or phase positions between the at least two transceiver units (S1, S2) are determined from the received transmission signals (sigRX1, sigRX2).

6. The method of claim 1, wherein in the reconstruction step (VS4), distances and/or relative velocities and/or phase positions and/or angles of passive objects are determined, wherein in particular further transceiver units (S3) communicate via transmission signals (sigTX3), which are modulated according to the OFDM method, with the at least two transceiver units (S1, S2).

7. The method of claim 1, wherein each of the transceiver units (S1, S2) sends and/or receives at least essentially simultaneously via at least one channel transmission signals (sigTX1, sigTX2) via antenna elements (A1, A2) of the transceiver units (S1, S2) designed for this purpose.

8. The method of claim 1, wherein in the transmission step (VS1), the first transceiver unit (S1) and the second transceiver unit have a line of sight connection to one another.

9. The method of claim 1, which furthermore comprises a synchronization step (VSSync), in which a time offset, a time drift, and/or a sending frequency of clock sources (S11, S21), which the transceiver units (S1, S2) have, is/are exchanged, in particular via radio.

10. The method of claim 1, wherein a simultaneous transmission of useful data takes place on subchannels and/or symbols provided for this purpose.

11. The method of claim 2, wherein only a certain number of send-receive channels is used, wherein in particular a compressed sensing method is used in the reconstruction step (VS3).

12. The method of claim 1, wherein the send-receive channels are assigned in a chronologically varying manner, in particular according to a stepped carrier method.

13. The method of claim 6, wherein the transceiver units (S1, S2), in particular during the transmission step (VS1), move in relation to one another and positions and alignments of the transceiver units (S1, S2) are detected at this time, wherein the reconstruction step (VS4) is carried out multiple times in succession, so that a radar image resolution of the reconstruction step (VS4) is increased by means of a synthetic aperture calculation.

14. The method of claim 5, wherein the transceiver units (S1, S2), in particular during the transmission step (VS1), move in relation to one another and positions and alignments of the transceiver units (S1, S2) are detected at this time, wherein the reconstruction step (VS3) is carried out multiple times in succession, so that an accuracy of the detected positions and/or alignments of the transceiver units (S1, S2) is increased by means of an inverse synthetic aperture calculation.

15. A radar system, in particular a secondary radar system, for determining a distance and/or a relative velocity, in particular for carrying out a method, comprising: at least two transceiver units (S1, S2), which are in particular spatially separated from one another and are designed to send and receive transmission signals (sigTX1, sigTX2), which are modulated according to an orthogonal frequency multiplex method, OFDM method; a signal processing unit, which is designed to determine correction values (γ1, γn, γ2) from the received transmission signals (sigRX1, sigRX2) and to postprocess the received transmission signals (sigRX1, sigRX2) by means of the correction values (γ1, γn, γ2), so that influences of interference variables, in particular phase noise and/or a time offset and/or unknown initial phase positions, are reduced.

16. The radar system of claim 15, wherein the signal processing unit is furthermore designed to: form a comparison signal (sigC21) from a first corrected, received transmission signal (sigCOR1) and the second transmission signal (sigTX2), wherein the comparison signal (sigC21) is transmitted, in particular communicated, from the second transceiver unit (S2) to the first transceiver unit (S1); and form a comparison signal (sigC12) from a second corrected, received transmission signal (sigCOR2) and the first transmission signal (sigTX1), wherein the comparison signal (sigC12) is transmitted, in particular communicated, from the first transceiver unit (S1) to the second transceiver unit (S2); wherein the signal processing unit is designed to reduce deviations of the comparison signals (sigC21, sigC12), which are induced by systematic deviations in the transceiver units (S2, S1), and to use at least one complex value from a first of the two comparison signals or from a signal which was derived from this first comparison signal, to adapt at least one complex value of the second of the two comparison signals or a value of a signal which was derived from this second comparison signal, and thus to form an adapted signal (sigCC), wherein the adaptation takes place in such a way that the vectorial sum or the difference of the complex values is formed or the sum or the difference of the phases of the complex values is formed by a mathematical operation.

17. The radar system of claim 15, wherein the transceiver units (S1, S2) are designed to communicate with a clock unit, wherein the clock unit is designed to provide a system clock and/or a sending starting point to the transceiver units (S1, S2).

18. The radar system of claim 15, wherein each of the transceiver units (S1, S2) furthermore has antenna elements (A1, A2), which are designed to send and to receive at least essentially simultaneously via at least one channel.

Description

[0065] The invention is explained further hereinafter on the basis of nonrestrictive exemplary embodiments with reference to the appended drawings. In the figures:

[0066] FIG. 1 shows two transceiver units, as are found in the prior art;

[0067] FIG. 2 shows frequency curves in the baseband for non-synchronized clock sources of the transceiver units as are found in the prior art;

DERIVATION OF THE SIGNAL MODEL

[0068] The signal model is derived hereinafter on the basis of an exemplary embodiment of the radar system R according to the invention. In this exemplary embodiment, the radar system R has two transceiver units S1, S2, which send and receive transmission signals sigTX1, sigTX2 modulated according to an OFDM method, wherein the transmission signals sigTX1, sigTX2 have multiple subcarriers subC1, subC2 orthogonal to one another. The orthogonal subcarriers are allocated in this exemplary embodiment in such a way that the transceiver unit S1 sends on the subcarriers n.sub.1=0, 2, 4, . . . , N−2 and transceiver unit S2 sends on the subcarriers n.sub.2=1, 3, 5, . . . , N−1.

[0069] The transmission signals sigTX1, sigTX2 having the subcarriers n.sub.1, n.sub.2 are each received by the other transceiver unit S1, S2, wherein non-occupied subcarriers n.sub.u can be used for a monostatic radar measurement, thus for detecting passive objects (radar targets).

[0070] The occupancy of this exemplary embodiment is shown in FIG. 2, wherein the number of the occupied subchannels is N=8 for better visibility. In practice, the number of the occupied subcarriers N is a significantly greater value than 8, for example, 1024 or more.

[0071] In addition, it is also possible that both transceiver units S1, S2 send on the same subcarriers n.sub.1=n.sub.2. The time offset of the transmission signals sigTX1, sigTX2 can be set matching for this purpose in such a way that homodyne and heterodyne radar signals are separable. Furthermore, a part of the subcarriers nit can also be used simultaneously for communication, which can be advantageous (in particular for the exchange of data or signals).

[0072] Before the actual radar measurement, it is preferably presumed that the time drift, which causes intercarrier interference (ICI), by which the orthogonality condition between the subcarriers is disturbed, is or was corrected. The time drift can result, on the one hand, in an error in the sampling times and, on the other hand, in a frequency error, since the phase-locked loops S102, S202 multiply the existing system clock.

[0073] In addition, it is preferably assumed that the time offset Δτ is sufficiently small, which in particular means that the uniqueness range of a distance measurement, in consideration of the time offset Δτ, is not exceeded. The time offset Δτ can be, for example, at most 1 μs, preferably 100 ns or in particular less.

[0074] Both above-mentioned conditions can be achieved, for example, by a synchronization method, as described, for example in H. Abdzadeh-Ziabari and M. G. Shayesteh, “Robust Timing and Frequency Synchronization for OFDM Systems,” IEEE Trans. Veh. Technol., vol. 2, no. 4, pp. 822-839, 2003.

[0075] In addition, T.sub.OFDM=T+T.sub.G preferably applies for the sending duration of a OFDM symbol, wherein T denotes the elementary symbol duration and T.sub.G corresponds to the duration of a so-called guard interval, also called cyclic prefix.

[0076] The additional time which was added by the guard interval (cyclic prefix) to each OFDM symbol is preferably used to compensate (or reduce) effects which can arise due to multipath propagation. In particular in the case of a rough pre-synchronization, the guard interval can be taken into consideration in addition to a maximum propagation time in the transmission channel and the expected time offset Δτ.

[0077] Furthermore, in particular Δf=1/T applies for the frequency offset Δf of N orthogonal subcarriers. The individual subcarriers n.sub.1, n.sub.2 then send at the frequencies f.sub.n=n.Math.Δf=n/T.

[0078] To take into consideration a possible synchronization error, the times t.sub.1 and t.sub.2 are introduced at the transceiver units S1 and S2. In the baseband, the transmission signals of the two stations S1, S2 can thus be described as

[00001]$\begin{array}{cc}\text{?}& \left(1\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(2\right)\end{array}$

[0079] wherein a total of M successive symbols are sent. The function rect (t/T.sub.0) describes a rectangular window of the duration T.sub.0 and f.sub.n1 and f.sub.n2 are the frequency offset between the subcarriers of the transceiver units S1 and S2.

[0080] In contrast to a communication application or a homodyne radar system for detecting passive objects (radar targets), the modulation of the subcarriers a.sub.1 and a.sub.2 of the respective other station is preferably known. Known symbols or symbol subcarriers are referred to as pilot symbols or pilot subcarriers. The type of modulation of the transceiver units S1 and S2 is preferably selected to be identical. An in-phase quadrature (IQ) modulation is preferred, using a sufficiently low peak to average power ratio (PAPR), the ratio of power to the average power, can be achieved.

[0081] For transmission via a radio channel, the baseband signals are each modulated using the high-frequency carriers

s.sub.1,lo(t.sub.1)=e.sup.j(2πf.sup.x.sup.t.sup.1.sup.+φ.sup.PN1,m.sup.(t.sup.1.sup.)+φ.sup.01,m.sup.)und  (3)

s.sub.2,lo(t.sub.2)=e.sup.j(2πf.sup.x.sup.t.sup.2.sup.+φ.sup.PN2,m.sup.(t.sup.2.sup.)+φ.sup.02,m.sup.)  (4)

[0082] which have the carrier frequency f.sub.c. Since these are spatially distributed clock sources S101, S102 (oscillators), the initial phases φ.sub.01,m and φ.sub.02,m of the symbols are different and unknown. Furthermore, the phase noise processes φ.sub.PN1,m(t.sub.1) and φ.sub.PN2,m(t.sub.2) of the two spatially separated transceiver units S1, S2 are different and uncorrelated. After the sending-side modulation using the high-frequency (HF) carriers, the two transmission signals can be represented as

[00002]$\begin{array}{cc}\text{?}& \left(5\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(6\right)\end{array}$

[0083] The time offset between the two transceiver units can be taken into consideration via the relationship

t.sub.1=t or t.sub.2=t−Δτ  (7)

wherein t corresponds to the time in the physical meaning here. It would also be conceivable that transceiver unit S1 is operated offset by ±Δτ/2 and transceiver unit S2 is operated offset by ±Δτ/2. After using the representation given in equation (7), the following applies for the transmission signals of the two transceiver units S1, S2:

[00003]$\begin{array}{cc}\text{?}& \left(8\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(9\right)\end{array}$

[0084] The signals represented in equations (8) and (9) are sent simultaneously via a radio channel and each received by the other transceiver unit.

[0085] It is assumed here that there is precisely one signal path or at least one dominant signal path. If an arbitrary scenario having random propagation paths is studied, an antenna element is necessary at each transceiver unit S1, S2, which is used simultaneously for sending and receiving.

[0086] Furthermore, it is assumed that the transceiver units (radar units) or the passive objects (radar targets) move at constant velocity.

[0087] A position change during the entire sending sequence can be assumed to be sufficiently small. A runtime in the transmission channel can thus be specified as τ(t)=τ+v.Math.t wherein the relative velocity v results in a carrier frequency-dependent Doppler shift f.sub.D of the reception signals of the transceiver units. Since the channel damping in both directions is assumed to be at least approximately identical and can be expressed by a complex number, it is neglected in the derivation. The reception signals of the transceiver units may thus be specified as follows:

[00004]$\begin{array}{cc}\text{?}& \left(10\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(11\right)\end{array}$

[0088] The received signals are downmixed with the HF signal of the respective transceiver unit S1, S2 in the mixer S105 or S205. For this purpose, an in-phase and quadrature mixer is necessary on the receiving side. The baseband signals after the mixing process and a suitable low-pass filtering can be described as

[00005]$\begin{array}{cc}\text{?}& \left(12\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(13\right)\end{array}$

[0089] The phase noise during an OFDM symbol is assumed to be sufficiently small. Furthermore, it is assumed that the time offset and the runtime in the transmission channel are sufficiently small. The phase noise is therefore strongly correlated in both reception channels and can be approximated as

e.sup.j(φ.sup.PN2,m.sup.(t)−φ.sup.PN3,m.sup.(t−τ+Δτ))≈e.sup.j(φ.sup.PN2,m.sup.(t−Δτ−τ)−φ.sup.PN1,m.sup.)≈1+jε.sub.m(t)  (14)

[0090] Using this approximation, the reception signal in the transceiver unit S1 may be represented as the following:

[00006]$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(15\right)\end{array}$

[0091] In addition, the difference of the initial phases is represented in abbreviated form in equation (15) by the variable

φ.sub.0,m=φ.sub.01,m−φ.sub.02,m  (16)

[0092] For the reception signal of the transceiver unit S2, it is additionally taken into consideration that the sampling is also time delayed, which can be expressed by the (back) transformation t|.fwdarw.t+Δτ. The influence of this time delay on the Doppler shift is assumed to be practically negligible and the phase noise is also approximated, which results in

[00007]$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(17\right)\end{array}$

[0093] The signals described in equations (15) and (17) are now sampled, wherein f.sub.s=T/N is to apply for the sampling frequency.

Pre-Correction Step

[0094] The signals represented in equations (15) and (17) in particular have the following interference variables: [0095] time offset Δτ; [0096] unknown phase shift φ.sub.0,m; and/or [0097] phase noise (fast time) ε.sub.m(t);

[0098] which are corrected first. That is, a fast Fourier transform (FFT) of the received and sampled signals is performed in the respective transceiver unit S1, S2.

[0099] In the present exemplary embodiment, in a pre-correction step VS2, first the effect of the above-mentioned interference variables on the signals received in the transceiver units is reduced (compensated), wherein the interference variables of both transceiver units are moved toward one another (equalized).

[0100] In a reconstruction step VS4, an accurate estimation of distance, velocity, and initial phase is possible by a comparison of the (corrected) signals of the two transceiver units.

[0101] In a further exemplary embodiment, a comparison signal can be generated, which has comparable properties like, for example, signals from a homodyne mixing process in homodyne radar systems.

[0102] To calculate the interference variables, first an FFT of the signals from equations (15) and 17) is calculated. The expanded sending symbols, thus the cyclic prefix, are removed. At the two transceiver units S1 and S2, neglecting ε.sub.m(t) and with f.sub.n1=2.Math.n and f.sub.n2=2.Math.n+1, the following discrete signals result

[00008]$\begin{array}{cc}\text{?}& \left(18\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(19\right)\end{array}$

[0103] An effect of the Doppler effect is assumed to be small, which has the result that the phase shift caused in this way only affects successive symbols. If this assumption does not apply, the Doppler shift in particular results in ICI. Compensation methods for the ICI occurring due to Doppler shift are found in J. Lim, S. R. Kim, and D. J. Shin, “Two-Step Doppler Estimation Based on Intercarrier Interference Mitigation for OFDM Radar,” IEEE Antennas Wirel. Propag. Lett., vol. 14, pp. 1726-1729, 2015, and by G. Hakobyan and B. Yang, “A Novel Intercarrier-Interference Free Signal Processing Scheme for OFDM Radar,” IEEE Trans. Veh. Technol., vol. 67, no. 6, pp. 5158-5167, 2018, wherein the latter method does not require specific coding, however, due to which it can preferably be combined in an exemplary embodiment with the method according to the invention.

[0104] In particular the distance between the transceiver units and/or passive objects or the runtime in the transmission channel can be calculated by an FFT over the index n.

[0105] In a further exemplary embodiment, the result of the FFT is improved, so that in particular an accurate and non-integer value is obtained, in that the result of the FFT is preferably ascertained by zero padding and parabolic interpolation. Furthermore, the use of a window function in multiple signals can result in a suppression of secondary maxima.

[0106] In addition, the accuracy of the results can be further increased in particular by averaging over multiple symbols, wherein first the time offset Δτ has to be determined. The determination of the time offset Δτ can preferably be enabled via a comparison of both received signals.

[0107] After the FFT of the received signals has been calculated, in the frequency spectrum, the spectral frequency peaks of the transceiver unit S1 and the transceiver unit S2 can be determined:

k.sub.1=2N(τ+Δτ)/Tund  (20)

k.sub.2=2N(τ−Δτ)/Tund  (21)

[0108] From the previously determined frequency peaks k.sub.1, k.sub.2, two correction values γ.sub.1, γ.sub.n, can then be determined:

[00009]$\begin{array}{cc}\text{?}& \left(22\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(23\right)\end{array}$

[0109] Since the transceiver unit S2 sends in this exemplary embodiment on the odd subcarriers n.sub.2, the third correction value γ.sub.2

[00010]$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(24\right)\end{array}$

[0110] is determinable independently of the frequency peaks k.sub.1, k.sub.2, thus the measured scenario. The received transmission signals, which are specified in equations (15) and (17), are now corrected using the correction values γ.sub.1, γ.sub.n, γ.sub.2:

[0111] First, for example, the offset of the subcarriers in the first transceiver unit S1 can be corrected using the correction value γ.sub.2 from equation (24). With f.sub.n=(2.Math.n+1)/T, this results in the following:

[00011]$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(25\right)\end{array}$

[0112] The two further correction values γ.sub.1, γ.sub.n from equations (22) and (23) can now be applied to the signal from equation (25). This results in the following:

[00012]$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(26\right)\end{array}$

[0113] If the correction values γ.sub.1 and γ.sub.n are applied to the received signal of the second transceiver unit S2, this results in the following:

[00013]$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(27\right)\end{array}$

[0114] In the last two equations, the known data symbols a.sub.1 and a.sub.2 were replaced by a. After the application of the previously determined correction values γ.sub.1, γ.sub.n, γ.sub.2, the pre-correction step is completed. It is apparent from equations (26) and (27) that the interference terms now contained in the corrected are complex conjugated to one another and all terms dependent on τ and f.sub.D are in phase.

Reconstruction Step

[0115] In a reconstruction step, further calculations can be carried out using the corrected signals z.sub.1(t), z.sub.2(t) from equations (26) and (27), wherein the discrete time sampling using f.sub.s=T/N has the result that the time variable t is replaced in each case by k.Math.T/N. Sample values (data) of the transceiver units S1 and transceiver units S2 after a discrete Fourier transform (DFT) or an FFT can be represented as

[00014]$\begin{array}{cc}\text{?}& \left(28\right)\end{array}$$\begin{array}{cc}\text{?}\text{?}\text{indicates text missing or illegible when filed}& \left(29\right)\end{array}$

[0116] The phase noise can result, on the one hand, in a constant phase offset per symbol, which is expressed by the variable φ.sub.0,m.sup.n and on the other hand, in ICI, which is represented by the variable ε.sub.m[n,m].

[0117] The signal model from equations (28) and (29) can possibly be further processed using further methods, for example, as in M. Gottinger, F. Kirsch, P. Gulden, and M. Vossiek, “Coherent Full-Duplex Double-Sided Two-Way Ranging and Velocity Measurement Between Separate Incoherent Radio Units,” IEEE Trans. Microw. Theory Tech., vol. 67, no. 5, pp. 2045-2061, 2019, so that possible remaining interference variables can be compensated or the influence of interference variables can be reduced. A detection of distance, relative velocity, and/or runtime-dependent phase between the transceiver units S1 and S2 and/or to passive objects (radar targets) is enabled.

LIST OF REFERENCE SIGNS

[0118] S1 first transceiver unit [0119] S2 second transceiver unit [0120] S101 clock source (local oscillator) of the first transceiver unit [0121] S201 clock source (local oscillator) of the second transceiver unit [0122] S102 phase-locked loop (PLL) of the first transceiver unit [0123] S202 phase-locked loop (PLL) of the second transceiver unit [0124] S103 modulator of the first transceiver unit [0125] S203 modulator of the second transceiver unit [0126] S104, S105 mixer of the first transceiver unit [0127] S204, S205 mixer of the second transceiver unit [0128] S106 analog-to-digital (A/D) converter of the first transceiver unit [0129] S206 analog-to-digital (A/D) converter of the second transceiver unit [0130] sigRX1 transmission signal of the first transceiver unit [0131] sigRX2 transmission signal of the second transceiver unit [0132] sigRX1 reception signal of the first transceiver unit [0133] sigRX2 reception signal of the second transceiver unit [0134] sigTX1b baseband transmission signal of the first transceiver unit [0135] sigTX2b baseband transmission signal of the second transceiver unit [0136] sigRX1b baseband reception signals of the first transceiver unit [0137] sigRX2b baseband reception signals of the first transceiver unit [0138] A1 HF antenna of the first transceiver unit [0139] A2 HF antenna of the second transceiver unit [0140] R radar system [0141] n.sub.1 subcarrier of the first transceiver unit [0142] n.sub.2 subcarrier of the second transceiver unit [0143] n.sub.u subcarrier which is not used by any transceiver unit [0144] τ(t) radio channel